Patent Description:
Interferences on CDMA signals emitted by GNSS satellites are prone to impulsed noise environments, for example in the vicinity of ignition systems, power lines, current switches or microwave inferred surroundings. Interferences occur usually in a bursed like form, such that they cannot be modelled as Gaussian. This affects the reception performance of a CDMA signal. This can be circumvented by a blanker. Commonly, a blanker sets received signal samples to zero, when the signal samples comprise high values of power. These high values indicate interferences. Typically, a blanker uses two fixed thresholds BTH+ and BTH-, which are symmetrical to zero. The high power value sample exceeds this threshold and is consequently set to zero by the blanker. CDMA signal receivers are usually equipped with such a blanker.

In scenarios, where no interferences occur, the blanker still sets signals to zero, which are affected by large thermal noise. However, this may lead to an undesired reduction of signal to noise and interference ratio.

In the acquisition phase during reception of the CDMA signal, for example when the receiver tries to synchronize the received signal with a replica, a right code delay and Doppler offset are searched. The positions of the chip transitions and the chip amplitude are not known. Consequently, the blanker needs to be modified in order to maximize possibility of detection.

Prior art can be found in <CIT> which generally relates to a method and device for signal acquisition of a generalized BOC-modulated signal, in <CIT> which generally relates to noise distribution shaping for signals, particularly CDMA signals, with mitigation of artefact signals, in <CIT> which generally relates to blanking using signal-based thresholding schemes and in <CIT> which generally relates to noise distribution shaping for signals, particularly spread spectrum signals like CDMA signals, with improved robustness.

According to a first aspect of the present invention, a method of noise distribution shaping for signal acquisition of a Code Division Multiple Access (CDMA) signal comprises demodulating the CDMA signal by multiplying with a carrier signal modulated with an estimated Doppler frequency. The carrier of the carrier signal is the CDMA signal carrier. The method further comprises generating a test signal. The test signal has a spreading sequence according to the CDMA signal. The test signal is shifted by an estimated code delay. The method further comprises modifying the noise distribution of the CDMA signal to generate a modified CDMA signal by blanking samples of the CDMA signal, when the samples of the CDMA signal exceed an upper or lower blanking threshold. The upper and lower blanking thresholds are offset by a square root of an estimated power (Pest) of the CDMA signal multiplied by a chip value polarity of the generated test signal and a predefined scaling factor, α. The upper blanking threshold is further offset by a positive value and the lower blanking threshold is further offset by a negative value, and the positive and negative values are set by a scaled standard deviation of thermal noise and circumferential interference.

The method further comprises acquiring the CDMA signal, when a correlation between the modified CDMA signal and the generated test signal is above a predefined detection threshold.

Setting the rising and falling edges of the upper and lower blanking thresholds according to the test signal's rising and falling edges leads to more precise correlation results with the CDMA signal. α can be positive, negative, constant, time dependent, or changing its sign over time as function of the generated test signal.

The carrier signal can further be shifted by an estimated carrier phase. This carrier phase can be modulated on the carrier signal, which is multiplied with the CDMA signal.

The estimated power can be an estimate of the CDMA signal power or is an estimate being at least <NUM> times greater than the CDMA signal power. The estimated power can further be greater than the CDMA signal power.

This can lead to better receiver operating characteristics during acquisition.

The number of blanked samples can indicate detection according to the estimated carrier phase, the estimated Doppler frequency, the estimated code delay and/or the estimated power. The number of blanked samples can be acquired by a signal processor. The number of blanked samples can be stored in a storage device.

The demodulating step can comprise separating the CDMA signal into an inphase and a quadrature component by multiplying the CDMA signal with two carrier signals phase-shifted by <NUM>-degree to each other. Each of the two carrier signals can have a carrier frequency of the CDMA signal. The carrier frequency can be shifted by an estimated Doppler frequency. The step of modifying can comprise modifying the noise distribution of the CDMA signal to generate a modified CDMA signal by blanking samples of the inphase and quadrature components, when the samples of the inphase and quadrature components exceed an upper or lower blanking threshold respectively. The upper and lower blanking thresholds can be offset by a square root of an estimated power (Pest) of the CDMA signal. The CDMA signal can be multiplied by a chip value polarity of the generated test signal. The CDMA signal can be multiplied by a predefined scaling factor, α. The step of acquiring can further comprise acquiring the CDMA signal, when a correlation between the CDMA signal according to the inphase and quadrature component and the generated test signal is above a predefined detection threshold.

The upper and lower blanking thresholds can be offset according to the estimated Doppler frequency.

The upper and lower blanking thresholds can be offset according to the estimated carrier phase. This has the advantage to ensure that the upper and lower blanking thresholds both applied on the inphase and quadrature branch are synchronized to the actual CDMA signal.

The corresponding upper and lower thresholds for the inphase and quadrature component of the CDMA signal can be further expressed as:.

BTHi+ can describe the upper blanking threshold for the inphase component. BTHi-can describe the lower blanking threshold for the inphase component. BTHq+ can describe the upper blanking threshold for the quadrature component. BTHq- can describe the lower blanking threshold for the quadrature component.

The predefined detection threshold can be set according to a probability of false alarm or a probability of missed detection.

According to a second aspect of the present invention, a computer program is provided comprising instructions which, when the program is executed by a computer, cause the computer to carry out the steps of a method according to the first aspect.

According to a third aspect of the present invention, a storage device stores a computer program according to the second aspect.

According to a fourth aspect of the present invention, a device for noise distribution shaping and signal acquisition of a Code Division Multiple Access (CDMA) signal comprises a demodulator, a signal generator, a blanker and an acquisition processor. The demodulator is adapted to demodulate the CDMA signal by multiplying with a carrier signal modulated with an estimated Doppler frequency. The carrier of the carrier signal is the CDMA signal carrier. The signal generator is adapted to generate a test signal. The test signal has a spreading sequence according to the CDMA signal. The test signal is shifted by an estimated code delay. The blanker is adapted to modify the noise distribution of the CDMA signal to generate a modified CDMA signal by blanking samples of the CDMA signal, when the samples of the CDMA signal exceed an upper or lower blanking threshold. The upper and lower blanking thresholds are offset by a square root of an estimated power (Pest) of the CDMA signal multiplied by a chip value polarity of the generated test signal and a predefined scaling factor, α. The upper blanking threshold is further offset by a positive value and the lower blanking threshold is further offset by a negative value, and the positive and negative values are set by a scaled standard deviation of thermal noise and circumferential interference.

The acquisition processor is adapted to acquire the CDMA signal, when a correlation between the modified CDMA signal and the generated test signal is above a predefined detection threshold.

The blanker can be adapted to be a passive matched filter. Coefficients of the passive matched filter can correspond to the upper and lower blanking thresholds. This has the advantage to avoid a penalizing effect on the mean time to acquire the CDMA signal.

Even if the foregoing described aspects with respect to the method were only described for the method, these aspects can further relate to the foregoing described device and vice versa.

The present invention will be further described in more detail hereinafter with reference to the figures. The figures schematically show:.

<FIG> schematically shows a GNSS receiver <NUM> with main functional blocks. A Code Division Multiple Access (CDMA) signal is first received via an antenna <NUM>. The CDMA signal is then fed to a pre-amplifier stage <NUM>. The pre-amplifier stage <NUM> aims at increasing the received signal to a level compatible with the following components of the GNSS receiver <NUM>, the receiver front-end respectively. The pre-amplifier stage <NUM> can comprise a single or several amplifiers mounted in cascade. The first amplifier arranged downstream to the antenna <NUM> is usually a low noise amplifier (LNA) and is characterized by a small noise figure (NF). The received signal, which is usually in a high frequency domain, for example <NUM> for GPS C/A signals, is firstly down converted to an intermediate frequency (IF). This downconversion is usually performed before an analogue to digital converter (ADC) <NUM> via a down converter <NUM>. This step can also be performed in the digital domain, if the sampling frequency of the A/D converter <NUM> is large enough following the Nyquist condition. In order to perform a down conversion from an RF signal to an IF signal, the RF signal must be multiplied with a cosine at a frequency corresponding to RF-IF, for example <NUM> - IF. A reference oscillator <NUM> provides a frequency synthesizer <NUM> with a reference frequency, which frequency synthesizer <NUM> is then outputting a desired (RF-IF) signal for the down converter <NUM> for down conversion from RF to IF. After the signal has been down converted, it is in an IF domain, which is then sampled via the A/D converter <NUM>. Further, an automatic gain control (AGC) <NUM> is used to adapt the power of the received signal in mere real time. The AGC <NUM> monitors the power level of the samples and provides information for multiplying the received signal in RF domain with a variable gain. The part before the A/D converter <NUM> is usually called the analogue front-end, wherein the part downstream to the analogue front-end is usually called digital front-end. In the digital front-end, the output of the ADC <NUM> is then fed to a blanker <NUM>. The blanker aims at setting samples to zero which contain large interference signals comprised in the received signal. In <FIG> the blanker is implemented in the digital front-end, however the blanker can also be implemented into an analogue front-end. After the blanker <NUM>, the remaining digital samples are then injected to N digital receiver channels, where N represents the number of line-of-sight satellite signals. As for the case of satellite signals, they need to be tracked in order to make a position estimation. This requires position and timing performance, which is usually more accurate using more digital receiver channels <NUM>. Each digital receiver channel <NUM> aims at processing the IF signals, by first wiping-off the remaining carrier frequency of the received signal. This is necessary for correlator channels provided downstream to the digital receiver channels <NUM>. The correlated channels are necessary for signal acquisition but also for code and carrier estimations and navigation data demodulation. The acquisition process is further described in the following <FIG>.

<FIG> schematically illustrates a device for signal acquisition without a blanker in the digital domain. A digital IF signal from the block Digital IF <NUM> is provided after an analogue to digital conversion. The aim of the acquisition of a CDMA signal is to estimate roughly a code delay and a Doppler offset of the received signal. For this purpose, a set of code delay estimates and Doppler offset estimates are tested. For each pair of code/Doppler estimates, a replica is firstly generated by shifting the spreading sequence with the corresponding estimated code delay and multiplying the shifted sequence with a carrier modulated at the estimated Doppler frequency. Afterwards, the CDMA signal is correlated with the replica generated with the estimated code delay and Doppler frequency. The correlation is then squared via an incoherent integration and then compared to a threshold either set according to a probability of false alarm or according to a probability of missed-detection. The undesired false alarm is caused by unavoidable thermal noise. When no information about a code delay or a Doppler frequency for the CDMA signal is available, all estimated code delays corresponding to the whole spreading sequence have to be searched. The typical order of magnitude for the estimated Doppler frequencies during a cold start is ±<NUM>. To limit the code delay and Doppler frequency miss-alignment losses during the correlation process to a few decibels, one estimated code delay is proposed every Tc/<NUM> or TC/<NUM>, wherein Tc is the chip duration of a corresponding spreading sequence of the CDMA signal. An estimated Doppler frequency is tested every <NUM> to <NUM>. As an example for the GPS C/A signal having a spreading sequence containing <NUM> chips, at least <NUM> estimated code delays can be tested. Those sampling intervals for the estimated code delay and Doppler frequencies are called respectively code and Doppler binwidths. To improve the performances for this conventional acquisition, it is possible to add non-coherently the detector output for successive correlation of the received signal with the same replica generated with the estimate code delay and Doppler frequency offset to be tested. Due to the non-coherent summations, so called squaring losses have to be taken into account. To avoid such squaring losses, it is possible to increase the coherent correlation time. This means, that the replica and received signals are correlated over longer correlation time, but in that case the code delay miss-alignment losses will increase. The principle is described in <FIG> in more detail with the corresponding functional blocks. The digital IF <NUM> delivers the CDMA signal, which is then demodulated via a cosine <NUM> and a sine <NUM>. The cosine <NUM> and the sine <NUM> are two functions <NUM> degree phase shifted to each other, which lead the CDMA signal to have an inphase and quadrature component. Further, the square grid <NUM> symbolizes estimated code delays and Doppler frequencies. An estimated Doppler frequency is modulated with the CDMA signal carrier in order to derive the inphase and quadrature component by demodulation of the digital IF signal. The test signals, the so called replicas, with the estimated code delays are generated via the Code NCO <NUM> and the Code Generator <NUM>. The test signals are then correlated with the inphase and quadrature component of the CDMA signal via a coherent integration <NUM>. In the acquisition processor <NUM> an incoherent integration and summation leads to a comparing result which is then compared to a detection threshold. In the following <FIG>, different replicas (test signals) and the corresponding received signal are schematically illustrated with a corresponding correlation function.

<FIG> schematically illustrates diagrams of different replicas of a CDMA signal. A typical example of a situation for signal acquisition via a correlation process using replicas, so called test signals, in order to derive a correlation function. The received signal S30 is illustrated on top of <FIG> with <NUM> tested code delay hypothesis, the so called estimated code delay. Those code delay hypothesis are Tc/<NUM> part. This is symbolized via crosses. R30, R31, R32, R33 until R3M describe the <NUM> different replicas. These replicas lead to a correlation function resulting in C30. The best case is represented by the exact estimated code delay at the peak of the correlation function. The further estimated code delays lead to smaller correlation results indicating that the estimated code delay is not matching the CDMA signal. The signal acquisition is then extended by an adaptive blanker in <FIG>.

<FIG> schematically illustrates the use of an adaptive blanker according to an embodiment of the present invention. S40 illustrates a received CDMA signal with respective upper and lower blanking thresholds BTH+ and BTH-. BTH+ describes the upper blanking threshold, wherein BTH- describes the lower blanking threshold. The remaining signals R41 until R4M describe the test signals with corresponding estimated code delays. Two blanking thresholds BTH+ and BTH- are built based on an initial blanking threshold B0 and an offset β. According to the claimed invention, the initial blanking threshold B0 is expressed as a product of the standard deviation of the thermal noise and additional interference contribution (σnoise) comprised in the received CDMA signal and a scaling factor KB0, such that B0 = KB0 * σnoise. The offset β can be expressed as a product between the square root of an estimated received signal power Ps,est, the polarity of the chips of the spreading sequence, C(T), and a scaling factor α. T describes the time expressed in the time scale of the receiver. This leads to β=α * square root (PS, est (t)) * C(T). The upper blanking threshold BTH+ and the lower blanking threshold BTH- can be set as follows: <MAT> <MAT>.

Both thresholds BTH+ and BTH- vary according to the chip polarity, for example the chip value, and vary at each chip transition, which are spaced by Tc, if a polarity of consecutive chips are of opposite sign. Under consideration of a true code delay of the CDMA signal between two estimated code delays, for example Tc/<NUM>, these two closest estimated code delays are the ones which will lead to highest values of the detector output. This is referred to maximal energy. If the detector output recites below the detection threshold, then a missed-detection can be declared, else a detection is achieved. All other <NUM>-<NUM> = <NUM> test signals correspond to wrong alignments of the CDMA signal and the corresponding <NUM> wrong test signals. If the detector outputs recite above the detection threshold, false alarm is declared. In the best case scenario, if one of the <NUM> estimated code delays exactly matches the true code delay of the CDMA signal, the corresponding correlation function will be maximal at a peak. If the detector output for this exact match recites below the detection threshold a missed-detection is declared. For the <NUM>-<NUM> = <NUM> other tested estimated code delays, if the detector outputs recite above the detection threshold, a false alarm is declared. In <FIG> the upper and lower blanking thresholds are perfectly symmetrical to the chip values. This corresponds to the scenario, wherein the estimated power exactly matches the true CDMA signal power Pest = P and α = <NUM>. Both upper and lower blanking thresholds are very close to the chip amplitude. This means, that either σnoise is very small with respect to the CDMA signal amplitude [square root (P) (and KB0 = <NUM>)], which is not a usual case for satellite navigation signals. The usual case for satellite navigation signals is σ<NUM>noise < < P, or KB >> <NUM> with σ<NUM>noise << P. Other settings for blanking thresholds are schematically illustrated in <FIG>.

<FIG> schematically illustrates two different blanking threshold adaptation settings. Two different scenarios of applied blanking thresholds are illustrated in the respective upper R50 and lower R51 plot. In R50, the upper and lower blanking thresholds are aligned above or below with respect to the functional slope of the test signal. In the case of R51, the upper and lower thresholds are above the test signal for positive chip values of the spreading sequence and are below the functional slope for negative chip values of the spreading sequence. In the case of R50 the scaled estimated amplitude according to a square root of an estimated power corresponds exactly to the received signal power. In the case of R51 the scaled estimated amplitude according to a square root of an estimated power is much greater than the actual CDMA signal power. In both cases R50 and R51 the scaled noise distribution is much greater than the actual signal power of the CDMA signal. The upper and lower blanking thresholds BTH+ and BTH- in <FIG> are no more symmetrical with respect to the true chip amplitude [square root (P (t))], contrary to the scenario in <FIG>. The reason lies in that the receiver also ignores the exact signal power or a signal amplitude in a cold acquisition, for example without a priori information about the code delay, Doppler frequency or signal power. In the same way that the receiver tests different estimated code delays during acquisition, which leads to an adaptation of the upper and lower blanking thresholds, the receiver also has to test different estimated signal amplitudes according to a square root of an estimated power. This will further discipline the amplitude of the upper and lower blanking thresholds. Hence, in acquisition processing with such an adaptive blanker, a third degree of freedom with the signal power is introduced besides estimated code delays and Doppler frequencies. Although, intuition would have expected at the acquisition performances are the best when the tested signal power (Pest) is close or very close to the actual signal power of the CDMA signal (Pest ~ P), it appears from simulations that in fact detection performances are best when the tested signal power is significantly larger than the actual signal power of the CDMA signal, for example <NUM> times larger than the actual signal power of the CDMA signal. The following <FIG> provides the adaptation of a conventional digital receiver channel used for acquisition with the implementations of an adaptive blanker introduced in <FIG> and <FIG>.

<FIG> schematically illustrates the acquisition process for a CDMA signal according to an embodiment of the present invention. The digital IF <NUM> delivers the CDMA signal which is then demodulated by the COS map <NUM> and SIN map <NUM> leading to an inphase and quadrature component. The COS map <NUM> and the SIN map <NUM> demodulate the digital IF signal to a base band via a Carrier NCO <NUM> comprising additional estimated Doppler frequencies and estimated phase shifts from the illustrated cubicle <NUM> and estimated phase shifts <NUM>. A noise distribution estimator <NUM> estimates the noise distribution, which is then fed to an adaptive blanker <NUM>. The adaptive blanker is further provided by the cubicle <NUM> with the estimated signal power Pest. After blanking of the inphase and quadrature component of the CDMA signal, the inphase and quadrature components of the CDMA signal are correlated with the test signals. The test signals have been generated by a code NCO <NUM> and a code generator <NUM>. The cubicle <NUM> schematically illustrates and symbolizes the three dimensions which have to be investigated for signal acquisition according to the present invention. After correlation the signal can be detected, when the detector indicates that the test signal is the most similar signal with respect to the CDMA signal. If the detector does not give relevant feedback, the acquisition process can include another detection process. This process comprises detection with the number of blanked samples. The number of blanked samples can indicate alignment of the received signal and the test signal. This kind of active acquisition is characterized by the fact that a snapshot of the received CDMA signal is correlated over a coherent integration time with a test signal generated with an estimated code delay, estimated Doppler frequency and estimated power for the same integration time. Hence, for each new set of estimated code delay, estimated Doppler frequency and estimated power to be tested, a new snapshot of the received CDMA signal is tested. The advantage of the active correlation is that the detector outputs are independent from each newly tested set of hypotheses. This yields better statistical results according to the detector. The drawback is that for each newly tested hypothesis the integration time multiplied with a number of non-coherent summations, also called dwell time, has to be spent. This reduces the mean time to acquire the CDMA signal. To avoid this penalizing effect on the mean time to acquire the CDMA signal, another architecture is illustrated in <FIG>.

In <FIG> another architecture called passive matched filter is introduced. In the passive matched filter <NUM> the received CDMA signal is directed to a delay line made of shift registers. The output of each shift register is then multiplied with a coefficient corresponding to the spreading code to be tested. The number of shifter registers and coefficients correspond to the number of estimated code delays to be tested per chip multiplied with the number of chips in the spreading sequence. For example <NUM> shift registers in the case of two code hypothesis for each chip of the GPS C/A sequence. Then, the outputs of the shift registers multiplied by the chip coefficients are added and directed to the power detector <NUM>. This is performed for both inphase and quadrature components for the active acquisition technique. Each new entering sample, a new estimated code delay is tested. This is symbolized by different arrows originating from the code delay grid <NUM>. This principle is further extended in <FIG>.

<FIG> schematically illustrates the acquisition process according to an embodiment of the present invention with added noise distribution estimator <NUM> and the passive matched filter <NUM> according to <FIG>. The code delay grid of <FIG> is replaced by a cubicle <NUM> corresponding to the cubicle in <FIG>. The principle corresponds to the principle, which was introduced in <FIG>. The coefficient cells, so called coefficients, are replaced by another cell implementing the two blanking thresholds, the upper and lower blanking thresholds. These upper and lower blanking thresholds are derived from the chip polarity (c(t)), the power to be tested (Ps, est), the estimated noise standard deviation (σnoise). A more detailed illustration is given by <FIG> schematically illustrates a part of the passive matched filter <NUM> for further illustration. Even if the estimated Doppler frequency would exactly match the actual (true) Doppler frequency of the received CDMA signal, it cannot be ensured that the major part of the signal power recites on the inphase branch, which is referred to the inphase component of the CDMA signal. It can further not be ensured that the quadrature branch, which is referred to the quadrature component, merely contains a residual part of the signal power. In order to ensure that the upper and lower blanking thresholds both applied on the inphase and quadrature branch are synchronized to the actual signal, a fourth degree of freedom, namely a carrier phase is introduced. This is shown in <FIG> and <FIG>. In that case both upper and lower blanking thresholds for the inphase and quadrature branches are expressed as:.

This principle can then be extended by analyzing a number of blanked samples for each set of estimated code delay, estimated Doppler frequency and estimated power. As a consequence on <FIG> and <FIG>, the number of blanked samples is collected for each branch, the inphase and quadrature branch. Then, this number is compared to a threshold which will help taking a decision regarding the right estimated code delay, estimated Doppler frequency and estimated power. The number of blanked samples can also represent a figure of merit which enables a further detection method of the right estimation of code delay, Doppler frequency and signal power of the received CDMA signal.

Claim 1:
A method of noise distribution shaping for signal acquisition of a Code Division Multiple Access, CDMA, signal the method comprising:
demodulating (S101) the CDMA signal by multiplying with a carrier signal modulated with an estimated Doppler frequency, wherein the carrier of the carrier signal is the CDMA signal carrier;
generating (S102) a test signal having a spreading sequence according to the CDMA signal, wherein the test signal is shifted by an estimated code delay;
modifying (S103) the noise distribution of the CDMA signal to generate a modified CDMA signal by blanking samples of the CDMA signal, when the samples of the CDMA signal exceed an upper or lower blanking threshold, wherein the upper and lower blanking thresholds are offset by a square root of an estimated power, Pest, of the CDMA signal multiplied by a chip value polarity of the generated test signal and a predefined scaling factor, α, wherein the upper blanking threshold is further offset by a positive value and the lower blanking threshold is further offset by a negative value, and wherein the positive and negative values are set by a scaled standard deviation of thermal noise and circumferential interference; and
acquiring (S104) the CDMA signal, when a correlation between the modified CDMA signal and the generated test signal is above a predefined detection threshold.