Patent Description:
Direct conversion transceivers have the advantage over intermediate frequency transceivers that they have a simplified hardware configuration, are less cost- and power-consuming, and may be implemented in integrated circuits. However, one of the main disadvantages of the direct conversion transceivers is an I/Q imbalance which significantly deteriorates their characteristics. The occurrence of the I/Q-imbalance in the direct conversion transceivers is associated, for example, with the use of analog up/down-converters and low-pass filters. In addition, for broadband devices, it is necessary to take into account the frequency dependence of the I/Q imbalance in a signal modulation band.

The existing methods for I/Q imbalance estimation and compensation rely on using additional external hardware (e.g., such as an envelope detector) that is often unavailable for regular data transmission. Moreover, the existing methods usually involve performing the I/Q estimation and compensation in two steps: for a transmitter path first and then for a receiver path. Therefore, it takes relatively long time to calibrate the direct conversion transceivers before their normal use.

For example, <CIT>refers to systems and techniques relating to calibrating I/Q mismatches in communication systems. In some implementations, a reference signal to be transmitted by a transmitter (Tx) is identified. A loopback signal corresponding to the reference signal is generated by passing the reference signal through the TX and a receiver (Rx). The loopback signal includes an additional signal that distinguishes an in-phase/quadrature (I/Q) mismatch caused by the Tx (Tx I/Q mismatch) from an I/Q mismatch caused by the Rx (RX I/Q mismatch). A set of Tx I/Q mismatch parameters and a set of Rx I/Q mismatch parameters are determined based on the reference signal and the loopback signal, using the additional signal. A Tx signal is calibrated based on the set of Tx I/Q mismatch parameters, while a Rx signal is calibrated based on the set of Rx I/Q mismatch parameters, independently from the calibrating the Tx signal.

This summary is not intended to identify key features of the present disclosure, nor is it intended to be used to limit the scope of the present disclosure.

It is an objective of the present disclosure to provide a solution that enables I/Q estimation and compensation in a communication apparatus (especially, a direct conversion transceiver).

The objective is achieved by the features of the independent claims in the appended claims. Further embodiments and examples are apparent from the dependent claims, the detailed description, and the accompanying drawings.

According to a first aspect, a communication apparatus is provided. The communication apparatus comprises a transmitter (TX), a receiver (RX), a phase-locked loop (PLL), and a RF loopback path. The PLL is configured to generate a TX local oscillator (LO) signal and a RX LO signal, and to successively provide a first phase shift and a different second phase shift between the TX LO signal and the RX LO signal. The TX is configured to generate a reference signal and to obtain a TX RF signal by quadrature mixing of the reference signal with the TX LO signal. The RF loopback path is configured to output the TX RF signal to the RX as a RX RF signal. The RX is configured to obtain a representation of the reference signal by quadrature mixing of the RX RF signal with the RX LO signal. The TX comprises an adaptive filter configured to:.

With this configuration, the communication apparatus may estimate the combined TX/RX frequency-dependent I/Q imbalance without having to use additional external hardware (e.g., such as an envelope detector). The knowledge of the contributions made by the TX and the RX in the combined TX/RX frequency-dependent I/Q imbalance may be then used to compensate the combined TX/RX frequency-dependent I/Q imbalance itself in the time domain.

In one embodiment of the first aspect, the TX is configured to generate the reference signal as a pseudo-noise signal or an Orthogonal Frequency-Division Multiplexing (OFDM) signal. By so doing, it is possible to study the combined TX/RX frequency-dependent I/Q imbalance in a wideband of the communication apparatus more efficiently. Furthermore, the OFDM signal is a signal used for regular data transmission in a wireless communication system, thereby meaning that the apparatus according to the first aspect does not need to generate any special signal for the purpose of the I/Q imbalance estimation and compensation in the time domain.

In one embodiment of the first aspect, the adaptive filter is configured to obtain the first estimate of the combined TX/RX frequency-dependent I/Q imbalance and the second estimate of the combined TX/RX frequency-dependent I/Q imbalance by using one of a Least-Mean-Square (LMS) algorithm, a normalized LMS algorithm, a leaky LMS algorithm, and a recursive least squares algorithm. This may make the apparatus according to the first aspect more flexible in use because it may decide on one of these several algorithms depending on particular application.

In one embodiment of the first aspect, the second phase shift differs from the first phase shift by <NUM> degrees. This preferred phase shift provides the highest accuracy of the I/Q imbalance estimation and compensation.

In one embodiment of the first aspect, the TX further comprises a pre-compensating finite impulse response (FIR) filter configured to compensate the determined contribution made by the TX in the combined TX/RX frequency-dependent I/Q imbalance, and the RX comprises a post-compensating FIR filter configured to compensate the determined contribution made by the RX in the combined TX/RX frequency-dependent I/Q imbalance. By using such FIR filters, the compensation of the contributions made by each of the TX and the RX in the combined TX/RX frequency-dependent I/Q imbalance may be performed more efficiently.

In one embodiment of the first aspect, each of the pre-compensating FIR filter and the post-compensating FIR filter has a filter length predefined based on a required accuracy of I/Q imbalance contribution compensation and/or a required amount of computational resources to be used for the I/Q imbalance contribution compensation. This may provide an acceptable tradeoff between the accuracy of I/Q imbalance contribution compensation and the computational resources used therefor.

In one embodiment of the first aspect, the pre-compensating FIR filter and the post-compensating FIR filter are configured to simultaneously compensate the determined contributions made by the TX and the RX, respectively, in the combined TX/RX frequency-dependent I/Q imbalance. By so doing, it is possible to reduce the time and number of measurements and calculations which are required for the I/Q imbalance compensation in the apparatus according to the first aspect.

In one embodiment of the first aspect, the pre-compensating FIR filter and the post-compensating FIR filter are configured to compensate the determined contributions made by the TX and the RX, respectively, in the combined TX/RX frequency-dependent I/Q imbalance when the communication apparatus operates in a time division duplexing (TDD) mode. This may make the apparatus according to the first aspect more flexible in use. In particular, the fact that the apparatus according to the first aspect may perform the I/Q imbalance estimation and compensation in the TDD mode means that it may do the same in a frequency division duplexing (FDD) mode. It should be noted that the opposite is not true, i.e. the possibility of the I/Q imbalance estimation and compensation in the FDD mode does not mean the possibility of doing the same in the TDD mode.

In one embodiment of the first aspect, the TX further comprises a nonlinear power amplifier configured to amplify the TX RF signal. In this embodiment, the apparatus according to the first aspect further comprises a digital signal processing unit configured to reduce nonlinear distortions occurring in the amplified TX RF signal by using a digital pre-distortion algorithm. By using the I/Q imbalance estimation and compensation in conjunction with the DPD algorithm, it is possible to improve the performance of the apparatus according to the first aspect.

According to a second aspect, a method for I/Q imbalance estimation in a communication apparatus is provided. The communication apparatus comprises a TX, a RX, a PLL, and a RF loopback path connecting the TX and the RX.

According to the method, steps (a)-(e) are performed successively at a first phase shift and a different second phase shift between the TX LO signal and the RX LO signal. The method further comprises the steps of:.

By so doing, it is possible to estimate and compensate the combined TX/RX frequency-dependent I/Q imbalance in the communication apparatus without having to use additional external hardware (e.g., such as an envelope detector).

In one embodiment of the second aspect, step (a) comprises generating the reference signal as a pseudo-noise signal or an OFDM signal. By so doing, it is possible to study the combined TX/RX frequency-dependent I/Q imbalance in a wideband of the communication apparatus more efficiently. Furthermore, the OFDM signal is a signal used for regular data transmission in a wireless communication system, thereby meaning that the communication apparatus does not need to generate any special signal for the purpose of the I/Q imbalance estimation and compensation in the time domain.

In one embodiment of the second aspect, steps (f) and (g) are performed by using one of a Least-Mean-Square (LMS) algorithm, a normalized LMS algorithm, a leaky LMS algorithm, and a recursive least squares algorithm. This may make the method according to the second aspect more flexible in use because it may use one of these several algorithms depending on particular application.

In one embodiment of the second aspect, the second phase shift differs from the first phase shift by <NUM> degrees. This preferred phase shift provides the highest accuracy of the I/Q imbalance estimation and compensation.

In one embodiment of the second aspect, the method further comprises the step of compensating the determined contribution made by each of the TX and the RX in the combined TX/RX frequency-dependent I/Q imbalance using a pre-compensating FIR filter in the TX and a post-compensating FIR filter in the RX. Each of the pre-compensating FIR filter and the post-compensating FIR filter has a filter length predefined based on a required accuracy of I/Q imbalance contribution compensation and/or a required amount of computational resources to be used for the I/Q imbalance contribution compensation. By using such FIR filters, the compensation of the contributions made by each of the TX and the RX in the combined TX/RX frequency-dependent I/Q imbalance may be performed more efficiently. Furthermore, by selecting the filter length in this manner, it is possible to provide an acceptable tradeoff between the accuracy of I/Q imbalance contribution compensation and the computational resources used therefor.

In one embodiment of the second aspect, the determined contributions made by the TX and the RX in the combined TX/RX frequency-dependent I/Q imbalance are compensated simultaneously. By so doing, it is possible to reduce the time and number of measurements and calculations which are required for the I/Q imbalance compensation in the communication apparatus.

In one embodiment of the second aspect, the determined contributions made by the TX and the RX in the combined TX/RX frequency-dependent I/Q imbalance are compensated when the communication apparatus operates in a TDD mode. This may make the method according to the second aspect more flexible in use. In particular, the fact that the method according to the second aspect may be used to perform the I/Q imbalance estimation and compensation in the TDD mode means that it may be used for the same purpose in an FDD mode. It should be noted that the opposite is not true, i.e. the possibility of the I/Q imbalance estimation and compensation in the FDD mode does mean the possibility of doing the same in the TDD mode.

In one embodiment of the second aspect, the method further comprises the steps of amplifying the TX RF signal by using a nonlinear power amplifier in the TX, and reducing nonlinear distortions occurring in the amplified TX RF signal by using a DPD algorithm. By using the I/Q imbalance estimation and compensation in conjunction with the DPD algorithm, it is possible to improve the performance of the method according to the second aspect.

According to a third aspect, a computer program product is provided. The computer program product comprises a computer-readable storage medium storing a computer code. When executed by at least one processor of a communication apparatus, the computer code causes the communication apparatus to perform the method according to the second aspect. By using such a computer program product, it is possible to simplify the implementation of the method according to the second aspect in any computing device.

Other features and advantages of the present disclosure will be apparent upon reading the following detailed description and reviewing the accompanying drawings.

The present disclosure is explained below with reference to the accompanying drawings in which:.

Various embodiments of the present disclosure are further described in more detail with reference to the accompanying drawings. However, the present disclosure may be embodied in many other forms and should not be construed as limited to any certain structure or function discussed in the following description. In contrast, these embodiments are provided to make the description of the present disclosure detailed and complete. The present invention is defined by the scope of the appended claims.

According to the detailed description, it will be apparent to the ones skilled in the art that the scope of the present disclosure encompasses any embodiment thereof, which is disclosed herein, irrespective of whether this embodiment is implemented independently or in concert with any other embodiment of the present disclosure. For example, the apparatus and method disclosed herein may be implemented in practice by using any numbers of the embodiments provided herein. Furthermore, it should be understood that any embodiment of the present disclosure may be implemented using one or more of the features presented in the appended claims.

The word "exemplary" is used herein in the meaning of "used as an illustration". Unless otherwise stated, any embodiment described herein as "exemplary" should not be construed as preferable or having an advantage over other embodiments.

As used in the embodiments disclosed herein, a communication apparatus may refer to a transceiving apparatus configured to employ two parallel channels I (in-phase) and Q (quadrature) to perform complex signal processing. For example, the I/Q channels may be used to perform a quadrature modulation/demodulation function. Non-restrictive examples of such a communication apparatus include a direct conversion transceiver, a low intermediate frequency (low-IF) transceiver, etc. Moreover, such a communication apparatus may be implemented as part of a user equipment (UE) or a network node.

The UE may refer to a mobile device, a mobile station, a terminal, a subscriber unit, a mobile phone, a cellular phone, a smart phone, a cordless phone, a personal digital assistant (PDA), a wireless communication device, a desktop computer, a laptop computer, a tablet computer, a gaming device, a netbook, a smartbook, an ultrabook, a medical device or medical equipment, a biometric sensor, a wearable device (for example, a smart watch, smart glasses, a smart wrist band, etc.), an entertainment device (for example, an audio player, a video player, etc.), a vehicular component or sensor, a smart meter/sensor, an unmanned vehicle (e.g., an industrial robot, a quadcopter, etc.), industrial manufacturing equipment, a global positioning system (GPS) device, an Internet-of-Things (loT) device, an Industrial IoT (IIoT) device, a machine-type communication (MTC) device, a group of Massive IoT (MIoT) or Massive MTC (mMTC) devices/sensors, or any other suitable device configured to support wireless communications. In some embodiments, the UE may refer to at least two collocated and inter-connected UEs thus defined.

The network node may relate to a fixed point of communication for the UE in a particular wireless or wired communication network. In case of a wireless communication network, the network node may implemented as a Radio Access Network (RAN) node referred to as a base transceiver station (BTS) in terms of the <NUM> communication technology, a NodeB in terms of the <NUM> communication technology, an evolved NodeB (eNodeB) in terms of the <NUM> communication technology, and a gNB in terms of the <NUM> New Radio (NR) communication technology. The RAN node may serve different cells, such as a macrocell, a microcell, a picocell, a femtocell, and/or other types of cells. The macrocell may cover a relatively large geographic area (for example, at least several kilometers in radius). The microcell may cover a geographic area less than two kilometers in radius, for example. The picocell may cover a relatively small geographic area, such, for example, as offices, shopping malls, train stations, stock exchanges, etc. The femtocell may cover an even smaller geographic area (for example, a home). Correspondingly, the RAN node serving the macrocell may be referred to as a macro node, the RAN node serving the microcell may be referred to as a micro node, and so on.

According to the embodiments disclosed herein, an I/Q imbalance may refer to an interference between an I (in-phase) channel and a Q (quadrature) channel in the communication apparatus, which is caused by gain imbalance between the I channel and the Q channel and by a quadrature error between the I channel and the Q channel. The I/Q imbalance occurs particularly frequently when a high frequency, such as a frequency in the millimeter band, is used for communication, when a broadband signal is handled, or when inexpensive (low-quality) components are employed in the communication apparatus.

Many studies have been focused on improving I/Q imbalance estimation and compensation. However, most of them are devoted to the I/Q imbalance estimation and compensation in a receiver (RX) path, and much less in a transmitter (TX) path. Furthermore, it is usually recommended to use external hardware, such as an envelope detector, to estimate the I/Q imbalance. In case of the frequency-dependent I/Q imbalance, it is additionally recommended to estimate it by using either multi-tones or OFDM pilot signals as a reference signal. In this case, the I/Q imbalance in a frequency domain (at individual frequencies) is estimated. This approach is poorly compatible, for example, with a digital pre-distortion (DPD) algorithm which is performed in a time domain to improve the linearity of power amplifiers.

There are also methods for I/Q imbalance estimation and compensation by using an OFDM signal in a time division duplexing (TDD) mode. However, these existing methods are mainly performed in two steps. There are various ways in which, as a rule, the TX path is estimated and calibrated first. And then, after the I/Q imbalance in the TX path is compensated, the I/Q imbalance in the RX path is estimated and compensated. Moreover, these existing methods also use additional external hardware, such as the envelope detector, which is unavailable for regular data transmission in a wireless communication system.

A very limited number of works is associated with simultaneous TX/RX calibration (in terms of the I/Q imbalance estimation and compensation) according to a loopback transceiver. However, most of them, which involve simultaneously estimating and compensating the TX and RX I/Q imbalance in the loopback transceiver, are related to frequency division duplexing (FDD) communication systems where there is a frequency offset between the TX path and the RX path. Thus, using such methods for TDD communication systems also requires using an additional external hardware, such as an external down-converter.

The exemplary embodiments disclosed herein provide a technical solution that allows mitigating or even eliminating the above-sounded drawbacks peculiar to the prior art. In particular, the technical solution disclosed herein involves using an adaptive filter in a loopback architecture of a communication apparatus to estimate and compensate a combined TX/RX frequency-dependent I/Q imbalance. The combined TX/RX frequency-dependent I/Q imbalance is estimated by using a reference signal and two different phase shifts between a TX local oscillator (LO) signal and a RX LO signal. The resulting two estimates of the combined TX/RX frequency-dependent I/Q imbalance are further used to calculate separately contributions made by a TX and a RX in the combined TX/RX frequency-dependent I/Q imbalance. By so doing, it is possible to estimate the combined TX/RX frequency-dependent I/Q imbalance without having to use additional external hardware. The calculated contributions may be then used to compensate the combined TX/RX frequency-dependent I/Q imbalance itself in a time domain.

<FIG> shows a schematic block diagram of a communication apparatus <NUM> in accordance with one exemplary embodiment. The communication apparatus <NUM> may be implemented as part of a UE or a network node, as discussed above. As shown in <FIG>, the communication apparatus <NUM> comprises a transmitter (TX) <NUM>, a receiver (RX) <NUM>, a phase-locked loop (PLL) <NUM>, and a RF loopback path <NUM>. It is assumed that each of the TX <NUM> and the RX <NUM> comprises an I (in-phase) channel and a Q (quadrature) channel. It is further assumed that there is a combined TX/RX frequency-dependent I/Q imbalance caused, for example, by a digital-to-analog converter (DAC), a low-pass filter (LPF) and an up-converter included in the I and Q channels of the TX <NUM> and by an analog-to-digital converter (ADC), an LPF and a down-converter included in the I and Q channels of the RX <NUM>. As also shown in <FIG>, the TX <NUM> comprises an adaptive filter <NUM> configured to determine contributions <NUM> made by each of the TX <NUM> and the RX <NUM> in the combined TX/RX frequency-dependent I/Q imbalance in the time domain, as will be discussed further in more detail. It should be noted that the RF loopback path <NUM> is used in the communication apparatus <NUM> only for the purpose of estimating and compensating the contributions <NUM>, which means that the RF loopback path <NUM> may be excluded from the communication apparatus <NUM> after the estimating and compensating procedure is completed. Furthermore, the number, arrangement and interconnection of the constructive elements constituting the communication apparatus <NUM>, which are shown in <FIG>, are not intended to be any limitation of the present disclosure, but merely used to provide a general idea of how the constructive elements may be implemented within the communication apparatus <NUM>. It should be also noted that the PLL <NUM>, the RF loopback path <NUM>, the adaptive filter <NUM> and typical different elements of the TX <NUM> and the RX <NUM>, for example, such as the DAC/ADC, the LPF, the up-/down-converters, are well-known in the art, for which reason their detailed description is omitted herein.

<FIG> shows a flowchart of a method <NUM> for I/Q imbalance estimation in accordance with one exemplary embodiment. The method <NUM> as such describes the calibration of the communication apparatus <NUM>, which is aimed at estimating and compensating the contributions <NUM> made by the TX <NUM> and the RX <NUM> in the combined TX/RX frequency-dependent I/Q imbalance. As shown in <FIG>, the method <NUM> starts with a step S202, in which the TX <NUM> generates a reference signal. In some embodiments, the step <NUM> may involve generating the reference signal as a pseudo-noise signal or an Orthogonal Frequency-Division Multiplexing (OFDM) signal. After or in parallel to the step S202, the PLL <NUM> generates a TX LO signal for the TX <NUM> and a RX LO signal for the RX <NUM> in a step S204 of the method <NUM>. Next, the method <NUM> proceeds to a step S206, in which the TX <NUM> obtains a TX RF signal by quadrature mixing of the reference signal with the TX LO signal. Further, the method <NUM> goes on to a step S208, in which the TX RF signal is propagated towards the RX <NUM> through the RF loopback path <NUM>. It should be noted that the TX RF signal is outputted from the RF loopback path <NUM> as a RX RF signal. The RX RF signal is then used in a step S210, in which the RX <NUM> obtains a representation of the reference signal by quadrature mixing the RX RF signal with the RX LO signal. The representation of the reference signal comprises the reference signal itself plus a noise component caused by the combined TX/RX frequency-dependent I/Q imbalance. According to the method <NUM>, the steps S202-S210 are performed successively at a first phase shift and a different second phase shift between the TX LO signal and the RX LO signal. In a preferred embodiment, the second phase shift differs from the first phase shift by <NUM> degrees; this phase difference value the highest accuracy of the I/Q imbalance estimation and compensation in accordance with the method <NUM>.

Once the steps <NUM>-<NUM> are repeated for the two different phase measurements, the method <NUM> goes on. In particular, it proceeds to a step S212, in which the adaptive filter <NUM> obtains a first estimate of the combined TX/RX frequency-dependent I/Q imbalance in the time domain based on the reference signal and the representation of the reference signal in case of the first phase shift between the TX LO signal and the RX LO signal. Then, in a step S214, the adaptive filter <NUM> obtains a second estimate of the combined TX/RX frequency-dependent I/Q imbalance in the time domain based on the reference signal and the representation of the reference signal in case of the second phase shift between the TX LO signal and the RX LO signal. It should be noted that the steps S212 and <NUM> may be implemented by the adaptive filter <NUM> in parallel, if required. Further, in a step S216, in which the adaptive filter <NUM> uses the first and second estimates to determine the contributions <NUM> made by the TX <NUM> and the RX <NUM> in the combined TX/RX frequency-dependent I/Q imbalance in the time domain. The method <NUM> may end up in a step S218, in which the determined contributions are compensated. By using the method <NUM>, it is possible to estimate and compensate the combined TX/RX frequency-dependent I/Q imbalance in the communication apparatus <NUM> without having to use additional external hardware (e.g., such as the envelope detector).

In one exemplary embodiment, the steps S212 and <NUM> of the method <NUM> are performed by using one of a Least-Mean-Square (LMS) algorithm, a normalized LMS algorithm, a leaky LMS algorithm, and a recursive least squares algorithm. Each of these several algorithms may be used depending on particular application.

In one exemplary embodiment, the step <NUM> of the method <NUM> is performed by using a pre-compensating FIR filter in the TX <NUM> and a post-compensating FIR filter in the RX <NUM>. In this case, the term "pre-compensating" means that the corresponding filter is located before an I/Q imbalance source (e.g., before the DAC and the up-converter) in the TX <NUM>, while the term "post-compensating" means that the corresponding filter is located after an I/Q imbalance source (e.g., after the down-converter and the ADC) in the RX <NUM>. Each of the pre-compensating FIR filter and the post-compensating FIR filter has its own filter length predefined based on a required accuracy of I/Q imbalance contribution compensation and/or a required amount of computational resources to be used for the I/Q imbalance contribution compensation. By using such FIR filters, the compensation of the contributions made by each of the TX <NUM> and the RX <NUM> in the combined TX/RX frequency-dependent I/Q imbalance may be performed more efficiently. Furthermore, by selecting the filter length in this manner, it is possible to provide an acceptable tradeoff between the accuracy of I/Q imbalance contribution compensation and the computational resources used therefor.

In one exemplary embodiment, the determined contributions <NUM> made by the TX <NUM> and the RX <NUM> in the combined TX/RX frequency-dependent I/Q imbalance are compensated simultaneously in the step S218. By so doing, it is possible to reduce the time and number of measurements and calculations required for the I/Q imbalance compensation in the communication apparatus <NUM>.

In one exemplary embodiment, the determined contributions <NUM> made by the TX <NUM> and the RX <NUM> in the combined TX/RX frequency-dependent I/Q imbalance are compensated in the step S218 when the communication apparatus <NUM> operates in a TDD mode. The possibility of using the method <NUM> in the TDD mode means that it may be also used in an FDD mode. It should be noted that the opposite is not true, i.e. the possibility of the I/Q imbalance estimation and compensation in the FDD mode does not mean the possibility of doing the same in the TDD mode. Therefore, the method <NUM> is more flexible in use compared to the existing methods for the I/Q imbalance estimation and compensation.

In one exemplary embodiment, when the communication apparatus <NUM> further comprises a nonlinear power amplifier in the TX <NUM> and a digital signal processing (DSP) unit, the method <NUM> comprises the following additional steps, in which the TX RF signal obtained in the step S206 is amplified by using the nonlinear power amplifier and nonlinear distortions occurring in the amplified TX RF signal are reduced by using a DPD algorithm executed by the DSP unit. By using the I/Q imbalance estimation and compensation in conjunction with the DPD algorithm, it is possible to improve the performance of the method <NUM>. It should be also noted that the DSP unit may be implemented by using a CPU, general-purpose processor, single-purpose processor, microcontroller, microprocessor, application specific integrated circuit (ASIC), field programmable gate array (FPGA), digital signal processor (DSP), complex programmable logic device, etc. In some embodiments, the DSP unit may be implemented as any combination of the aforesaid, e.g., as two or more microprocessors.

Let us now consider how to calculate the contributions <NUM> made by the TX <NUM> and the RX <NUM> in the combined TX/RX frequency-dependent I/Q imbalance. For this purpose, let us assume that the adaptive filter <NUM> uses the LMS algorithm, and the DSP unit uses the DPD algorithm. The first estimate of the combined TX/RX frequency-dependent I/Q imbalance is performed with the first phase shift between the TX LO signal and the RX LO signal, for example, equal to zero. Let x be the reference signal, then an I/Q imbalance βtx in the TX <NUM> is defined by: <MAT> where x* is the conjugate of x. An estimate β<NUM> of the combined TX/RX frequency-dependent I/Q imbalance, which is obtained by the adaptive filter <NUM> at the first phase shift, is equal to: <MAT>.

Then, the TX RF signal after the I/Q imbalance source in the TX <NUM> is obtained as: <MAT>.

The output signal of the RF loopback path <NUM>, i.e. the RX RF signal, is obtained as: <MAT> where βrx is the I/Q imbalance in the RX <NUM>. Therefore, the following expression should be executed in the I/Q imbalance compensation case: <MAT>.

Then, the PLL <NUM> sets the second phase shift between the TX LO signal and the RX LO signal. In this case, the TX RF signal after the I/Q imbalance source in the TX <NUM> is obtained as: <MAT> where Φ = |Φ| · ej·ϕ is the transfer function of an RF phase shifter. As noted earlier, the PLL <NUM> sets the phase shift between the TX LO and RX LO signals, and this operation of the PLL <NUM> is equivalent to the operation of some RF phase shifter with the transfer function Φ.

Hence, the RX RF signal outputted from the RF loopback path <NUM> is obtained as: <MAT>.

The output signal y should be aligned with the reference signal as follows: <MAT> where G = Φ-<NUM>; <MAT>; Φ* = |Φ| · e-j·ϕ.

Therefore, the following expression should be executed in the I/Q imbalance compensation case at the second phase shift: <MAT>.

As a result, one can obtain the following system equation: <MAT>.

The results of these two measurements (i.e. at the first and second phase shifts between the TX LO signal and the RX LO signal) make it possible to calculate separately the contributions <NUM> made by the TX <NUM> and RX <NUM> in the combined TX/RX frequency-dependent I/Q imbalance, namely: <MAT>.

The following notes should be taken into consideration. Firstly, as noted earlier, the best calculation accuracy is ensured when the difference between the first and second phase shifts is <NUM> degrees. Secondly, in the presence of the nonlinear power amplifier, the level of I/Q imbalance compensation is limited by the level of nonlinear products after the DPD algorithm. On the other hand, the presence of the I/Q imbalance leads to additional nonlinear distortion products (due to the occurrence of complex conjugate components in the signal spectrum). Therefore, the simultaneous operation of the DPD algorithm and the compensation of I/Q imbalance is correct.

<FIG> shows topologies of the TX <NUM> and the RX <NUM> included in the communication apparatus <NUM> in accordance with one exemplary embodiment. As shown in <FIG>, the TX <NUM> of the communication apparatus <NUM> comprises a DAC <NUM>, an up-converter <NUM>, a nonlinear power amplifier <NUM>, a switch <NUM>, a pre-compensating FIR filter <NUM>, an adder <NUM>, an antenna <NUM>, and the adaptive filter <NUM>. The RX <NUM> of the communication apparatus <NUM> comprises a down-converter <NUM>, an ADC <NUM>, a switch <NUM>, and a post-compensating FIR filter <NUM>. The I/Q imbalance sources in the TX <NUM> and the RX <NUM> are schematically shown in <FIG> as "TX IQimb" and "RX IQimb", respectively. Moreover, as follows from <FIG>, the communication apparatus <NUM> also comprises a DSP unit <NUM> configured to perform the DPD algorithm. The adaptive filter <NUM> is implemented as an adaptive FIR (AFIR) filter configured to use the above-obtained mathematical expressions for βtx, βrx in order to compensate the contributions <NUM> made by the TX <NUM> and the RX <NUM> in the combined TX/RX frequency-dependent I/Q imbalance.

Accordingto <FIG>, the communication apparatus <NUM> operates as follows. When it is required to estimate the combined TX/RX frequency-dependent I/Q imbalance (i.e. perform the steps S212-S216), the switch <NUM> is off, so that the AFIR filter <NUM> is used. When it is required to compensate the determined contributions <NUM> made by the TX <NUM> and the RX <NUM> in the combined TX/RX frequency-dependent I/Q imbalance (i.e. perform the step S218), the switch <NUM> is on, so that the pre-compensating FIR filter <NUM> is used. It should be noted that the "on" position of the switch <NUM> corresponds to the situation when the communication apparatus <NUM> is used for regular data transmission via the antenna <NUM>. Irrespective of the position of the switch <NUM>, the reference signal x(t) goes to the adder <NUM> where it is added with a predistortion signal pd(t) from the DSP unit <NUM> to reduce the nonlinearity distortions caused by the amplifier <NUM>. The reference signal then goes from the adder <NUM> though the DAC <NUM> to the up-converter <NUM>. The up-converter <NUM> performs the quadrature mixing of the reference signal with the TX LO signal generated by the PLL <NUM>, thereby obtaining the TX RF signal. Further, the TX RF signal goes through the RF loopback path <NUM> to the RX <NUM>.

More specifically, the RF loopback path <NUM> outputs the TX RF signal as the RX RF signal to the down-converter <NUM> in the RX <NUM>. The down-converter <NUM> performs the quadrature mixing of the RX RF signal with the RX LO signal generated by the PLL <NUM>, thereby obtaining the representation of the reference signal x(t), i.e. the reference signal x(t) with some signal errors caused by the combined TX/RX frequency-dependent I/Q imbalance. The representation of the reference signal then goes to the ADC <NUM>. When the combined TX/RX frequency-dependent I/Q imbalance is estimated (i.e. the switch <NUM> is off), the switch <NUM> is off, so that an output signal of the ADC <NUM>, which is denoted as y_fb(t) in <FIG>, is aligned to the reference signal x(t) (multiplied by a complex coefficient, which minimizes the squared error between the input samples of the reference signal x(t) and the samples of the output signal y_fb(t)). A specified error signal e(t) is used in the DSP unit <NUM> to update a predistortion function by using the LMS or LS algorithm. The specified error signal e(t) is also used to adapt coefficients of the AFIR filter <NUM> in the TX <NUM>.

As noted earlier, the combined TX/RX frequency-dependent I/Q imbalance should be estimated twice for different phase shifts. The "phase shifter" is implemented by the phase shift between the TX LO signal and the RX LO signal. By using two AFIR impulse responses for the two phase shifts, the impulse responses of the pre-compensating FIR filter <NUM> in the TX <NUM> and the post-compensating FIR filter <NUM> in the RX <NUM> are calculated. Then, the contributions <NUM> made by the TX <NUM> and the RX <NUM> in the combined TX/RX frequency-dependent I/Q imbalance are compensated simultaneously, for which purpose the switches <NUM> and <NUM> should be both in the "on" position.

<FIG> shows a flowchart <NUM> explaining how the I/Q imbalance estimation and compensation may be used in concert with the DPD algorithm in the method <NUM> in accordance with one exemplary embodiment. The flowchart <NUM> starts with a step S402, in which the DPD algorithm is trained based on the reference signal to find an acceptable predistortion signal pd(t). Then, the flowchart <NUM> goes to a step S404, in which the combined TX/RX frequency-dependent I/Q imbalance is estimated twice at the first and second phase shifts, as discussed above (i.e. the steps S202-S216 are performed), and the resulting two estimates of the combined TX/RX frequency-dependent I/Q imbalance are used to calculate the impulse responses of the pre-compensating FIR filter <NUM> and the post-compensating FIR filter <NUM>. After that, the flowchart <NUM> proceeds to a step S406, in which the DPD algorithm and the I/Q imbalance compensation (i.e. the step <NUM> performed based on the calculated impulse responses of the FIR filters <NUM>, <NUM>) are used together.

<FIG> shows dependences of an error value magnitude (EVM) on an average output power of the nonlinear power amplifier <NUM> included in the TX <NUM>, which are obtained on a test platform by using the method <NUM> in view of the flowchart <NUM>. In other words, the dependences show the results of the joint application of the I/Q imbalance estimation and compensation and the digital predistortion of the power amplifier nonlinearity. In <FIG>, "IQ on/off" means that the method <NUM> is used/not used, and "DPD on/off" means that the DPD algorithm is used/not used. As can be seen, the best results, i.e. the largest reduction of the EVM, are achieved when the method <NUM> is used together with the DPD algorithm (see the dependence corresponding to "IQ on, DPD on" in <FIG>). At the same time, the method <NUM>, as used without the DPD algorithm, also allows significantly reducing the EVM.

<FIG> and <FIG> show a power spectral density of a residual I/Q imbalance signal at the output of the TX <NUM> and the RX <NUM>, respectively, which is resulted from the use of the method <NUM> without considering the nonlinearity of the power amplifier <NUM>. To distinguish an I/Q imbalance signal caused by the I/Q imbalance source(s) provided in each of the TX <NUM> and the RX <NUM> from the reference signal represented by an OFDM signal, the reference signal is shifted in frequency to the region of positive frequencies. The I/Q imbalance signal is presented as a weighted complex conjugate of the reference signal. In this case, the complexconjugate signal is located in the region of negative frequencies. Thus, the reference signal (shown by using a gray solid line) and the I/Q imbalance signal (shown by using a black dashed line) are located in different frequency ranges. The residual I/Q imbalance signal obtained by using the method <NUM> is shown in <FIG> and <FIG> by using a black solid line. As can be seen, the method <NUM> almost eliminates the I/Q imbalance signal (see its normalized mean-square error (NMSE) values for cases with and without the I/Q imbalance compensation performed by using the method <NUM>). It should be also noted that such estimates (i.e. NMSE values) cannot be obtained in the presence of the nonlinear power amplifier <NUM>, since the nonlinear distortions caused by the power amplifier <NUM> do not allow the residual I/Q imbalance signal to be seen against their background (i.e. the error in the reference signal will be mostly determined by these nonlinear distortions).

It should be noted that one or more steps or operations of the method <NUM> and the flowchart <NUM> may be implemented by various means, such as hardware, firmware, and/or software. As an example, one or more of the steps or operations described above may be embodied by a computer code, processor executable instructions, data structures, program modules, and other suitable data representations. Furthermore, the computer code which embodies the step(s) or operation(s) described above may be stored on a corresponding data carrier and executed by at least one processor of a computing device. This data carrier may be implemented as any computer-readable storage medium configured to be readable by said at least one processor to execute the computer code. Such computer-readable storage media may include both volatile and nonvolatile media, removable and non-removable media. By way of example, and not limitation, the computer-readable media comprise media implemented in any method or technology suitable for storing information. In more detail, the practical examples of the computer-readable media include, but are not limited to information-delivery media, RAM, ROM, EEPROM, flash memory or other memory technology, CD-ROM, digital versatile discs (DVD), holographic media or other optical disc storage, magnetic tape, magnetic cassettes, magnetic disk storage, and other magnetic storage devices.

Claim 1:
A communication apparatus (<NUM>) comprising:
a transmitter,TX (<NUM>);
a receiver, RX (<NUM>); and
a phase-locked loop (<NUM>) configured to:
generate a TX local oscillator, LO, signal and a RX LO signal, and
successively provide a first phase shift and a different second phase shift between the TX LO signal and the RX LO signal,
wherein the TX is configured to:
generate a reference signal, and
obtain a TX radio frequency, RF, signal by quadrature mixing of the reference signal with the TX LO signal,
wherein the communication apparatus (<NUM>) further comprises a RF loopback path (<NUM>) configured to output the TX RF signal to the RX as a RX RF signal,
wherein the RX is configured to obtain a representation of the reference signal by quadrature mixing of the RX RF signal with the RX LO signal, and
wherein the TX (<NUM>) comprises an adaptive filter (<NUM>) configured to:
based on the reference signal and the representation of the reference signal in case of the first phase shift between the TX LO signal and the RX LO signal, obtain a first estimate of a combined TX/RX frequency-dependent In-phase and Quadrature, I/Q, imbalance in a time domain,
based on the reference signal and the representation of the reference signal in case of the second phase shift between the TX LO signal and the RX LO signal, obtain a second estimate of the combined TX/RX frequency-dependent I/Q imbalance in the time domain, and
based on the first and second estimates, determine a contribution made by each of the TX and the RX in the combined TX/RX frequency-dependent I/Q imbalance in the time domain.