Patent Description:
Ultra-Wideband (UWB) technology is a wireless technology for the transmission of large amounts of digital data as modulated coded impulses over a very wide frequency spectrum with very low power over a short distance. Such pulse based transmission being an alternative to transmitting using a sinusoidal wave which is then turned on or off, to represent the digital states, as employed within today's wireless communication standards and systems such as IEEE <NUM> (Wi-Fi), IEEE <NUM> wireless personal area networks (PANs), IEEE <NUM> (WiMAX), Universal Mobile Telecommunications System (UMTS), Global System for Mobile Communications (GSM), General Packet Radio Service (GPRS), and those accessing the Industrial, Scientific and Medical (ISM) bands, and International Mobile Telecommunications-<NUM> (IMT-<NUM>).

UWB systems are well-suited to short-distance applications in a variety of environments, such as depicted in <FIG> including peripheral and device interconnections, as exemplified by first residential environment <NUM>, sensor networks, as exemplified by second residential environment <NUM>, control and communications, as exemplified by industrial environment <NUM>, medical systems, as exemplified by medical imaging <NUM>, and personal area networks (PAN), as exemplified by PAN <NUM>. Due to low emission levels permitted by regulatory agencies such UWB systems tend to be short-range indoor applications but it would be evident that a variety of other applications may be considered where such regulatory restrictions are relaxed and / or not present addressing military and civilian requirements for communications between individuals, electronic devices, control centers, and electronic systems for example.

Accordingly, it would be beneficial for UWB transmitters, UWB receivers and UWB transceivers to exploit multiple directive antennas oriented in different directions to ensure spatial filtering of undesired signals and increase signal strength. It would be further beneficial for the multiple directive antennas for incorporate integrated baluns.

It would be further beneficial for UWB transmitters, UWB receivers and UWB transceivers to exploit dynamic configuration of the multi-pulse bundles employed to transmit the bits / symbols within the packets. Such dynamic configuration being to enhance link quality of service.

It would be further beneficial for UWB transmitters, UWB receivers and UWB transceivers to exploit dynamic configuration of the band or bands which the transmitter operates upon.

It would be further beneficial for UWB transmitters, UWB receivers and UWB transceivers to exploit antenna sub-systems which provide an omnidirectional radiation pattern with printed circuit board implementations offering filtering and balun functions offering small footprints and low cost.

<CIT> discloses that Ultra-Wideband (UWB) technology exploits modulated coded impulses over a wide frequency spectrum with very low power over a short distance for digital data transmission, and such UWB systems through their receivers may operate in the presence of interfering signals and should provide for robust communications. Accordingly, an accurate and sharp filter that operates at low power is required and beneficially one that does not require a highly accurate power heavy clock. Further, many UWB applications require location and / or range finding of other elements and it would therefore be beneficial to provide a UWB based range finding and/or location capability removing the requirement to add additional device complexity and, typically significant, power consumption. <CIT> discloses that today's leading edge modulated sinusoidal wave wireless communication standards and systems achieve power efficiencies of 50nJ / bit employing narrowband signaling schemes and traditional RF transceiver architectures. However, such designs severely limit the achievable energy efficiency, especially at lower data rates such as below <NUM> Mbps. Further, it is important that peak power consumption is supportable by common battery or energy harvesting technologies and long term power consumption neither leads to limited battery lifetimes or an inability for alternate energy sources to sustain them. Accordingly, it would be beneficial for next generation applications to exploit inventive transceiver structures and communication schemes in order to achieve the sub nJ per bit energy efficiencies required by next generation applications.

Other aspects and features of the present invention will become apparent to those ordinarily skilled in the art upon review of the following description of specific embodiments of the invention in conjunction with the accompanying figures.

It is an object of the present invention to mitigate limitations within the prior art relating to ultra-wideband wireless communication systems and more particularly to configuring ultra-wideband transmitters and receivers for enhanced ultra-wideband wireless link performance and antennas for said communication systems.

Examples disclosed herein of background relevance include an antenna comprising:.

In accordance with an embodiment of the invention there is provided a transmitter for an impulse radio according to appended claim <NUM>. The transmitter comprises:.

Examples disclosed herein of background relevance include an antenna sub-system comprising:.

Embodiments and examples of background relevance will now be described, by way of example only, with reference to the attached Figures, wherein:.

The present invention is directed to ultra-wideband wireless communication systems and more particularly to configuring ultra-wideband transmitters and receivers for enhanced ultra-wideband wireless link performance and antennas for said communication systems.

The ensuing description provides exemplary embodiment(s) only, and is not intended to limit the scope, applicability or configuration of the disclosure. Rather, the ensuing description of the exemplary embodiment(s) will provide those skilled in the art with an enabling description for implementing an exemplary embodiment. It being understood that various changes may be made in the function and arrangement of elements without departing from the scope as set forth in the appended claims.

As discussed supra UWB offers many potential advantages such as high data rate, low-cost implementation, and low transmit power, ranging, multipath immunity, and low interference. The Federal Communications Commission (FCC) regulations for UWB reserved the unlicensed frequency band between <NUM> and <NUM> for indoor UWB wireless communication system wherein the low regulated transmitted power allows such UWB systems to coexist with other licensed and unlicensed narrowband systems. Therefore, the limited resources of spectrum can be used more efficiently. On the other hand, with its ultra-wide bandwidth, an UWB system has a capacity much higher than the current narrowband systems for short range applications. Two possible techniques for implementing UWB communications are Impulse Radio (IR) UWB and multi-carrier or multi-band (MB) UWB. IR-UWB exploits the transmission of ultra-short (of the order of nanosecond) pulses, although in some instances in order to increase the processing gain more than one pulse represents a symbol. In contrast MB-UWB systems use orthogonal frequency division multiplexing (OFDM) techniques to transmit the information on each of the sub-bands. Whilst OFDM has several good properties, including high spectral efficiency, robustness to RF and multi-path interferences. However, it has several drawbacks such as up and down conversion, requiring mixers and their associated high power consumption, and is very sensitive to inaccuracies in frequency, clock, and phase. Similarly, nonlinear amplification destroys the orthogonality of OFDM. Accordingly, MB-UWB is not suitable for low-power and low cost applications.

In contrast IR-UWB offers several advantages, including unlicensed usage of several gigahertz of spectrum, offers great flexibility of spectrum usage, and adaptive transceiver designs can be used for optimizing system performance as a function of the data rate, operation range, available power, demanded quality of service, and user preference. Further, multi-Gb/s data-rate transmission over very short range is possible and due to the ultra-short pulses within IR-UWB it is very robust against multipath interference, and more multipath components can be resolved at the receiver in some implementations, resulting in higher performance. Further, the ultra-short pulses support sub-centimeter ranging whilst the lack of up and down conversion allows for reduced implementation costs and lower power transceiver implementations. Beneficially, ultra-short pulses and low power transmissions make IR-UWB communications hard to eavesdrop upon.

An IR-UWB transmitter as described below in respect of embodiments of the invention in with reference to <FIG> and <FIG> respectively exploits an on-demand oscillator following a pulse generator in order to up-convert the pulses from the pulse generated whilst avoiding the requirement of a separate mixer. Implementable in standard CMOS logic both the pulse generator and the on-demand oscillator are digitally tunable in order to provide control over the pulse bandwidth and center frequency. Further, by exploiting a digitally controlled ring oscillator for the on-demand oscillator the IR-UWB transmitter is designed to allow very quick frequency adjustments on the order of the pulse repetition rate (PRR). Beneficially this technique provides the same advantages as MB-OFDM in respect of spectrum configurability, achieved by sequentially changing the transmitted spectrum using a frequency hopping scheme, whilst maintaining the benefits of IR-UWB. Further, by providing advanced duty cycling with fast power up time combined with On-Off Shift Keying (OOK) modulation the IR-UWB according to embodiments of the invention allows significant reductions in power consumption by exploiting the low duty cycle of a UWB symbol and the fact that only half the symbols require sending energy.

In addition to defining the operating frequency range for UWB systems the different regulatory bodies all specify and enforce a specific power spectral density (PSD) mask for UWB communications. A PSD mask as may be employed in respect of embodiments of the invention is the FCC mask for which mask data are summarized in Table <NUM> below for the <NUM>-<NUM> (<NUM>-<NUM>) range.

Accordingly, it would be evident that the upper limit of -<NUM> dB/MHz across the <NUM>-<NUM> frequency range is the same limit imposed on unintentional radiation for a given frequency in order not to interfere with other radios. Basically, for a given frequency, the UWB radio operates under the allowed noise level which creates the relationship presented in Equation (<NUM>) between Ep, the transmitted energy per pulse, the maximum spectral power S , the bandwidth B , the bit rate Rb and the number of pulses per bits Nppb.

The IEEE has published a few standards for a physical layer (PHY) for UWB radio in Personal Area Networks (IEEE <NUM>. 4a-<NUM>), Body Area Networks (IEEE <NUM>. 4a-<NUM>) and Radio-Frequency Identification (IEEE <NUM>. These standards use mostly relatively large pulses resulting in relatively narrow bandwidth which is up-converted to a specific center frequency in order to fill predetermined channels. The data is encoded using pulse-position-modulation (PPM) and bi-phasic shift keying (BPSK) is used to encode redundancy data. Every bit consists of one or more pulses scrambled in phase depending on the target data rate. These standards allow considerable flexibility on channel availability and data rates. The standard also defines the preamble, headers for the data packet and ranging protocol.

These IEEE standards are designed with multiple users in mind and use different channels to transmit the data, thereby putting a heavy constraint on pulse bandwidth and limiting the transmitted energy. Prior art on non-standard transmitter attempts to make better use of the available spectrum by using narrow pulses, which therefore have a larger bandwidth thereby increasing the maximum transmitted energy according to Equation (<NUM>). Accordingly, these transmitters are non-standard and were also designed for different data rates, frequencies, pulse width, etc. Additionally, they also used various encoding schemes, most notably PPM, OOK or BPSK.

Within the work described below the inventors have established improvements with respect to UWB systems, UWB transmitters and energy based UWB receivers which are capable of generating and adapting to a variety of IR-UWB pulses and bit encoding schemes thereby supporting communications from both IR-UWB transmitters compliant to IEEE standards as well as those that are non-standard. These improvements are made with respect to UWB transmitters, UWB receivers, UWB transceivers and UWB systems such as those described and depicted by the inventors within <CIT> "Energy Efficient Ultra-Wideband Impulse Radio Systems and Methods" (<CIT>), <CIT> "Systems and Methods for Spectrally Efficient and Energy Efficient Ultra-Wideband Impulse Radios with Scalable Data Rates" (<CIT>), and <CIT> "Systems and Methods Relating to Ultra Wideband Broadcasting comprising Dynamic Frequency and Bandwidth Hopping" (<CIT>).

Referring to <FIG> there is depicted schematically an exemplary architecture for an IR-UWB transmitter <NUM> according to embodiments of the invention which is composed of five main blocks plus the antenna. First a programmable impulse is produced by a pulse generator <NUM> at clocked intervals when the data signal from AND gate <NUM> is high based upon control signals presented to the AND gate <NUM>. The pulses from the pulse generator <NUM> are then up-converted with a programmable multi-loop digitally controlled ring oscillator (DCRO) <NUM>. The output from the DCRO <NUM> is then coupled to a variable gain amplifier (VGA) <NUM> in order to compensate for any frequency dependency of the pulse amplitude. Finally, a driver <NUM> feeds the antenna <NUM>, overcoming typical package parasitics, such as arising from packaging the transceiver within a quad-flat no-leads (QFN) package. In order to further reduce the power consumption of the IR-UWB transmitter (IR-UWB-Tx) <NUM> according to embodiments of the invention a power cycling controller <NUM> dynamically switches on or off these functional blocks when the data signal is low.

Now referring to <FIG> there is depicted schematically a block diagram <NUM> of an exemplary IR-UWB transmitter according to embodiments of the invention supporting biphasic phase scrambling. In comparison to the IR-UWB transmitter <NUM> in <FIG> for an IR-UWB according to embodiments of the invention without biphasic phase shifting rather than being composed of five main blocks plus the antenna the Biphasic Phase Shifting IR-UWB (BPS-IR-UWB) transmitter comprises <NUM> main blocks. First a programmable impulse is produced by a pulse generator <NUM> at clocked intervals when the data signal from AND gate <NUM> is high based upon control signals presented to the AND gate <NUM>. The pulses from the pulse generator <NUM> are then up-converted with a programmable multi-loop digitally controlled ring oscillator (DCRO) <NUM>. The output from the DCRO <NUM> is then coupled to a dual-output amplifier (VGA) <NUM> both in order to compensate for any frequency dependency of the pulse amplitude but also to generate dual phase shifted output signals that are coupled to a switch <NUM> which selects one of the two signals to couple to the output power amplifier (driver) <NUM> under the action of the switch control signal "S" applied to the switch <NUM>. Note that a similar phase selection scheme could be implemented by affecting the startup conditions for DCRO <NUM> in order to provide the two phases. This would preclude the need for switch <NUM> at the cost of an added control startup condition control signal on DCRO <NUM>.

The output power amplifier <NUM> feeds the antenna <NUM>, overcoming typical package parasitics, such as arising from packaging the transceiver within a quad-flat no-leads (QFN) package. In order to reduce the power consumption of the BPS-IR-UWB transmitter represented by block diagram <NUM> according to an embodiment of the invention a power cycling controller <NUM> dynamically switches on or off these functional blocks when the data signal "PC" is low. Accordingly, a BPS-IR-UWB transmitter according to embodiments of the invention transmits pulses with or without phase shift based upon the control signal "S" applied to switch <NUM>. If this control signal is now fed from a random data generator or a pseudo-random data generator then the resulting pulses coupled to the antenna of the BPS-IR-UWB transmitter will be pseudo-randomly or randomly phase shifted.

Now referring to <FIG> there is depicted schematically a block diagram <NUM> of an exemplary IR-UWB transmitter according to embodiments of the invention. As depicted a Pulse Pattern block <NUM> holds a configuration for the pulses used to represent the current symbol. From the symbol-rate clock (i.e. <NUM>), multiple phases are generated by a Delay Locked Loop (DLL) <NUM>. The rising edge of each clock phase represents the start of one pulse in the symbol pulse bundle. A multiplexer <NUM> is triggered by the edges of the clock phases and selects the configuration of the current pulse out of the Pulse Pattern block <NUM>. A pulse generator (Pulser) <NUM> generates pulses with a pulse width set by the multiplexer <NUM> and enables the Digitally Controlled Oscillator (DCO) <NUM> and Power Amplifier (PA) <NUM>. When enabled, the DCO <NUM> generates a Gaussian shaped pulse with frequency set by the multiplexer <NUM>, which is then amplified by the PA <NUM> and radiated by the antenna <NUM>.

Accordingly, the Pulse Pattern block <NUM> establishes the pulses for a symbol or sequence of symbols. In this manner updating the Pulse Pattern block <NUM> adjusts the pulse sequence employed for each symbol and accordingly the Pulse Pattern block <NUM> may be dynamically updated based upon one or more factors including, but not limited to, network environment data, predetermined sequence, date, time, geographic location, signal-to-noise ratio (SNR) of received signals, and regulatory mask.

Referring to <FIG> there is depicted schematically a multi-pulse symbol UWB protocol according to an embodiment of the invention. Referring to first image 3100A there is depicted a bit <NUM> comprising a series of sub-pulses 3160A to 3160C which are each at frequencies f<NUM>;f<NUM>;f<NUM>. Accordingly, the multi-pulse spectrum <NUM> of a symbol (bit <NUM>) is depicted in second image 3100B as obtained conceptually (phase scrambling is omitted for clarity) by summing the individual pulse spectra of the sub-pulses 3160A to 3160C, which increases the bandwidth whilst increasing the total symbol duration, in contrast with single-pulse prior art methods, whilst maintaining the maximum power below the UWB mask <NUM>. This allows the symbol energy to be maximized while relaxing the timing requirements and level of synchronization required at the receiver. An arbitrary number of pulses with different sets of parameters may be included within a bundle to tailor the pulse spectrum to a given requirement.

Now referring to <FIG> there are depicted the measured pulse shapes for three different frequency setting of a UWB transmitter according to an embodiment of the invention such as described and depicted in <FIG> without biphasic phase scrambling according to <CIT>. First to third traces 3200A to 3200C respectively representing single pulses at <NUM>, <NUM>, and <NUM> respectively.

Referring to <FIG> there is depicted in first trace 3300A a pulse bundle representing a bit being transmitted such as described within <CIT> is depicted whilst second trace 3300B depicts the resulting power spectrum density (PSD). It is evident that this allows for managing the PSD of the final signal through the parameters for each pulse within the frequency hopping sequence. The pulse sequence depicted comprising <NUM> pulses at <NUM>, <NUM> pulses at <NUM>, and <NUM> pulses at <NUM>. The resulting PSD fills the spectrum at around - 58dBm over the entire band.

Now referring to <FIG> there are depicted the power spectrum and pulse train for a pulse bundle according to <CIT> and as depicted in <FIG> supporting operating over a frequency range from approximately <NUM> to approximately <NUM>. First and second images 3400A and 3400B respectively representing the power spectrum and pulse sequence wherein there is no random frequency or phase scrambling during the generation and transmission. Third and fourth images 3400C and 3400D depict the results for random frequency and random phase scrambling of the pulses wherein phase is set per pulse through data established by a pseudo-random data generator. Accordingly, it would be evident that when comparing first and third images 3400A and 3400C that the introduction of random frequency and random phase shifting reduces the spectral lines significantly within the emitted spectrum of a UWB transmitter according to embodiments of the invention.

Referring to <FIG> there is depicted schematically the architecture of an IR-UWB receiver <NUM>. Accordingly, the signal from an IR-UWB transmitter is received via an antenna <NUM> and coupled to a low noise amplifier (LNA) <NUM> followed by first amplifier <NUM> wherein the resulting signal is squared by squaring circuit <NUM> in order to evaluate the amount of energy in the signal. The output of the squaring circuit <NUM> is then amplified with second amplifier <NUM>, integrated with integration circuit <NUM> and evaluated by a flash ADC <NUM> to generate the output signals. Also depicted is Power Cycling Controller <NUM> which, in a similar manner to the power cycling controller <NUM> of IR-UWB transmitter <NUM> in <FIG>, dynamically powers up and down the LNA <NUM>, first and second amplifiers <NUM> and <NUM> respectively, squaring circuit <NUM>, and flash ADC <NUM> to further reduce power consumption in dependence of the circuit's requirements.

Referring to <FIG> there is depicted a schematic of a receiver <NUM> The RF signal from the antenna <NUM> is initially amplified by a Low Noise Amplifier (LNA) <NUM> before being passed to a two stage RF amplifier (AMP1) <NUM>. A first squaring mixer (MIX1) <NUM> multiplies the signal with itself to convert to the Intermediate Frequency (IF). A three-stage Variable Gain Amplifier (VGA) <NUM> amplifies the signal further and implements a bandpass filter function. The VGA <NUM> output is then coupled to a second squaring mixer (MIX2) <NUM> which down-converts the signal to the baseband frequency. A parallel integrator (INT1 and INT2) sums the signal energy, which is digitized by the Analog-to-Digital Converters (ADC1 and ADC2) and sent to a digital processor (not depicted for clarity).

As described within <CIT> and <CIT> the inventors have established design parameters of millisecond range start-up time from sleep mode and microsecond range start-up time from idle mode by establishing a custom integrated DC/DC converter and duty cycled transceiver circuitry that enables fast circuit start-up / shut-down for optimal power consumption under low (<NUM> kbps ) and moderate data rates (<NUM> Mbps ).

In order to sustain good energy efficiency, the elements of a total UWB transceiver, such as depicted with transceiver <NUM> in <FIG>, has been designed for low static sleep current and fast startup/sleep times. Referring to <FIG>, a battery (<NUM> V ≤ VBATT ≤ <NUM> V ) (not depicted for clarity) powers a low-frequency crystal oscillator <NUM>, sleep counter <NUM> and bandgap reference <NUM>, all of which are typically always operational although the bandgap reference <NUM> could be duty cycled ). Their power consumption limits the minimum power consumption of the system to sub-microwatt level. An integrated buck DC-DC converter <NUM> is powered by the battery when the system is not in sleep mode, and this provides the supply voltage to the rest of the system with high conversion efficiency. The startup time of the DC-DC converter <NUM> is on the order of several symbol periods in order to minimize wasted energy. Between sleep periods, the PLL <NUM> is active to provide the base clock for the system. The receiver <NUM> and DLL <NUM> have dedicated power down controls and are only activated during frame transmission / reception. Further, the transmitter is also power cycled through its all-digital architecture which is not depicted as having a separate control. The power consumption of the digital synthesized blocks is low due to the low base clock (e.g. <NUM>).

In principle, a power-cycled transceiver achieves linear scaling of power consumption with data rate, thus achieving constant energy efficiency. With a fixed frame size, multiple data rates are obtained by adjusting the length of the sleep period, with the maximum attainable data rate determined by the symbol rate in the frame itself. In order to preserve energy efficiency, the power consumption during sleep must be lower than the average power consumption. For high data rates, powering down the PLL is not required when its consumption does not significantly degrade the overall efficiency. For low data rates, the whole system except the bandgap reference, crystal oscillator, and sleep counter can be shut down during sleep mode. In this case, the millisecond range startup time of the PLL can be insignificant compared to the sleep period, and overall efficiency is also not significantly degraded.

As depicted the UWB transceiver <NUM> also comprises a receive / transmit switch <NUM> coupled to the antenna to selectively couple the transmitter <NUM> or receiver <NUM> to the antenna during transmission and reception respectively. The UWB transceiver <NUM> also comprises a spectrum configuration circuit <NUM> (equivalent to Pulse Pattern <NUM> in transmitter <NUM> in <FIG>), PHY Processing circuit <NUM>, Link Controller <NUM>, Buffer and Interface circuit <NUM>, and PHY Formatting circuit <NUM>. The UWB transceiver <NUM> communicates via Link Controller <NUM> to the Client <NUM>. As such, Link Controller <NUM> may communicate using a wired protocol (e.g., serial peripheral interface (SPI)) to Client <NUM>, for example.

Indoor wireless communication is often affected by various propagation phenomena that can mitigate the signal quality if not properly addressed. Multipath interference is an important source of fading that can seriously degrade the link quality in an indoor environment. When radio waves are transmitted in the air, they will be received at the receiver from different paths. Based on their travelling path, obstacles, and reflecting media, these received signals will have various amplitudes and time delays (phase difference). When combined at the receiver, these signals from different paths might partially or, in some cases, completely cancel each other resulting in a significant drop of the signal quality. In dynamic environments where there are moving objects in the room, this fading effect is accentuated. Therefore, it is crucial to address this problem in system level as well as radio and antenna level.

Antenna diversity techniques are often used to mitigate this problem. In this method more than one antenna is used to transmit and receive the wireless signal. Usually in order to have the maximum possible diversity, the antennas are oriented orthogonally to provide polarization diversity. Moreover, placing the antennas at about a quarter wavelength distance between adjacent antennas ensures a spatial diversity which enhances the efficiency of the system. In this structure, the signal will be received from both antennas and compared with each other and the stronger signal will be selected between the two. This technique has been proven to be efficient in mitigating multipath fading in dynamic and static environments.

Many structures have been presented for Ultra-Wide Band (UWB) antenna diversity in the literature. However, most of these structures are based on monopole antennas with a relatively omnidirectional radiation pattern. Using omnidirectional antennas for diversity reduces the efficiency of this technique since the radiation patterns of the antennas will overlap and the signal will still be reaching both antennas from various paths. Accordingly, it is preferred to use multiple directive antennas oriented in different directions to ensure spatial filtering of the undesired signals.

Accordingly, the inventors have established a new type of compact antenna diversity structure to employ with UWB transmitters, UWB receivers and UWB transceivers exploiting techniques according to embodiments of the invention, those described and depicted within <CIT>, <CIT>, and <CIT>, together with other aspects of these UWB devices not described here or elsewhere. The inventive antenna system is composed of two UWB planar microstrip dipole antennas that provide spatial and polarization diversity by having a directive radiation pattern. A compact structure, wideband impedance matching, stable gain over the bandwidth, and perfect envelope correlation between antennas are the main features of this structure making it a strong candidate for high data rate applications.

The structure of the antenna is depicted in <FIG>. It is composed of two wideband dipole antennas with integrated baluns. The two dipoles are placed with <NUM>° angle relative to each other to ensure maximum polarization diversity. As can be observed in <FIG>, the whole structure is compact, low profile, and low cost making it suitable for various high data rate indoor applications.

The antenna has been simulated using full wave FEM method in ANSYS HFSS, a 3D electromagnetic field simulator for RF and wireless design. The |S11| (dB) results are depicted in <FIG> showing that both antennas are perfectly matched from <NUM> to <NUM> covering a desired frequency band of interest of UWB wireless radios developed by Spark Microsystems of Montreal, Canada. The antennas are perfectly isolated with a S21 value below -<NUM> dB from <NUM> to <NUM> as evident from <FIG>. This is mainly due to the directive nature of the antennas that radiate in two distinct directions. A good isolation between antennas will ensure a low envelope correlation and therefore higher diversity gain.

The radiation patterns of the antenna are presented in <FIG> respectively for different frequencies and three different planes: φ=<NUM>°, φ=<NUM>°, θ=<NUM>° respectively. As it can be observed from these <FIG> respectively the radiation pattern is completely directional and only one side of the UWB radio is illuminated at once. This allows for the attenuation of the multipath signals that are being received from the other direction. When the antenna is switched, the other side of the radio will be active. Therefore, a complete spatial diversity is achieved while the combination of the two antennas cover the whole space around the wireless radio module.

In addition to the above-mentioned features, the antenna has a uniform radiation pattern in various frequencies which ensures a stable link quality all over the desired frequency band.

The peak realized gain of the antenna is depicted in <FIG> where it is evident that the peak realized gain remains around <NUM> dB from <NUM> to <NUM>. Therefore, by using this antenna the wireless radio will be able to use a single output power profile over the bandwidth. Moreover, the antenna will attenuate the signal outside of the desired bandwidth. Due to the high realized gain of the antenna, the coverage range of the wireless radio is enhanced in indoor as well as outdoor environments. The radiation efficiency of the antenna is also depicted in <FIG> with more than <NUM>% over the desired range of frequency.

The antenna diversity performance can be analyzed using a metric called the Envelope Correlation Coefficient (ECC). It is a measurement that shows how correlated the antennas are. Since the antennas have completely distinct polarizations and are well isolated, the ECC is expected to be very low for this configuration. ECC can be calculated using the S-parameters of the two antennas using Equation (<NUM>).

The resulting ECC is depicted in <FIG>. As expected, very low values of ECC are achieved over the operating bandwidth resulting in a very good diversity efficiency of the antenna.

This inventive antenna uses a loop element driven differentially. As depicted in <FIG> the loop element <NUM> is coupled to a differential feed <NUM>. Disposed either side of the loop are first and second parasitic (non-driven) elements <NUM> and <NUM> respectively together with a third parasitic element in the center of the loop. The first and second parasitic elements <NUM> and <NUM> are coupled to the ground plane underneath and are used to tune the lower frequency cutoff. The center third parasitic element <NUM> employed to tune the higher cutoff frequency. Further the first to third parasitic elements <NUM> to <NUM> respectively allow for a reduction in the overall size of the antenna.

<FIG> depicts the |S11dd| (dB) results for UWB loop antenna with parasitic elements over the frequency range <NUM> - <NUM> as measured and simulated. Referring to <FIG> the influence of the central resonator on the nominal |S11dd| is depicted whereas <FIG> depicts the influence of the side resonators on the nominal |S11dd|. Accordingly, <FIG> depicts the performance of the antenna with and without all resonators.

The antenna is omnidirectional and has a low gain that is stable in frequency. It is possible to tune the gain pattern by changing the position and length of the resonators. <FIG> depicts the gain of the antenna with respect to frequency. Referring to <FIG> it is evident that the antenna matching is minimally affected by the width of the ground plane wherein |S11dd| (dB) results over the frequency range <NUM> - <NUM> are depicted for different ground plane sizes, from <NUM> to <NUM>. <FIG> and <FIG> depict |S11dd| which is a mixed-mode S-parameter relating to the power reflected in the differential mode.

Similar results were obtained when flesh like substances were near the antenna indicating the antenna can be used in proximity of body parts without being detuned. These results being depicted in <FIG>.

Referring to <FIG> there are depicted two antenna designs employing the same design methodology. In both cases, the substrate has a width of <NUM> (<NUM> inches). The antenna in <FIG> is designed for operation over a frequency range of <NUM> - <NUM> and measures <NUM> x <NUM> (~<NUM> inch x <NUM>. 65inch) whilst that in <FIG> is designed for operation over <NUM> - <NUM> and measures <NUM> by <NUM> (~<NUM> inch x <NUM> inch). Schematics of the two antennas being depicted in <FIG> respectively.

The inventors have further modified the loop antenna to add band-stop filtering as well as allowing for lower frequency cutoff tuning and high frequency cutoff tuning through the parasitic elements, such as first to third parasitic elements <NUM> to <NUM> respectively in <FIG>. Accordingly, the inventors have employed a stub-line filter as depicted in <FIG> and a slot-line filter as depicted in <FIG>. As evident in <FIG> the stub-line filter is implemented as a notched region within the central third parasitic element whilst the slot-line filter is implemented within the edge portion of the ground plane beneath the differential feed to the loop antenna. Accordingly, the band stop filter increase the height of the antenna by <NUM> (~<NUM> inch) whilst the common mode rejection slotline does not change the dimensions.

The band stop filters can reduce the efficiency near the <NUM> and <NUM> ISM bands as evident from <FIG>. The slotline filter reflects the common mode resulting in the performance depicted in <FIG>.

As noted above in Section <NUM> a loop antenna forming part of a diversity antenna has a single feed. Accordingly, a wireless radio exploiting such antennas may require a BALUN (BALanced to UNbalanced) to couple differential circuit of the wireless radio electronics to the single ended feed of the loop antenna. This being, for example, differential outputs of a RF signal generator may be coupled to a single ended transmit antenna or a single ended receive antenna is coupled to differential inputs of a RF receiver front end. A balun is a reciprocal device meaning that it can be used in both directions and accordingly in embodiments of the invention a wireless radio may employ a single diversity antenna for both transmit and receive or within other embodiments of the invention different antennas may be employed for transmit and receiver functionality whilst in other embodiments of the invention a single antenna may be used in conjunction with a receive only wireless radio or a transmit only wireless radio.

If the balanced ports of the balun are excited with two similar signals with the same phase, they will be either rejected or absorbed. This is called the common mode rejection feature of the balun. In some situations, it is desired to absorb the common mode signal and guide it to the ground plane. There are several approaches to design a balun in microwave frequencies. In most of these methods, the balun is usually designed with three ports and the common mode is usually reflected. However, where it is desired to absorb the common mode signal then it is necessary to have a four-port structure from which both the differential and common mode signals can be extracted.

Within this section the inventors describe an innovative compact folded planar balun design based upon a microstrip <NUM>° rat-race hybrid with low loss and wideband performance. The <NUM>° hybrid junction is a four-port network that can be designed in several forms. The rat-race hybrid junction can be designed in planar form using microstrip or coplanar transmission lines and has the benefits of being low profile and compact.

The main feature of a rat-race hybrid is that if it is excited on ports <NUM> and <NUM>, the sum of the signals will be at port <NUM> whereas port <NUM> will output their difference. These ports being depicted in <FIG> which depicts a plan view of the rat-race hybrid balun. Accordingly, this feature can be used to design a balun from a rat-race hybrid. If the ports <NUM> and <NUM> are excited with a differential signal, then the unbalanced signal will be seen at port <NUM> and the common mode signal can be absorbed by a load connected to at port <NUM> as also depicted in <FIG> with a resistor, R_absorb, coupled between port <NUM> and ground. If the hybrid is completely matched, there will be no reflection of the common mode and signal integrity problems can be avoided.

<FIG> depicts a three dimensional depiction of the structure of the rat-race balun which can be seen to employ a folded structure in a rectangular shape thereby allowing the overall dimensions of the structure to be reduced. The dimensions shown in <FIG> of <NUM> x <NUM> (<NUM> inch x <NUM> inch) representing a folded rat-race balun designed for operation over <NUM> to <NUM>. The simulated differential and common mode reflection coefficients are depicted in <FIG> whilst <FIG> depicts the insertion loss of the rat-race balun.

As evident from <FIG> a good differential S11 of below -<NUM> dB from <NUM> to <NUM> and a common mode matching of below -<NUM> dB from <NUM> and above are achieved using this structure. This means that the reflection of the common mode is avoided and, as explained before, can be absorbed from port <NUM>. Moreover, as can be observed in <FIG>, this structure has a very low insertion loss with a maximum of -<NUM> dB at <NUM>. This ensures the efficiency of the system when the balun is integrated with other circuit elements such as filters and antenna.

Another two important features of a balun to be analyzed are the amplitude and phase balance with simulated values being depicted in <FIG> respectively. The amplitude balance indicates the difference in the amplitude of the received signal at port <NUM> when excited from ports <NUM> and <NUM>. Ideally this value should be <NUM> dB over the entire frequency band of interest but typically a maximum difference of about <NUM> dB is acceptable in most applications. This value is respected in the proposed design from <NUM> to <NUM> which is considered a wideband performance.

The phase difference at the output port should be exactly <NUM>° at the center frequency and close to this value for the rest of the desired band. This balun presents a phase difference value of <NUM>° at <NUM> whilst at the lower and upper frequencies of interest at <NUM> and <NUM> respectively the phase difference is <NUM>° and <NUM>° degrees respectively. Accordingly, the phase variation is acceptable over the desired bandwidth.

Beneficially the inventive folded rat-race balun whilst demonstrating good performance over a wideband frequency range of interest, from <NUM> to <NUM>, provides significant benefits over current commercial baluns in that the structure is low cost, it can be printed onto a PCB, has low loss, and does not suffer from a common mode reflection.

As described above in respect of Sections <NUM> to <NUM> the inventors have established innovative wireless radios exploiting multi-frequency pulse bundles per transmitted bit allowing the PSD of each bit to be established through the cumulative spectral properties of the multiple pulses within each multi-pulse bundle. These concepts also being described within <CIT>, <CIT>, and <CIT>.

Referring to <FIG> in first image 2000A there is depicted a frequency band from <NUM> to <NUM> together with the FCC regulated lower and upper frequency limits for the unreserved frequency band within which wireless radio emission spectra (WRES) are depicted. First WRES <NUM> representing a wireless radio operating such that the pulse bundles substantially fill the FCC unregulated spectrum from <NUM> to <NUM>. In contrast, second WRES <NUM> and third WRES <NUM> represent wireless radios operating upon smaller portions of this FCC unregulated spectrum, these being <NUM> - <NUM> and <NUM> - <NUM> respectively. Third WRES <NUM> also depicting the scenario where the wireless radio has established a notch in its PSD at <NUM>. It would be evident to one of skill in the art that the WRES for a wireless radio would be established by local emission regulations for wireless devices. It would be evident to one skilled in the art that wireless radios exploiting embodiments of the invention may operate over different total frequency ranges that that regulated by the FCC according to their deployment scenario such as defined by factors such as interoperability standards, application, environment, and jurisdiction for example. Further, it would be evident to one skilled in the art that wireless radios exploiting embodiments of the invention may operate over different frequency ranges which are portions of the total frequency ranges within those regulated or unregulated according to factors such as interoperability standards, application, environment, and jurisdiction for example. Accordingly, wireless radios according to embodiments of the invention may operate bandwidths such as <NUM> - <NUM>, <NUM> - <NUM>, and <NUM> - <NUM> for example within the environments covered by FCC regulations.

Referring to second image 2000B the frequency range from <NUM> to <NUM> is depicted as being divided into <NUM> bands, I,II,. Each band being <NUM> wide. It would be evident to one skilled in the art that the number of bands may be changed such that either a defined number of bands exist of a width defined by total frequency range divided by the number of bands, or that the bands are specified at a defined frequency range (e.g. <NUM>, <NUM>, <NUM>, <NUM>, etc.) and that the number of bands are therefore defined by the total frequency range divided by this defined frequency range. Whilst the descriptions in respect of <FIG> are described with respect to bands that are all of equal bandwidth it would be evident that within other embodiments of the invention the bands may be unequally defined in bandwidth upon a regular grid or an irregular grid. For example, each band may be consecutive even if equally or unequally defined in bandwidth or they may be non-consecutive with equal or unequal bandwidths on a regular grid or irregular grid.

Accordingly, referring to <FIG> a transmitter Tx_1 <NUM> according to an embodiment of the invention may transmit a pulse bundle or pulse bundles for each data block which provides an emission covering the same band for each sequential data block, in this instance Band IV. Within embodiments of the invention a data block may be a bit, a byte, a packet, or a predetermined portion of data being transmitted. A receiver within another wireless radio may be similarly tuned to Band IV to receive the transmitted pulse bundle(s) from the transmitter Tx_1 <NUM>.

However, another transmitter, Tx_3 <NUM>, may transmit a pulse bundle or pulse bundles for each data block which provides an emission covering the same band for each sequential data block, in this instance Band VIII. Again, a data block may be a bit, a byte, a packet, or a predetermined portion of data being transmitted. Accordingly, a receiver within another wireless radio may be similarly tuned to Band VIII to receive the transmitted pulse bundle(s) from the transmitter Tx_3 <NUM>.

However, within another embodiment of the invention a transmitter Tx_2 <NUM> may transmit a pulse bundle or pulse bundles for each data block which provides an emission range which varies in the band occupied for each sequential data block within an overall frequency range. As depicted Tx_2 <NUM> transmits data blocks upon a recurring <NUM> block sequence of Bands II, VII, XIII and X. Again, a data block may be a bit, a byte, a packet, or a predetermined portion of data being transmitted. However, now in contrast to Tx_1 <NUM> and Tx_3 <NUM> this transmitter Tx_2 <NUM> hops bands per data block. Accordingly, a receiver within another wireless radio may be similarly tuned to Bands II, VII, XII and X in the same sequence as transmitter Tx_2 <NUM> IV to receive the transmitted pulse bundle(s) from the transmitter Tx_2 <NUM>.

It would be evident to one skilled in the art that the number of bands available within the recurring sequence may vary according to factors such as the manner the frequency band has been subdivided, the total frequency range, the bandwidth of each band etc..

Further, would be evident to one skilled in the art that the number of bands available within the recurring sequence may vary and may be <NUM>, <NUM>, <NUM>, <NUM>, <NUM>, <NUM>, <NUM>, etc. and that in some embodiments of the invention each band within the sequence is unique whereas within other embodiments of the invention a portion of the bands within the sequence may occur multiple times although their sequence of occurring changes such that the overall length of the recurring sequence is large. Within embodiments of the invention the band sequence such as number of steps within the recurring sequence, the number of bands used, the actual sequence of the bands within a repetition, etc. may be determined by the transmitter based upon a configuration setting provided to the transmitter or as described below in respect of <FIG> through a spectrum sensing process as part of the configuration process for the transmitter.

As evident from first and second insert images <NUM> and <NUM> then the pulse bundle for a data block within each of Band VII and Band XII is designed by the transmitter to fill the band. Accordingly, for example Band VII is established through a pulse bundle comprising frequencies f<NUM>,f<NUM>,f<NUM> whilst Band XII is established through a pulse bundle comprising frequencies f<NUM>,f<NUM>,f<NUM>. The number of frequencies employed for each band may be constant or variable. The number of pulses of each frequency with the group of frequencies employed for a band may be constant or variable. Further, the parameters of each pulse for a given frequency may be constant or as described below in respect of other embodiments of the invention the parameters for each pulse may be varied deterministically or pseudo-randomly, such parameters including, but not limited to, pulse width, pulse power, dithered frequency offset from nominal frequency, dither time offset from nominal pulse position, and phase. Further the sequence of frequencies within a band may be varied deterministically or pseudo-randomly such that for example, in one sub-set of the pulse bundles for Band VII for example, the pulse sequence employs pulses at frequencies f<NUM>,f<NUM>,f<NUM> whereas another sub-set of pulse bundles may employ pulses at frequencies f<NUM>,f<NUM>,f<NUM>, and yet another sub-set of the pulse bundles may employ f<NUM>,f<NUM>,f<NUM>. It would be evident to one of skill in the art that the more frequencies employed within a pulse bundle the more combinations of frequency sequences can be employed within different pulse bundles even though they are all nominally the same band. It would also be evident that within some embodiments of the invention the pulse sequence for a given band may employ varying number of pulses at each frequency. For example, with a frequency sequence of f<NUM>,f<NUM>,f<NUM> and a pulse count sequence of <NUM>,<NUM>,<NUM> (i.e. <NUM> pulses at f<NUM>, <NUM> pulses at f<NUM>, and <NUM> pulses at f<NUM> that other pulse bundles may be <NUM>,<NUM>,<NUM>; <NUM>,<NUM>,<NUM>; <NUM>;<NUM>;<NUM>; <NUM>,<NUM>,<NUM>; etc. provided that the accumulated power spectrum density (PSD) complies with the regulator requirements. Optionally, the frequency span employable may be defined by the bandwidth of the receiver filter.

Now referring to <FIG> there is depicted schematically a transmission sequence from a transmitter according to an embodiment of the invention wherein during a first sequence <NUM> the transmitter transmits pulse bundles for each sequential data block according to a sequence of Bands II, VII, XII, and X. However, subsequently the transmitter changes to a second sequence <NUM> of Bands IV, VII, XII, and X. In a similar manner as described above a receiver within another wireless radio may be similarly tuned to the first sequence <NUM> of Bands II, VII, XII, and X to receive the transmitted pulse bundle(s) from the transmitter and then be advised of the change to the second sequence <NUM> of Bands IV, VII, XII, and X.

A similar scenario is depicted in <FIG> wherein during a first sequence <NUM> the transmitter transmits pulse bundles for each sequential data block according to a sequence of <NUM> bands comprising Bands II, VII, VIII, and X but the actual sequence of bands within each <NUM> band sequence now varies. Accordingly, within the first sequence <NUM> for bundles N-<NUM> to N-<NUM> the bands are in sequence II, VII, VIII, and X whereas for bundles N-<NUM> to N they are X, VIII, VII, II. The sequence for each band within the <NUM> band sequence may be deterministic or pseudo-random. As noted above the number of bands within the sequence and the actual bands may vary in other scenarios. However, subsequently the transmitter changes to a second sequence <NUM> of Bands II, IV, VIII, and X based upon a configuration adjustment either communicated to the wireless radio by another wireless radio, network controller, etc. or established by the wireless radio through spectrum sensing or monitoring Quality of Service (QoS), for example. Then subsequently at another point in time the transmitter changes to a third sequence <NUM> employing Bands III, IV, VIII and X for bundles N+M, N+M+<NUM>, N+M+<NUM>, and N+M+<NUM>. Accordingly, in a similar manner as described above a receiver within another wireless radio may be similarly tuned to the first sequence <NUM> of Bands II, VII, XII, and X to receive the transmitted pulse bundle(s) from the transmitter, then be advised of the change to the second sequence <NUM> of Bands II, IV, VIII, and X, and then be advised of the third sequence <NUM> employing Bands III, IV, VIII and X.

Now referring to <FIG> there is depicted a schematic band transmission sequence according to embodiments of the invention wherein a transmitter prior to transmitting data blocks using pulse bundles initiates a spectrum sensing phase, Sense #<NUM><NUM>, wherein the transmitter transmits pulse bundles that cover all or a predetermined portion of each Band I through VII using spectral sense pulse bundles Spec_Tx(<NUM>) <NUM> to Spec_Tx(N) <NUM>. Accordingly, the transmitter is established to operate in Band I, for example in dependence upon link data provided to the transmitter from another transmitter associated a receiver within another wireless radio receiving the spectral sense pulse bundles Spec_Tx(<NUM>) <NUM> to Spec_Tx(N) <NUM> or through a communications interface from a network control receiving data from one or more receivers (wireless radios) which receive the spectral sense pulse bundles Spec_Tx(<NUM>) <NUM> to Spec_Tx(N) <NUM>. Once configured to transmit in Band I the transmitter transmits signals within a sequence Transmit #<NUM><NUM> comprising a recurring sequence of Tx_A <NUM> and Tx_B <NUM> each of which comprises a series of four pulse bundles emitted upon s-bands of Band I, these being Sub-Bands A-H wherein Tx_A <NUM> comprises Sub-Bands A, D, F, and H whereas Tx_B <NUM> comprises Sub-Bands B, C, E, and G. These may for example as described and depicted in <FIG> and <FIG> may repeat continuously until at a subsequent point in time a second spectrum sensing phase, Sense #<NUM><NUM>, wherein the transmitter transmits pulse bundles that cover all or a predetermined portion of each Band I through VII using spectral sense pulse bundles Spec_Tx(<NUM>) <NUM> to Spec_Tx(N) <NUM>.

As depicted, this results in the transmitter switching to a second sequence Transmit #<NUM><NUM> wherein the pulse bundles are now within Band IV and sequentially pulse bundles are in Sub-Bands A, D, F, and H of this new band. Alternatively, the second spectrum sensing phase, Sense #<NUM><NUM>, may have resulted in a decision for no change in band or it may have determined a transition to one of the other bands II, III, and V to VII was more appropriate. The subsequent spectrum sensing phases may, for example, be triggered automatically after either a predetermined number of pulse bundles have been transmitted, a predetermined number of packets have been transmitted or a predetermined period of time has elapsed since the last spectrum sensing process was executed. Alternatively, the determination to trigger another spectrum sensing process may be established in dependence upon data provided to the transmitter from a receiver forming part of a link with the transmitter, data received from a network controller, or another receiver.

Optionally, the decision making process within UWB wireless radios according to embodiments of the invention may be dictated by monitoring QoS via techniques such as packet error rate (PER) monitoring or signal-to-noise ratio (SNR) monitoring, for example, either alone or in combination with spectrum sensing etc..

Accordingly, referring to <FIG> there are depicted three different exemplary scenarios. Within the first scenario the transmitter transmitting with pattern Transmit #<NUM><NUM> determines after a spectrum sense process that no reconfiguration is required and so continues to transmit with Transmit #<NUM><NUM> which is the same as Transmit #<NUM><NUM>. Within the second scenario the transmitter transmitting with pattern Transmit #<NUM><NUM> determines after a spectrum sense process that no reconfiguration is required and so shifts to transmit with Transmit #<NUM><NUM> which is now in Band II. Within the third scenario the transmitter transmitting with pattern Transmit #<NUM><NUM> determines after a spectrum sense process that a reconfiguration is required and so transmits with Transmit #<NUM><NUM> which is now in Band VII.

Within <FIG> there is depicted another scenario according to embodiment of the invention wherein a transmitter according to an embodiment of the invention wherein the transmitter operates upon Bands I to XV and is initially configured to transmit upon Bands II, VII, X, and XIII with sequence Transmit #<NUM><NUM>. Subsequently, a spectrum sense process results in the decision to shift to Transmit #<NUM><NUM> wherein the sequence comprises Bands III, VI, XI and XII. Then again at another subsequent point in time another spectrum sense process results in the decision to shift to Transmit #<NUM> wherein the sequence comprises Band IV, VI, VII, and X. Accordingly, the scenario depicted in <FIG> may relate to a plurality of bands such as described within <FIG> respectively or it may relate to a series of sub-bands within a band such as described within <FIG> respectively. However, within other embodiments of the invention the spectrum sensing and adjustments may relate to frequencies within a band or sub-band.

Accordingly, a wireless link according to embodiments of the invention may be configured to operate upon a plurality of bands and/or a plurality of sub-bands wherein the transmitter and receiver are configured to operate upon the same bands and/or sub-bands such that the transmitter and receiver are aligned to the same bands and/or sub-bands by one or processes including, but not limited to:.

Within embodiments of the invention a transmitter within a first wireless radio and a receiver within a second wireless radio may exploit a discovery protocol at the onset of communications in order to ensure that the transmitter and receiver find each other not only in time but also in respect of which frequency band(s).

Referring to <FIG> there is depicted an exemplary pulse bundle according to an embodiment of the invention comprising first to ninth pulses <NUM> to <NUM> respectively which are positioned within the pulse bundle as being centered upon times t<NUM>, t<NUM>, t<NUM>, t<NUM>, t<NUM>, t<NUM>, t<NUM>, t<NUM>, and t<NUM> respectively. These pulses having widths W<NUM>, W<NUM>, W<NUM>, W<NUM>, W<NUM>, W<NUM>, W<NUM>, W<NUM>, and W<NUM> respectively. First to third pulses <NUM> to <NUM> being at a first frequency f<NUM>, fourth to sixth pulses <NUM> to <NUM> being at second frequency f<NUM>, and seventh to ninth pulses <NUM> to <NUM> being at f<NUM>. However, within embodiments of the invention in the generalized construction a pulse bundle may comprise:.

An RF signal generator according to an embodiment of the invention generating such a pulse bundle is as depicted in <FIG>. Further, each pulse within the pulse bundle may have a phase associated with it ϕ<NUM>, ϕ<NUM>, ϕ<NUM>, ϕ<NUM>, ϕ<NUM>, ϕ<NUM>, ϕ<NUM>, ϕ<NUM>, and ϕ<NUM>. With an RF signal generator as depicted in <FIG> then each of the pulses is assigned to either a first phase or a second phase according to the setting of the Switch <NUM>. As described within <CIT> the Switch <NUM> is fed with a pseudo-random signal such that the phases of the pulses within the pulse bundle have a pseudo-randomly assigned phase.

However, within a wireless link between a first wireless radio comprising a transmitter according to an embodiment of the invention and a second wireless radio comprising a receiver according to an embodiment of the invention then amongst the signal degradations that the wireless signals may experience are multipath interference and cross-fading.

Now referring to <FIG> there are depicted first to fourth pulse bundles <NUM> to <NUM> respectively each comprising <NUM> pulses a first frequency f<NUM>, <NUM> pulses at a second frequency f<NUM>, and <NUM> pulses at a third frequency f<NUM>. As first to fourth pulse bundles <NUM> to <NUM> respectively each comprise the same frequencies and the same number of pulses at each frequency then the emitted power spectrum for each pulse is the same independent of the phase of the pulses provided that the pulses in each of the first to fourth pulse bundles <NUM> to <NUM> respectively have the same amplitude. However, as evident the actual pulse sequences in each of the first to fourth pulse bundles <NUM> to <NUM> respectively are different, these being:.

Now referring to <FIG> there is depicted a packet <NUM> comprising a header <NUM> and a plurality of data blocks <NUM>(<NUM>), <NUM>(<NUM>), <NUM>(<NUM>) to <NUM>(N). As depicted:.

Referring to <FIG> there is depicted a series of packets <NUM> to <NUM> transmitted by a transmitter according to an embodiment of the invention. Within this embodiment of the invention the frequencies within each of the packets <NUM> to <NUM> are the same but their sequence varies. Accordingly, as depicted:.

Now referring to <FIG> there is depicted a series of packets <NUM> to <NUM> transmitted by a transmitter according to an embodiment of the invention. Within this embodiment of the invention not only do the sequences of the pulses vary (i.e. it is not a constant sequence with respect to frequencies) but the frequencies themselves vary as each packet is in a different band of the UWB link spectrum accessible according to the constraints defined, for example, by the UWB transmitter and receiver together with regulatory requirements. This, for example, being a repeating sequence of bands, such as described above in respect of <FIG> respectively. Accordingly, as depicted:.

Further, as described above with respect to <FIG> it would be evident that the pulses within each of the first to fourth packets <NUM> to <NUM> respectively in <FIG> or first to fourth packets <NUM> to <NUM> respectively in <FIG> may also have other aspects of the pulses such as inter-pulse spacing, phase, pulse width, etc. can be varied within the sequence either pseudo-randomly or programmatically. Optionally, within other embodiments of the invention a programmable or fixed time guard band between symbols may be employed. Optionally, within other embodiments of the invention the symbol repetition rate may be varied to accommodate locations that have degraded link quality through inter-symbol interference.

Within the preceding descriptions with respect to <FIG> multi-frequency pulse bundles are employed within each bit, within each symbol or within a packet or across multiple bits, multiple symbols, or multiple packets. However, in <FIG> and <FIG> the multi-frequency bundles may be within a single band until a decision to adjust to another band or to shift a band or bands within a repeating sequence of bands may be made through a method such as spectrum sensing as described within <FIG>. However, within another embodiment of the invention the decision to shift from one band to another, to shift a band or bands within a repeating sequence of bands, or shift the frequency sequence within a multi-pulse bundle may be made in dependence upon a degradation of the link properties between the transmitter and one or more receivers either fed back to the transmitter from the one or more receivers or through a network controller. Such degradations in the link properties arising from the aforementioned effects such as multi-path interference and cross-fading.

Accordingly, the transmitter may either within a training sequence or during transmission of data adjust the sequence of frequencies within a multi-pulse bundle such as described and depicted in <FIG> and <FIG> to determine whether an adjust in the frequency sequence of the pulses within the multi-frequency bundle results in mitigation of the signal degradations observed such as through absolute received power at the receiver and/or an effective signal to noise ratio between the pulse bundles for a "<NUM>" and the pulse bundles for "<NUM>".

In a similar manner a decision may be made to shift to another band of a plurality of bands to determine whether exploiting a different band similarly yields an improved in received signal quality.

The adjustment from one frequency sequence to another may be programmatically defined by a configuration sequence stored within a memory associated with the transmitter or it may be pseudo-randomly established from a series of configuration sequences stored within the memory. Alternatively, the adjustment from one frequency sequence to another may be determined from spectrum sensing discretely or in combination with one or more link metrics.

As noted in <FIG> the pulses within a multi-pulse bundle are defined by their position within the pulse bundle, which may be varied as described above in Section 8A, together with their frequency, pulse width, phase, and amplitude and the pulse repetition rate of the pulses from the RF signal generator within the transmitter.

However, these parameters may be dithered either programmatically or pseudo-randomly in order to adjust the actual content of the multi-frequency pulse bundle. For example, this may be by:.

Accordingly, adjustments in these parameters when programmatically controlled allow the receiver to determine how the link quality of service (QoS) metrics improve or degrade and sequentially adjust them to obtain improved link QoS. Further, adjusting the PRR allows the transmitter to adjust the spacing of pulses within a multi-pulse bundle to similarly determine how the link QoS metrics improve or degrade. Optionally, a transmitter according to embodiments of the invention may dynamically adjust the spacing of symbols dynamically in order to adjust and seek to enhance the QoS, albeit at the cost of maximal data rate sustainable.

Within the descriptions in respect of <FIG> each pulse within a multi-frequency bundle has been described and depicted as being at a single discrete frequency. However, within other embodiments of the invention the pulses may be chirped such that a predetermined chirp is applied to each pulse. This may be an increase of frequency during the pulse, up-chirp; a decrease of frequency during the pulse, down-chirp; an up-chirp followed by a down-chirp; or a down-chirp followed by an up-chirp. The applied chirp may be linear, wherein the chip varies linearly with time during the pulse or it may be geometric wherein it varies according to a geometric relationship with time during the pulse. For example, a geometric chirp may be sinusoidal or exponential.

Accordingly, a transmitter according to embodiments of the invention may apply varying degrees and form of chirp to the pulses within a multi-pulse bundle over time such that QoS metrics can be measured and the transmitter parameters adjusted for improved QoS and to continuously adjust the different aspects of the pulses to continuously tune for improved QoS.

Within the preceding descriptions with respect to <FIG> multi-frequency pulse bundles are employed within each bit, within each symbol or within a packet or across multiple bits, multiple symbols, or multiple packets. However, in <FIG> and <FIG> the multi-frequency bundles may be within a single band until a decision to adjust to another band or to shift a band or bands within a repeating sequence of bands may be made through a method such as spectrum sensing as described within <FIG>. However, within another example the decision to shift from one band to another, to shift a band or bands within a repeating sequence of bands, or shift the frequency sequence within a multi-pulse bundle may be made in dependence upon a degradation of the link properties between the transmitter and one or more receivers either fed back to the transmitter from the one or more receivers or through a network controller. Such degradations in the link properties arising from the aforementioned effects such as multi-path interference and cross-fading.

Alternatively, within other examples a decision may be made at the beginning of establishing a link between the transmitter and receiver wherein an initial training sequence can be employed such that the transmitter cycles through a predetermined sequence of variations in the properties of the multi-frequency pulse bundles such as stepping through the different frequency bands etc. in order for the receiver to determine QoS metrics for each band and establish a configuration which is subsequently employed in the link. Optionally, within examples the initial configuration may be employed until the QoS drops below a predetermined threshold wherein the transmitter may trigger a new training sequence or a spectrum sensing process etc. in order to establish new transmitter configuration settings.

As described above in respect of <FIG> diversity can provide enhanced link performance in the presence of signal degradations such as those arising from multi-path interference, fading, etc. Accordingly, when implementing UWB transmitters, UWB receivers, and UWB transceivers diversity antennas may be enhanced with directive antennas in order to provide improved spatial filtering. Accordingly, antennas which are referred to as Vivaldi antennas provide not only a directive radiation pattern but also an ultra-wideband performance. Accordingly, the inventors have established diversity antenna structures incorporating two oppositely directed Vivaldi antennas to provide a maximum coverage around the UWB wireless radio. As depicted in <FIG> a diversity antenna comprises a first Vivaldi antenna <NUM> and a second Vivaldi antenna <NUM> which are located side by side with respect to each other with each being fed by a co-planar waveguide transmission line. The co-planar waveguides are connected within UWB devices to an RF switch which will connect either of the antennas to the transmitter and/or receiver circuit with UWB transmitter, UWB receiver, and UWB transceiver etc..

The exemplary antenna diversity systems for which results are presented in <FIG> were designed to fit with the limited dimensions of the PCB hence making it a compact system. The structure is simulated in the FEM based simulation software, Ansys HFSS with the total dimensions of <NUM>×<NUM>. The simulated |S<NUM>|(dB) is depicted in <FIG> which depicts an ultra-wideband impedance matching performance from <NUM> to <NUM> ensuring low loss performance of the antennas. The two antennas are well isolated since their radiation pattern is directed in two distinct direction. This ensures a very low ECC hence increasing the diversity gain of the system. The simulated ECC being depicted in <FIG>.

<FIG> respectively depicted the antenna radiation patterns over the frequency band of interest for a diversity antenna exploiting Vivaldi. These being at <NUM>, <NUM>, and <NUM> respectively. From these it is evident that the radiation pattern uniformity is maintained over the desired bandwidth whilst the peak realized gain ranges between <NUM> dBi to <NUM> dBi is obtained over the range <NUM> to <NUM> which a sub-band of the overall operating bandwidth of the diversity antennas which is employed with UWB wireless devices manufactured by Spark Microsystems of Montreal, Canada. <FIG> depict the peak realized gain and radiation efficiency of the designed antenna.

As noted above in respect of the diversity antennas described within Section <NUM> and <NUM> respectively with the dual antennas a UWB transmitter can be connected to either one of the two antennas within the exemplary diversity antenna structures. However, where the UWB transmitter incorporates a differential structure then the UWB transmitter has a differential output. Accordingly, referring to <FIG> for a transmitter a UWB Circuit <NUM> provides a differential RF output which is coupled to an RF switch <NUM> which provides an equivalent functionality as double pole double throw switch wherein in one state the RF Switch <NUM> connects the differential RF output of the UWB Circuit <NUM> to the first antenna, Antenna #<NUM><NUM>, and then in the other state the RF Switch <NUM> connects the differential RF output of the UWB Circuit <NUM> to the second antenna, Antenna #<NUM><NUM>. Antenna #<NUM><NUM> and Antenna #<NUM><NUM> being the two antennas within Diversity Antenna <NUM>. Based upon the switching speed of the RF Switch <NUM> the UWB device employing the configuration depicted in <FIG> can reconfigure at the per-packet level. Optionally, where a UWB receiver supports an RF front end with differential capabilities then the circuit depicted in <FIG> can be employed to connect the dual outputs of an antenna to a receiver UWB circuit.

Now considering a UWB receiver then referring to <FIG> there is depict a Diversity Antenna <NUM> comprising Antenna #<NUM><NUM> and Antenna #<NUM><NUM>. Antenna #<NUM><NUM> is connected to a first RF front end, RF Receiver Signal Chain #<NUM><NUM>, whilst Antenna #<NUM><NUM> is connected to a second RF front end, RF Receiver Signal Chain #<NUM><NUM>. An output from each of RF Receiver Signal Chain #<NUM><NUM> and RF Receiver Signal Chain #<NUM><NUM> is coupled to a control circuit <NUM> which determines which of the Receiver Signal Chain #<NUM><NUM> and RF Receiver Signal Chain #<NUM><NUM> provides improved signal reception and accordingly powers down the other circuit. Due to the fast power cycling of UWB receivers established by the inventors then UWB receivers can determine, for example, from a preamble within each packet which antenna to employ, select the appropriate antenna, and power down the other antenna such that the UWB can establish a configuration per packet. Optionally, the Antenna #<NUM><NUM> and Antenna #<NUM><NUM> may each provide hemispherical radiation patterns such that the pair form a spherical coverage.

An antenna subsystem is required to establish a wireless link between a UWB transceiver (radio) and the outside world such as described and depicted in <FIG>, for example. This subsystem may include several components that are necessary for the performance of the whole system. Moreover, this subsystem should shape the output spectrum of the UWB radio in order to comply with the stringent requirements of a spectrum mask. This spectrum mask may, for example, be associated with a geographical region. Within <FIG> there are depicted exemplary architectures for IR-UWB transmitters with and without biphasic scrambling according to embodiments of the invention. Within such IR-UWB transmitters may employ a differential output signal therein requiring a Balun to convert the balanced signal to an unbalanced (single-ended) signal. As depicted in <FIG> in first and second images 3800A and 3800B respectively depicting exemplary architectures for UWB radio modules which exploit a chip (circuit) balun in first image 3800A or a printed circuit board (PCB) based balun in second image 3800B.

First image 3800A depicts a UWB radio <NUM> which provides a differential output signal to a second harmonic rejection filter <NUM> which then generates a differential output signal to a chip balun <NUM>. The chip balun <NUM> generates a single-ended signal which is passed to the antenna <NUM> via a low loss passband filter <NUM>. Second image 3800B depicts a UWB radio <NUM> which provides a differential output signal to a PCB balun/filter <NUM>. The output of the PCB balun/filter <NUM> is a single-ended signal which is coupled to an antenna <NUM>.

Referring to <FIG> there are depicted first to fourth images 3900A to 3900D respectively for a chip balun antenna sub-system wherein each of the first to fourth images 3900A to 3900D represents a different layer of a PCB employing a UWB radio in combination with a chip balun antenna subsystem. Within these first to fourth images 3900A to 3900D respectively there are depicted first to ninth elements <NUM> to <NUM> respectively. These comprise:.

Fourth image 3900D representing the bottom PCB layer. Accordingly, the subsystem presented in <FIG> provides a uniform and omni-directional radiation pattern over the whole frequency range of the UWB radio 3900E, an effective and low loss filtering of the second and third harmonics as well as low frequency undesired radiations, a shielded structure with suppressed cavity resonance modes for low electromagnetic interference, and a compact low loss balun.

Referring to <FIG> there are depicted antenna radiation patterns over the frequency band of interest for the UWB antenna subsystem as depicted in <FIG> over the frequency range <NUM> to <NUM>. <FIG> depict the S11 and peak gain for the UWB antenna subsystem as depicted in <FIG>.

Referring to <FIG> there is depicted a PCB balun/filter circuit <NUM> for transforming a balanced signal to a single-ended signal such as the PCB Balun / Filter <NUM> in second image 3800B in <FIG>. Within PCB balun/filter <NUM> there are depicted first to sixth elements <NUM> to <NUM> respectively. These comprise:.

Accordingly, the inventors have established a PCB balun/filter circuit <NUM> which provides excellent filtering capability of the high frequency spurious harmonics together with integration of filtering and balun function into one PCB structure leading to a compact and low-cost solution. The PCB balun/filter circuit <NUM> provides an omni-directional radiation pattern with uniform gain around the antenna module for the whole required frequency band.

Accordingly, the innovations within the PCB balun/filter circuit <NUM> include the use of ground plane slots for smoothing the radiation pattern and providing a uniform gain around the antenna for all frequencies in conjunction with a compact PCB balun/filter structure with excellent filtering of the high frequency spurious signals.

Referring to <FIG> there are depicted antenna radiation patterns over the frequency band of interest for the UWB antenna subsystem as depicted in <FIG> over the frequency range <NUM> to <NUM>. <FIG> depict the S11 and peak gain for the UWB antenna as depicted in <FIG>.

As described above a UWB radio may be directly connected to a differential antenna or indirectly to a single ended antenna using a balun. As described above these single ended antenna modules may have integrated filtering capabilities to reject undesired spurious signals. However, this is possible where the extra cost and insertion loss of a balun can be accepted. However, within some UWB systems these penalties cannot be accepted and accordingly examples disclosed herein have established new UWB antenna modules to address employing either a filtered loop antenna methodology as described below in section 13A or a differential antenna module such as described below in section 13B. Within each harmonics rejection is achieved using a differential low pass filter thereby removing the requirement for a balun which in turns reduces the losses and the fabrication costs associated with this additional component.

Referring to <FIG> there is depicted an exemplary filtered loop antenna module 4500A having overall dimensions of <NUM> (<NUM>") by <NUM> (<NUM>"). Within filtered loop antenna module <NUM> there are depicted first to third elements <NUM> to <NUM> respectively. These comprise:.

Beneficially the filtered loop antenna module <NUM> provides for a compact and low loss structure for the novel differential low pass filter which provides for a wide rejection region of the second and third harmonics together with a low cost efficient structure not requiring a balun. Further, the high pass nature of the antenna combined with the low pass filter provides a bandpass radiation of the incident wave such as depicted in Figure as it can be observed in <FIG>.

Referring to <FIG> there are depicted the S11 and radiation efficiency for the UWB antenna subsystem as depicted in <FIG>. Accordingly, it is evident that the filtered loop module <NUM> provides for good impedance matching from <NUM> to <NUM> which covers the required frequency bandwidth of an exemplary UWB radio such as that manufactured by Spark Microsystems of Montreal, Quebec, Canada. Moreover, the radiation efficiency plot in <FIG> shows the rejection of the high frequency signals at the harmonics as well as low frequency signals. Now referring to <FIG> depict antenna radiation patterns over the frequency band of interest for the UWB antenna subsystem as depicted in <FIG>.

A similar structure of the feedline and filter as described above in respect of <FIG> may also be employed with different differential antenna types such as the one depicted in the <FIG>. Accordingly, <FIG> depicts an exemplary differential patch antenna module. Within differential patch antenna module <NUM> there are depicted first to third elements <NUM> to <NUM> respectively. These comprise:.

As evident from <FIG> this antenna provides a more compact module with smaller overall dimensions whilst offering, as may be observed from the radiation pattern plots in <FIG> respectively, the antenna gain on the sides (Phi=<NUM>°, Theta=<NUM>°) is higher than compared to the loop antenna. This results in improved coverage within indoor environments for example.

Referring to <FIG> depict antenna radiation patterns over the frequency band of interest for the UWB antenna subsystem as depicted in <FIG>. Now referring to <FIG> there are depicted the S11 and radiation efficiency for the UWB antenna subsystem as depicted in <FIG>. Accordingly, it is evident that the antenna shows good impedance matching over the frequency range of interest, <NUM> to <NUM>, and that the radiation efficiency shows the bandpass nature of the overall structure which results in efficient rejection of undesired signals.

Implementation of the techniques, blocks, steps and means described above may be done in various ways. For example, these techniques, blocks, steps and means may be implemented in hardware, software, or a combination thereof. For a hardware implementation, the processing units may be implemented within one or more application specific integrated circuits (ASICs), digital signal processors (DSPs), digital signal processing devices (DSPDs), programmable logic devices (PLDs), field programmable gate arrays (FPGAs), processors, controllers, micro-controllers, microprocessors, other electronic units designed to perform the functions described above and/or a combination thereof.

The foregoing disclosure of the exemplary examples and embodiments of the present invention has been presented for purposes of illustration and description. It is not intended to be exhaustive or to limit the invention to the precise forms disclosed. Many variations and modifications of the embodiments described herein will be apparent to one of ordinary skill in the art in light of the above disclosure. The scope of the invention is to be defined only by the claims appended hereto.

Claim 1:
A transmitter (<NUM>, <NUM>) for an impulse radio comprising:
a radio frequency, RF, signal generator (<NUM>, <NUM>) for receiving a data signal to be transmitted and a clock signal characterised by a clock frequency and coupled to an RF antenna (<NUM>) and a control circuit where a pulse repetition rate of the RF signal generator (<NUM>, <NUM>) is determined in dependence of the clock frequency; where
the control circuit is adapted to configure the generation of the transmitted data by the RF signal generator (<NUM>, <NUM>) such that each bit being transmitted is comprised of a plurality N pulses generated by the RF signal generator within the duration of the bit of the data signal wherein each pulse of the N pulses is at a predetermined frequency of a plurality M frequencies, has a predetermined amplitude, and has a predetermined pulse length where N is an integer, N ≥ <NUM>, and N depends upon a duration of the bit of the data signal and the pulse repetition rate of the RF signal generator, M is an integer and M ≥ <NUM> and the plurality N pulses are within a predetermined frequency band;
wherein the predetermined frequency band is a sub-band of a plurality of sub-bands of an operating band of the transmitter (<NUM>, <NUM>);
said transmitter (<NUM>, <NUM>) being characterized in that:
the control circuit is adapted to configure the RF signal generator (<NUM>, <NUM>) to initially cycle through a first subset of the plurality of sub-bands in a predetermined order for a number of cycles and subsequently cycle through a second subset of the plurality of sub-bands in a predetermined order for a number of cycles where the control circuit is adapted to determine to reconfigure the RF signal generator (<NUM>, <NUM>) from the first subset of the plurality of sub-bands to the second subset of the plurality of sub-bands in dependence upon quality of service data received by the transmitter (<NUM>, <NUM>) from at least one of first receiver and a network controller comprising a second receiver; wherein
the at least one of the first receiver and the second receiver receive the transmitted data from the transmitter (<NUM>, <NUM>).