Patent Description:
Previous solutions involve low-noise amplifiers and down-conversion mixers followed by tunable analog baseband gm-C filters. Other solutions involve the use of N-path mixers running at a single local oscillator (LO) frequency. Higher order filtering in such receivers have been demonstrated by driving the N-path mixer with higher order impedance loads.

High-order analog baseband gm-C filters have inherent quality factor-dynamic range/distortion trade-offs. Those are elaborate analog designs that do not scale with process and require redesign for each process node. As such, those are sensitive to process corners that results in over-design which requires cost, area, and power. Additionally, those analog baseband filters come after a low noise amplifier (LNA) that amplifies not only the desired signal but also in-band blockers, which eventually limits the performance of the multi-channel transceiver.

Achieving a high tunable range in bandwidth, while maintaining higher order filtering, in analog baseband gm-C filters and higher-order single-frequency N-path mixers involves extensive tuning of capacitors, resistors, transistor sizes, and bias currents. Such filters also have a poor yield due to process variations. To account for all of these, an extensive overdesign of such filters is needed, leading to power penalty.

Switch size limits the far-out rejection offered by the single-frequency N-path mixers. A prior two-frequency N-path filter solution does not have a simple method of tuning the center frequency/bandwidth and has limited isolation between the two N-path filters.

<CIT> discloses an LNA for carrier aggregation. 6A and 6B show the structure of LNAs. <CIT> discloses that the LNA includes first and second amplifier stages, and the amplifier stages include an input capacitor, an input transistor, a degeneration inductor, and cascode transistors.

<CIT> relates to radio frequency (RF) low noise amplifier (LNA) and RF power splitters. <CIT> discloses a dual output RF LNA.

<CIT> discloses an amplifier including N-path filters that are coupled in series as shown in <FIG>, <FIG>, and <FIG>. <CIT> discloses that the frequency of local oscillation signals driving the N-path filters is at the first center frequency fLO and the second center frequency fLO + Δf.

<NPL>, discloses a wideband CMOS receiver frontend for radio applications. Araujo Filipe et al. discloses that in order to obtain channel selection with image-rejection and out-of-band interferers attenuation, both low noise amplifier (LNA) and mixer incorporate a N-Path signal processing technique.

<NPL>, discloses reconfigurable RF front-ends for cellular receivers. Mirzaei A et al. discloses that frequency translation technique enables fully integrated reconfigurable multi-band multi-mode cellular receivers that use no external RF components, and that utilizing the concept of N-path filtering high-Q bandpass filters can be implemented to attenuate strong blockers.

<CIT> discloses a low noise amplifier circuit that does not require a surface acoustic filter. <CIT> discloses a low noise amplifier circuit including a cascaded multistage amplification circuit, wherein one or two sets of N-path filters are bridged in the second stage amplifying circuit.

The invention is defined by the subject matter as claimed in claim <NUM>.

Accordingly, while further examples are capable of various modifications and alternative forms, some particular examples thereof are shown in the figures and will subsequently be described in detail. However, this detailed description does not limit further examples to the particular forms described. Further examples may cover all modifications, equivalents, and alternatives falling within the scope of the disclosure. Like numbers refer to like or similar elements throughout the description of the figures, which may be implemented identically or in modified form when compared to one another while providing for the same or a similar functionality.

It will be understood that when an element is referred to as being "connected" or "coupled" to another element, the elements may be directly connected or coupled or via one or more intervening elements. If two elements A and B are combined using an "or", this is to be understood to disclose all possible combinations, i.e. only A, only B as well as A and B. An alternative wording for the same combinations is "at least one of A and B". The same applies for combinations of more than <NUM> elements.

Examples are disclosed for a novel bandpass low-noise amplifier (LNA). A bandpass LNA is an LNA that embeds a high order digitally tunable bandpass filter in the LNA to allow improved rejection to external and internal blockers. The bandpass LNAs according to the examples disclosed herein are digitally tunable both for bandwidth and center frequency of the passband. In example LNAs, a digitally tunable bandpass filter that is embedded in the LNA is built by summing up multiple N-path filters. Each N-path filter is driven by a different local oscillator (LO) signal. The LO frequencies driving the multiple N-path filters may be precisely controlled by a frequency synthesizer (e.g., a digital-to-time converter (DTC)-based frequency synthesizer). The LO frequencies that drive the multiple N-path filters are spaced closely around the center frequency of the band of interest. The spacing between the LO frequencies driving the N-path filters may be varied to achieve a tunable bandwidth.

In examples, the bandpass LNA may include a plurality of cascode amplifiers and a plurality of N-path filters. A cascode amplifier is a two-stage amplifier comprising a transconductance amplifier followed by a buffer amplifier. A cascode amplifier may be constructed using field effect transistors (FETs) or bipolar junction transistors (BJTs). One stage of a cascode amplifier is configured in a common source/common emitter mode and the other stage of the cascode amplifier is configured in a common gate/common base mode. A cascode amplifier includes a common source (or common emitter) device feeding into a common gate (or common base) device. Hereafter, examples will be explained with reference to a cascode amplifier including a common source device feeding into a common gate device, but the examples are applicable to a cascode amplifier including a common emitter device feeding into a common base device.

The LNA is split into multiple segments. Each segment of the LNA includes one cascode amplifier and one N-path filter. In each segment, the N-path filter may be coupled to the source node of the common gate device of the cascode amplifier. The input signal may be a differential signal and the positive and negative polarity of the differential input signals may be appropriately fed to different segments, so that the summed output yields the desired bandpass response. The summed bandpass output may be down-converted to baseband by an N-path mixer driven by an LO frequency equal to the center frequency of the band of interest.

An N-path filter may be based on switched capacitors. Each of the plurality of N-path filters includes N paths coupled in parallel, N being a positive integer greater than one. Each path in an N-path filter may include a switch and a capacitor coupled in series. Each N-path filter is coupled to a different one of the plurality of cascode amplifiers. The plurality of N-path filters are driven by LO signals having different frequencies. The output nodes of the plurality of cascode amplifiers are coupled in parallel. The LO signals that drive the plurality of N-path filters may be symmetrically spaced around a desired frequency (fLO).

In examples, the LNA includes a linear combination of N-path filters driven by LO signals. The frequencies of the LO signals may be spaced with symmetrical offsets around the desired frequency (fLO) to implement a bandpass filter with higher order roll-off centered at fLO. The input signal to be amplified by the cascode amplifiers is a differential signal. The input signal is fed to one or more pairs of cascode amplifiers. In a non claimed example, the number of N-path filters is two for each half circuit and the two N-path filters may be driven by LO signals at fLO+Δf and fLO-Δf. According to one implementation of the invention the number of N-path filters is four for each half circuit and one pair of N-path filters coupled to a positive (or negative) port of a differential input may be driven by LO signals at fLO+Δf<NUM> and fLO-Δf<NUM>, and another pair of N-path filters coupled to a negative (or positive) port of a differential input may be driven by LO signals at fLO+Δf<NUM> and fLO-Δf<NUM>.

In example LNAs disclosed herein, the tunable bandwidths may be achieved through precise (digital) control of the LO frequencies (i.e., the LO frequency offset) driving the different N-path filters, with minimal tuning of capacitors and no tuning of other analog components like resistors, transistor sizes, bias currents, etc. This can lead to very high yield despite variations induced by processing and component mismatch. Alternatively, the baseband capacitance of the N-path filter may also be adjusted to control the gain of the LNA.

Summing up multiple N-path filters with appropriate phase results in higher order bandpass responses. For example, summing up four separate N-path filters, followed by down-conversion to baseband by an N-path mixer yields a response with <NUM> dB/decade RF selectivity. The filter also provides enhanced blocker rejection due to the higher order roll-off.

Access to only the <NUM>° and <NUM>° phase of the input RF signal may be sufficient to synthesize such a response, making it amenable to differential implementations.

A conventional scheme of synthesizing a <NUM>-frequency N-path bandpass filter exhibited limited isolation between the two N-path filters. The multi-frequency N-path bandpass filters in accordance with the invention disclosed herein have inherent isolation since it is placed at the source of the common gate device of the cascode LNA.

The examples disclosed herein have advantages over single-frequency N-path implementations that the far-out rejection is not limited by switch ON resistance but by mismatch between ON resistance of different switches, leading to enhanced far-out rejection.

<FIG> and <FIG> show a non claimed example LNA <NUM> for a <NUM>-frequency N-path bandpass filter. <FIG> shows a half circuit of an LNA <NUM> and <FIG> shows a full differential circuit of an LNA <NUM>. Each half circuit generates one polarity of the differential output. The structure of both halves 100a, 100b is the same but the polatiry of the input signals to the cascode amplifiers 120a, 120b and the LO signals driving the N-path filters 130a, 130b of each half 100a, 100b are inversed. Hereafter, the example will be explained with reference to the half circuit in <FIG> for simplicity.

The example LNA <NUM> includes two N-path filters driven by fLO - Δf and fLO + Δf to synthesize a bandpass filter (e.g., with <NUM> dB/decade high-frequency roll-off centered at fLO). In this example, the LNA <NUM> includes two cascode amplifiers 120a, 120b and two N-path filters 130a, 130b for each half circuit. The LNA <NUM> is split into two segments 110a, 110b for each half circuit. Each segment 110a/110b includes one cascode amplifier 120a/120b and one N-path filter 130a/130b.

The N-path filters 130a, 130b may be based on switched capacitors. Each N-path filter 130a, 130b includes N paths coupled in parallel, N being a positive integer greater than one. In this example, each N-path filter 130a, 130b includes four paths. Each path of an N-path filter 130a, 130b may include a switch <NUM> and a capacitor <NUM> (e.g., a variable capacitor) coupled in series. The plurality of N-path filters 130a, 130b are driven by LO signals having different frequencies. The LO signals driving the four switches <NUM> of each N-path filter 130a, 130b may have a <NUM> % duty cycle and are <NUM> degree out-of-phase, i.e., non-overlapping.

A cascode amplifier 120a, 120b may include a common source device <NUM> feeding into a common gate device <NUM>. Each N-path filter is coupled to a different one of the plurality of cascode amplifiers. In this example, the N-path filter 130a is coupled to the cascode amplifier 120a, and the N-path filter 130b is coupled to the cascode amplifier 120b. A single bandpass N-path filter is synthesized by placing an N-path filter 130a, 130b at the source of the common gate device <NUM> of a cascode amplifier 120a, 120b.

The LO signals may be generated by a frequency synthesizer <NUM> (e.g., a DTC-based frequency synthesizer) and supplied to the N-path filters 130a, 130b. A DTC is a device that can control time delay of an input signal by a digital code. The DTC outputs a delayed replica of an RF clock signal based on a control signal. The frequencies of the LO signals that drive the N-path filters 130a, 130b may be symmetrically spaced around the desired center frequency (fLO). For example, the N-path filter 130a may be driven by an LO signal at fLO-Δf and the N-path filter 130b may be driven by an LO signal at fLO-Δf, where fLO is the desired center frequency and Δf is a frequency offset. The input signal may be a differential signal (vin and vip). Outputs of the plurality of cascode amplifiers 120a, 120b are combined and may then be down-converted by an N-path mixer <NUM>.

Each N-path filter 130a, 130b is centered at a different LO frequency. These bandpass responses are combined with appropriate phase to realize an effective bandpass filter with higher bandwidth and steeper roll-off than the individual bandpass responses. The combination with different phases (<NUM>° and <NUM>°) may be done by using the two differential inputs of the LNA. One polarity (vin) of the differential input signal is fed to one cascode amplifier 110a and the other polarity (vip) of the differential input signal is fed to the other cascode amplifier 110b, and the output of the band-pass filters (vbp) may be then down-converted by the N-path mixer <NUM>.

In the example shown in <FIG>, two N-path filters 130a, 130b are driven by LO signals at fLO - Δf and fLO + Δf to synthesize a bandpass filter with <NUM> dB/decade high-frequency roll-off centered at fLO and the output is down converted by an N-path mixer at fLO. Consider the circuit in <FIG> where the N-path filters 130a, 130b with bandwidths equal to f<NUM> at center frequencies fLO + Δf and fLO - Δf are summed anti-phase. This anti-phase summing may be done in differential signaling. The effective bandpass response of the combination of the N-path filters 130a, 130b in <FIG> is given by: <MAT> where ω̃ and <MAT> are defined as, <MAT> <MAT><MAT> is a bandpass response centered at ωLO and shows a <NUM> dB/decade high-frequency roll-off. The bandwidth of <MAT> may be tuned by simply tuning the frequency at which the N-path filters 130a, 130b are driven, and this can be precisely controlled by the frequency synthesizer <NUM> (e.g., a DTC-based frequency synthesizer) with minimal need for capacitance tuning. A controller <NUM> may control the LO frequency (e.g., the LO frequency offset) for tuning the bandwidth. Tuning Δω changes the gain as well. In some examples, to keep the gain constant, the bandwidth ω<NUM> may be scaled. For example, the scaling may be achieved by scaling the baseband capacitors (the capacitors <NUM> of the N-path filters 130a, 130b) by the controller <NUM>.

<FIG> and <FIG> show an example LNA <NUM> for a <NUM>-frequency N-path bandpass filter according to the invention. <FIG> shows a half of the LNA <NUM> and <FIG> shows a full differential circuit of the LNA <NUM>. Each half circuit 200a, 200b generates one polarity of the differential output. The structure of both halves 200a, 200b is the same but the polarity of the input signals to the cascode amplifiers 220a, 220b, 220c, 220d and the LO signals driving the N-path filters 230a, 230b, 230c, 230d of each half 200a, 200b are inversed. Hereafter, the example will be explained with reference to the half circuit in <FIG> for simplicity.

The example LNA <NUM> includes four N-path filters driven by fLO - Δf<NUM>, fLO - Δf<NUM>, fLO + Δf<NUM>, and fLO + Δf<NUM> to synthesize a bandpass filter (e.g., with <NUM> dB/decade high-frequency roll-off centered at fLO). In this example, the LNA <NUM> includes four cascode amplifiers 220a, 220b, 220c, 220d and four N-path filters 230a, 230b, 230c, 230d for each half circuit. The LNA <NUM> is split into four segments 210a, 210b, 210c, 210d for each half circuit. Each segment 210a, 210b, 210c, 210d includes one cascode amplifier 220a, 220b, 220c, 220d and one N-path filter 230a, 230b, 230c, 230d.

Each N-path filter 230a, 230b, 230c, 230d includes N paths coupled in parallel, N being a positive integer greater than one. Each path includes a switch <NUM> and a capacitor <NUM> (e.g., a variable capacitor) coupled in series. The plurality of N-path filters 230a, 230b, 230c, 230d are driven by LO signals having different frequencies. The LO signals driving the four switches <NUM> of each N-path filter 230a, 230b, 230c, 230d may have a <NUM> % duty cycle and are <NUM> degree out-of-phase, i.e., non-overlapping.

A cascode amplifier 220a, 220b, 220c, 220d includes a common gate device <NUM> and a common source device <NUM>. Each N-path filter is coupled to a different one of the plurality of cascode amplifiers. The N-path filter 230a is coupled to the cascode amplifier 220a, the N-path filter 230b is coupled to the cascode amplifier 220b, the N-path filter 230c is coupled to the cascode amplifier 220c, and the N-path filter 230d is coupled to the cascode amplifier 220d. A single bandpass N-path filter is synthesized by placing an N-path filter 230a, 230b, 230c, 230d at the source of the common gate device <NUM> of a cascode LNA 220a, 220b, 220c, 220d, respectively.

The plurality of N-path filters 230a, 230b, 230c, 230d are driven by LO signals having different frequencies. The LO signals may be generated by a frequency synthesizer <NUM> (e.g., a DTC-based frequency synthesizer) and supplied to the N-path filters 230a, 230b, 230c, 230d. The frequencies of the LO signals may be symmetrically spaced around the desired center frequency. Two cascode amplifiers may be driven by the positive input of the differential input signal, and the other two cascode amplifiers may be driven by the negative input of the differential input signal. The two cascode amplifiers driven by one polarity input may have N-path filters at fLO - Δf<NUM> and fLO + Δf<NUM> at the source of the common gate device, and the two cascode amplifiers driven by the other polarity input may have N-path filters at fLO - Δf<NUM> and fLO + Δf<NUM> at the source of the common gate device. For example, the N-path filter 230a may be driven by an LO signal at fLO-Δf<NUM>, the N-path filter 230b may be driven by an LO signal at fLO-Δf<NUM>, the N-path filter 230c may be driven by an LO signal at fLO+Δf<NUM>, and the N-path filter 230d may be driven by an LO signal at fLO+Δf<NUM>, where fLO is the desired center frequency and Δf<NUM> and Δf<NUM> are frequency offsets. The four cascode amplifiers' output currents are then summed/combined together and may be down-converted to baseband by an N-path mixer <NUM> at fLO.

Each N-path filter 230a, 230b, 230c, 230d is centered at a different LO frequency. These bandpass responses are then combined with appropriate phase to realize an effective bandpass filter with higher bandwidth and steeper roll-off than the individual bandpass responses. The combination with different phases (<NUM>° and <NUM>°) may be done by using the two differential inputs of the LNA. One polarity (one of vin and vip) of the differential input signal is fed to two cascode amplifiers 210a, 210d and the other polarity (the other of vin and vip) of the differential input signal is fed to the other cascode amplifiers 210b, 210c and the output of the band-pass filters (vbp) is then down-converted to baseband by the N-path mixer <NUM>.

Two <NUM>-frequency N-path filters centered at different offset frequencies may be combined in an anti-phase fashion to obtain an effective <NUM>-frequency N-path filter as shown in <FIG>. The effective response of the <NUM>-frequency N-path filter-based bandpass filter is given by: <MAT>.

The transfer function <MAT> has a <NUM> dB/decade high frequency roll-off. The signal at node vbp may be down converted to baseband by an N-path mixer <NUM> at fLO, which further increases the order of bandpass filtering at node vbp, with an <NUM> dB/decade high-frequency roll-off. For the specific case where the bandwidth of this N-path mixers is exactly equal to f<NUM>, the transfer function to the node vbp is given by: <MAT>.

The bandwidth of <MAT> and <MAT> may be tuned by simply tuning the frequency at which the N-path filters 230a, 230b, 230c, 230d are driven, and this can be precisely controlled by the frequency synthesizer <NUM> (e.g., a DTC-based frequency synthesizer) with minimal need for capacitance tuning. The controller <NUM> may control the LO frequency (e.g., the LO frequency offset) for tuning the bandwidth. In some examples, to keep the gain constant, the bandwidth ω<NUM> may be scaled. The bandwidth scaling may be achieved by scaling the baseband capacitor (i.e., the capacitors <NUM> in the N-path filters 230a, 230b, 230c, 230d) by the controller <NUM>.

It should be noted that while <FIG>, <FIG>, <FIG>, and <FIG> illustrate the specific cases of combining two and four N-path filters, respectively, the examples can be extended to a linear combination of any number of N-path filters, for example <NUM>k N-path filters, k being a positive integer.

<FIG> shows simulation results of bandwidth tuning in the <NUM>-frequency N-path filter-based digital LNA as shown in <FIG>/<FIG> by tuning the offset frequencies (Δf<NUM> and Δf<NUM>) of the N-path filters alone, with no tuning of the bandwidth of each N-path filter. The simulation was run for fLO = <NUM>, a bandwidth of <NUM> for the four N-path filters at the source of the common gate device and a bandwidth of <NUM> for the N-path mixer at fLO. Under these conditions, <FIG> shows the response for the following <NUM> cases: a) Δf<NUM> = <NUM>, Δf<NUM> = <NUM>, b) Δf<NUM> = <NUM>, Δf<NUM> = <NUM>, c) Δf<NUM> = <NUM>, Δf<NUM> = <NUM> and d) Δf<NUM> = <NUM>, Δf<NUM> = <NUM>, yielding effective bandwidths of <NUM>, <NUM>, <NUM>, and <NUM>, respectively.

The bandwidth tuning for the <NUM>-frequency N-path filter in <FIG> is implemented by tuning the LO frequencies alone, while keeping the baseband capacitance constant. This results in bandwidth tuning with change in gain across different settings. The gain in the response corresponds to the gain of the passive N-path filter alone, and a higher gain may be achieved by the LNA. As shown in <FIG>, there is a variation of as much as <NUM> dB gain between the different settings. In this method of tuning the bandwidth by tuning the LO frequency of the different N-path filters alone, the amount of bandwidth tuning is limited by the amount of tolerable gain variation, as illustrated by the plot in <FIG> shows a plot illustrating the range of bandwidth tuning by tuning LO frequencies of N-path filters alone for different values of maximum tolerable gain variation. The higher the allowable gain variation, the higher the range of bandwidth tuning.

In addition to the LO frequency tuning (e.g., by adjusting the LO frequency offset Δf or Δf<NUM> and Δf<NUM>) as described above, a capacitor tuning (e.g., by adjusting the capacitance of the capacitors <NUM> or <NUM>) may be implemented to change the bandwidth of each N-path filter while maintaining the same gain and filter shape as the effective bandwidth is tuned. It should be noted that there will be no limitation in the range of bandwidth tuning when a hybrid approach of LO frequency and capacitor tuning is used.

<FIG> shows simulation results for bandwidth tuning of a <NUM>-frequency N-path filter as shown in <FIG>/<FIG> by tuning the LO frequencies and baseband capacitance. This results in bandwidth tuning while maintaining the same gain and filter shape. The gain in the response corresponds to the gain of the passive N-path filter alone, and a higher gain may be achieved by the LNA.

In <FIG>, the different plots indicate the bandpass response for fLO = <NUM> for the following conditions: a) Δf<NUM> = <NUM>, Δf<NUM> = <NUM>, f<NUM> = <NUM>, b) Δf<NUM> = <NUM>, Δf<NUM> = <NUM>, f<NUM> = <NUM>, c) Δf<NUM> = <NUM>, Δf<NUM> = <NUM>, f<NUM> = <NUM> and a) Δf<NUM> = <NUM>, Δf<NUM> = <NUM>, f<NUM> = <NUM>, yielding effective bandwidths of <NUM>, <NUM>, <NUM>, and <NUM>, respectively. <FIG> shows the simulated down-converted baseband response for the bandpass responses of <FIG>. In <FIG>, the <NUM> dB/decade (<NUM> dB/octave) high frequency roll-off is observed.

<FIG> illustrates a user device <NUM> in which the examples disclosed herein may be implemented. For example, the examples disclosed herein may be implemented in the radio front-end module <NUM>, in the baseband module <NUM>, etc. The user device <NUM> may be a mobile device in some aspects and includes an application processor <NUM>, baseband processor <NUM> (also referred to as a baseband module), radio front end module (RFEM) <NUM>, memory <NUM>, connectivity module <NUM>, near field communication (NFC) controller <NUM>, audio driver <NUM>, camera driver <NUM>, touch screen <NUM>, display driver <NUM>, sensors <NUM>, removable memory <NUM>, power management integrated circuit (PMIC) <NUM> and smart battery <NUM>.

In some aspects, application processor <NUM> may include, for example, one or more CPU cores and one or more of cache memory, low drop-out voltage regulators (LDOs), interrupt controllers, serial interfaces such as serial peripheral interface (SPI), inter-integrated circuit (I2C) or universal programmable serial interface module, real time clock (RTC), timer-counters including interval and watchdog timers, general purpose input-output (IO), memory card controllers such as secure digital / multi-media card (SD/MMC) or similar, universal serial bus (USB) interfaces, mobile industry processor interface (MIPI) interfaces and Joint Test Access Group (JTAG) test access ports.

In some aspects, baseband module <NUM> may be implemented, for example, as a solder-down substrate including one or more integrated circuits, a single packaged integrated circuit soldered to a main circuit board, and/or a multi-chip module containing two or more integrated circuits.

<FIG> illustrates a base station or infrastructure equipment radio head <NUM> in which the examples disclosed herein may be implemented. For example, the examples disclosed herein may be implemented in the radio front-end module <NUM>, in the baseband module <NUM>, etc. The base station radio head <NUM> may include one or more of application processor <NUM>, baseband modules <NUM>, one or more radio front end modules <NUM>, memory <NUM>, power management circuitry <NUM>, power tee circuitry <NUM>, network controller <NUM>, network interface connector <NUM>, satellite navigation receiver module <NUM>, and user interface <NUM>.

In some aspects, application processor <NUM> may include one or more CPU cores and one or more of cache memory, low drop-out voltage regulators (LDOs), interrupt controllers, serial interfaces such as SPI, I2C or universal programmable serial interface module, real time clock (RTC), timer-counters including interval and watchdog timers, general purpose IO, memory card controllers such as SD/MMC or similar, USB interfaces, MIPI interfaces and Joint Test Access Group (JTAG) test access ports.

In some aspects, baseband processor <NUM> may be implemented, for example, as a solder-down substrate including one or more integrated circuits, a single packaged integrated circuit soldered to a main circuit board or a multi-chip module containing two or more integrated circuits.

In some aspects, memory <NUM> may include one or more of volatile memory including dynamic random access memory (DRAM) and/or synchronous dynamic random access memory (SDRAM), and nonvolatile memory (NVM) including high-speed electrically erasable memory (commonly referred to as Flash memory), phase change random access memory (PRAM), magneto resistive random access memory (MRAM) and/or a three-dimensional crosspoint memory. Memory <NUM> may be implemented as one or more of solder down packaged integrated circuits, socketed memory modules and plug-in memory cards.

In some aspects, power management integrated circuitry <NUM> may include one or more of voltage regulators, surge protectors, power alarm detection circuitry and one or more backup power sources such as a battery or capacitor. Power alarm detection circuitry may detect one or more of brown out (under-voltage) and surge (over-voltage) conditions.

In some aspects, power tee circuitry <NUM> may provide for electrical power drawn from a network cable to provide both power supply and data connectivity to the base station radio head <NUM> using a single cable.

In some aspects, network controller <NUM> may provide connectivity to a network using a standard network interface protocol such as Ethernet. Network connectivity may be provided using a physical connection which is one of electrical (commonly referred to as copper interconnect), optical or wireless.

In some aspects, satellite navigation receiver module <NUM> may include circuitry to receive and decode signals transmitted by one or more navigation satellite constellations such as the global positioning system (GPS), Globalnaya Navigatsionnaya Sputnikovaya Sistema (GLONASS), Galileo and/or BeiDou. The receiver <NUM> may provide data to application processor <NUM> which may include one or more of position data or time data. Application processor <NUM> may use time data to synchronize operations with other radio base stations.

In some aspects, user interface <NUM> may include one or more of physical or virtual buttons, such as a reset button, one or more indicators such as light emitting diodes (LEDs) and a display screen.

Functions of various elements shown in the figures, including any functional blocks labeled as "means", "means for providing a sensor signal", "means for generating a transmit signal. ", etc., may be implemented in the form of dedicated hardware, such as "a signal provider", "a signal processing unit", "a processor", "a controller", etc. as well as hardware capable of executing software in association with appropriate software. When provided by a processor, the functions may be provided by a single dedicated processor, by a single shared processor, or by a plurality of individual processors, some of which or all of which may be shared. However, the term "processor" or "controller" is by far not limited to hardware exclusively capable of executing software but may include digital signal processor (DSP) hardware, network processor, application specific integrated circuit (ASIC), field programmable gate array (FPGA), read only memory (ROM) for storing software, random access memory (RAM), and non-volatile storage.

Claim 1:
A low-noise amplifier, LNA, comprising:
a plurality of N-path filters (130a/130b, 230a/230b/230c/230d); and
a plurality of cascode amplifiers (120a/120b, 220a/220b/220c/220d) configured to amplify an input signal, wherein each N-path filter (130a/130b, 230a/230b/230c/230d) is coupled to a different one of the plurality of cascode amplifiers (120a/120b, 220a/220b/220c/220d) and each cascode amplifier comprises an amplifier transistor for receiving a respective input signal of a differential input signal and an associated cascode transistor, and the plurality of N-path filters are connected in parallel with the amplifying transistor at a connection node of the amplifying transistor and the cascode transistor,
wherein the plurality of N-path filters (130a/130b, 230a/230b/230c/230d) are driven by local oscillator, LO, signals having different frequencies, and the plurality of cascode amplifiers (120a/120b, 220a/220b/220c/220d) are coupled in parallel at an output node of the LNA,
wherein one pair of N-path filters (230b, 230c) coupled to one polarity of a differential input signal are driven by LO signals at fLO+Δf<NUM> and fLO-Δf<NUM>, and another pair of N-path filters (230a, 230d) coupled to the other polarity of the differential input signal are driven by LO signals at fLO+Δf<NUM> and fLO-Δf<NUM>.