Patent Description:
One or more embodiments can be applied to driving bypass field-effect transistors in high-voltage startup regulators as used in a variety of products.

These products may include, for instance, chargers/adapters, home appliances such as refrigerators or TVs, industrial machinery, telecom apparatus and so on.

Regulator circuits such as high-voltage startup (HVS) regulators may include a bypass transistor (a MOSFET transistor, for instance), a pull-up resistor used to bias the gate of the bypass transistor, as well as a voltage clamp connected between the gate of the bypass transistor and ground in order to limit the gate voltage range and define a maximum supply voltage output.

Such an approach may exhibit certain limitations in terms of power consumption, application input range, and cost.

For instance, in low-power applications, a constraint may be placed on the lowest (minimum) resistance value for the pull-up resistor.

Under those circumstances, achieving a trade-off between such a constraint on the minimum resistance value for the pull-up resistor and the lowest (minimum) high-voltage input may become critical, while on the other hand a bypass component with a gate bias current still represents an attractive, cost-effective solution.

Document <CIT> discloses a power converter startup circuit that establishes an operating voltage for control circuitry during startup and is then disabled to reduce no-load power dissipation. The startup circuit has a normally on characteristic to automatically provide startup charging current for a startup capacitor. The control circuitry begins operating as the startup capacitor voltage reaches an operating value, and it generates an inhibitory signal that disables the startup circuit to stop the startup charging current and reduce power dissipation. The normally on characteristic is achieved by an emitter switched current source employing a normally on device such as a depletion-mode J-FET. A resistor divider network provides both biasing for the startup current source and a point of monitoring the power supply input voltage during steady state operation. A charge pump provides a voltage of the inhibitory control signal sufficiently above the auxiliary voltage to turn off the normally on switching transistor.

Also documents <CIT>, <CIT> and <CIT> are of interest for the invention.

An object of one or more embodiments is to contribute in addressing the issues discussed in the foregoing.

According to one or more embodiments, such an object can be achieved by means of a circuit having the features set forth in claim <NUM> that follows.

One or more embodiments relate to a corresponding system as claimed in claim <NUM>. A switched-mode power supply (SMPS) system may be exemplary of such a system.

One or more embodiments relate to a corresponding method as claimed in claim <NUM>.

The claims are an integral part of the technical teaching on the embodiments as provided herein.

One or more embodiments may provide a cost-effective solution, primarily in comparison with the use of a dedicated HV-MOS with no gate leakage.

One or more embodiments may use a charge pump to sustain the gate leakage current. Such a charge pump can be supplied by a transistor source and its activation may thus involve pulling-up the gate of the bypass transistor.

In one or more embodiments, an (electronic) switch may be interposed between the gate and the pull-up resistor.

The switch can be closed (that is, made conductive) by a latched comparator in response to the input voltage reaching (that is, rising up to) a comparator threshold.

The charge stored in a capacitor intermediate (between) the pull-up resistor (RHV) and ground facilitates pulling-up the gate with the capability of providing both supply energy as consumed by the comparator/latch and supporting gate leakage, before charge pump turn-on.

One or more embodiments can effectively address certain limitations in terms of power consumption, application input range, and cost.

The possible presence of an embedded charge pump may result in a negative current detected on a high-voltage pin when the high voltage startup regulator is active.

In the ensuing description one or more specific details are illustrated, aimed at providing an in-depth understanding of examples of embodiments of this description.

By way of introduction to the instant detailed description, reference may be had to <FIG>, which reproduces a circuit diagram of a conventional high-voltage startup (HVS) regulator designated <NUM> as a whole.

Such a regulator is intended to provide a low voltage supply to a low-voltage section LV from a high voltage source VHVIN. It is noted that in various types of high-voltage power converters, such a high-voltage startup regulator may be used only during a power-on phase to be subsequently replaced by a more efficient supply source.

A high-voltage startup regulator <NUM> as illustrated in <FIG> comprises a "bypass" electronic switch MBP (a MOSFET transistor, for instance) able to withstand a high voltage applied across it.

In that respect, those of skill in the art will appreciate that a MOSFET transistor is referred to herein as the switch MBP merely by way of example: other types of electronic switches (for instance, JFET, BJT, GaN) may be used as the switch MBP.

As illustrated in <FIG>, the switch MBP is arranged with the current path therethrough (source-drain, in the case of a field-effect transistor such as a MOSFET) coupled intermediate an input node to which a high voltage VHVIN can be applied and the low-voltage section LV, which can be assumed to be referred to ground GND.

As illustrated in <FIG>, the drain D of the MOSFET transistor MBP is connected to the high-voltage input node at VHVIN while the source S of the MOSFET transistor MBP supplies the low-voltage circuitry LV.

A pull-up resistor RHV is used to bias the control node G (gate, in the case of a field-effect transistor such as a MOSFET) of the bypass switch MBP.

As illustrated, the pull-up resistor RHV is arranged between the high-voltage input node at VHVIN and the gate G of MBP.

A voltage clamp DZ (a zener diode, for instance) is arranged between the gate G of MBP and ground GND in order to limit the gate voltage range of MBP and define a highest (maximum) supply voltage for the low-voltage section LV.

In order to reach this voltage value, the input voltage is selected above the DZ voltage clamp.

In low-power applications, a constraint on the minimum value of the resistance of RHV is set in order to limit waste of power. Another specification is the lowest (minimum) input voltage (VHVINMIN) for which the high-voltage regulator is expected to be capable of supplying the downstream circuits.

Assuming MBP is a component having a gate bias current (IGATE) a certain voltage drop on RHV may exist which establishes a judicious trade-off between the constraint on the minimum RHV and the minimum high-voltage input.

A problem arises when this trade-off cannot be satisfactorily reached.

For instance, with RHV=10MQ and IGATE =10µA, the minimum input voltage is above 100V.

In principle, using a lower resistance value for RHV may be considered, to the point of even dispensing with such resistance. It is otherwise noted that the minimum value for that resistance is related to a desirably reduced stand-by consumption, which results in a design constraint.

Also, using a component "with leakage" may be an attractive option in monolithic or System in Package (SiP) arrangements: for instance, a high-voltage component with non-zero gate current may represent the only component available in a certain technological process and/or may represent an advanteous choice in view of specific design characteristics.

For instance, gallium-nitride (GaN) technology may be selected for the main switch in a converter in view of its improved static and dynamic performance in comparison with silicon-based power MOS transistors. Adding a dedicated component (a SIP with a dedicated die and/or process) may be more expensive and complex.

Using a bypass component MBP with a gate bias current IGATE forced to flow through RHV and then into the gate of MBP may provide a solution which is cost-effective and attractive (for instance, resulting in a monolithic solution with only one component available). Such an approach may otherwise exhibit various drawbacks: for instance, it may be hardly suited for use in very-low standby applications and/or may end up by providing undesirably expensive and complex solutions.

One or more embodiments may address these issues adopting the approach illustrated in <FIG>.

In that respect, it will be appreciated that:.

In one or more embodiments as illustrated in <FIG>, circuitry designated <NUM> as a whole is coupled intermediate:.

As illustrated, the circuitry <NUM> comprises an electronic switch SW1 (a MOSFET transistor, for instance) with the current path therethrough (source-drain, in the case of a field-effect transistor such as a MOSFET) configured to connect - in response to the switch SW1 being "on", that is, conductive - the control node G of the "main" switch MBP to the node HV, and thus to the pull-up resistor RHV.

As illustrated, the circuitry <NUM> also comprises:.

<FIG> are exemplary of possible operation of a regulator <NUM> as exemplified in <FIG>.

Once again, unless the context indicates otherwise, parts or elements already discussed in connection with <FIG> are indicated in <FIG> with like reference symbols and will not be not described again for brevity.

More specifically, <FIG> are exemplary of possible current flows in a regulator <NUM> as per the embodiments exemplified in <FIG>:.

At power-on, SW1 is assumed to be open (nonconductive) and no current IGATE can flow through RHV towards the node G and the voltage at the node HV follows VHVIN.

In response to the voltage at the node HV reaching the threshold VTH of the comparator <NUM> (<NUM> V, by way of example), the related information is stored by the latch <NUM> and SW1 is closed (that is, made conductive) via a signal SW1_ON being asserted.

In this phase the charging of the capacitor CHV is used to pull-up the gate G of the switch MBP (exploiting the intrinsic gate-source capacitance, for instance) and a current IGATE is provided towards the node G.

When the voltage on the source S of MBP is enough (that is, it exceeds a lower threshold as desired), the charge pump <NUM> starts to supply current towards the node HV, drawing current from the source S of MBP (see <FIG>).

The charge pump <NUM> thus provides the current IGATE and a current in excess of IGATE used to charge the capacitor CHV up to the voltage clamp of HV.

That is, during high-voltage regulator operation, the resistor RHV does not contribute appreciably to supplying the current IGATE and the voltage drop across RHV does not limit the lowest (minimum) input voltage, which is independent both of the resistance value of RHV value and of the intensity of IGATE.

<FIG> is a transistor-level diagram exemplary of a of a possible embodiment of the layout illustrated in <FIG> based on a latched current comparator <NUM>, which is configured to facilitate having a current IHVtrig through the resistor RHV such that IHVtrig <<< IGATE.

Once again, unless the context indicates otherwise, parts or elements already discussed in connection with <FIG> and <FIG> are indicated in <FIG> with like reference symbols and will not be not described again for brevity.

Essentially, the latched comparator <NUM> is based on a current mirror M3 including two transistors Q1 (in a diode-like configuration) and Q2 having current flow paths therethrough (emitter-collector, in the case of bipolar transistors as exemplified herein) that define respective current flow lines:.

The mutually connected bases of the transistors Q1 and Q2 are coupled to the node HV via the current flow path (source-drain, in the case of a MOSFET transistor as exemplified herein) through a further electronic switch M1 whose control node (gate in the case of a MOSFET transistor as exemplified herein) is coupled to the control node of the switch SW1 vie the line where the switching signal SW1_ON is applied with the parallel connection of a further voltage clamp DZUP and further pull-up resistor Rup coupling that line with the line/node HV.

In <FIG>, the main currents before start-up are indicated in continuous lines. Dashes lines denote main currents at startup and chain lines denote the main currents in steady state after activation of the charge pump <NUM>.

The charge pump <NUM> can be of any conventional type, such as Dickson, for instance, as exemplified in <FIG>.

<FIG> comprises a set time diagrams exemplary of possible time behaviors, mapped against a common time (abscissa) scale of the following signals:.

The diagrams of <FIG> are exemplary of a possible start-up sequence including events designated with numbers from <NUM> to <NUM> such as:.

<FIG> is a circuit diagram exemplary of the possible use of circuit <NUM> as discussed in the foregoing within the framework of a switched-mode power supply (that is an electronic power supply system that incorporates a switching regulator to convert electrical power efficiently).

Once again, unless the context indicates otherwise, parts or elements already discussed in connection with the previous figures are indicated in <FIG> with like reference symbols and will not be not described again for brevity.

<FIG> illustrates the possible use of embodiments in connection with a system in package (SIP) switched-mode power supply controller genarelly designated SMPS.

<FIG> exemplifeis the possible use of embodiments in switching power controllers (with a flyback topology in the - purely exemplary - case illustrated) integrating a main power switch PM (a power MOSFET transistor, for instance), with part of the circuit <NUM> included in the controller (for instance the clamp DZ, the bypass FET driver <NUM>, and the switch MBP) while other components (such as the resistor RHV) may be external parts distinct from the circuit <NUM> (and <NUM>).

For instance, this may be the case of the source of the high-voltage VHVIN, represented in <FIG> by a bridge rectifier BR supplied by an AC source ACin (a mains distribution grid, for instance) and having associated therewith a smoothing capacitor Cin intended to be coupled with the line or node VHVIN.

In an arrangement as exemplified in <FIG>, in the place of being coupled to VHVIN directly as discussed previously, the switch MBP is coupled to VHVIN (at its drain) indirectly, that is, via the primary winding of the converter transformer T.

In an arrangement as exemplified in <FIG>, the low- voltage circuitry LV coupled to the switch MBP (here at the source) includes the supply portion og the SMPS controller, including a supply node VDD.

The external network coupled to the node VDD is exemplified as a winding driving a rectifier diode and a smoothing capacitor Cvdd (not to be confused with the capacitor CHV discussed previously), provides an auxiliary power supply active during regulation. A current generator Icharge is illustrated as exemplary of the node VDD being charged with a controlled current. The current generator could be replaced with a diode exemplary of the fact that the auxiliary power supply and the power supply from the HVS circuit are mutually decoupled.

The symbol PM in <FIG> denotes a main switch of the switching converter. Advantageously, this may be of the same type of switch MBP (a GaN transistor, for instance).

The reference <NUM> in <FIG> denotes the logic control circuitry of the switch PM, which applies to the switch a PWM-modulated control signal via a driver <NUM>.

This may occur in any conventional manner known to those of skill in the art. It will be otherwise appreciated that <FIG> provides a fairly general representation of a switching controller: reference to a SMPS converter controller as made in <FIG> is thus merely exemplary and not limiting of the embodiments.

One or more embodiments may in fact be applied to flyback, boost topologies and other types of bypass regulators as a high-voltage startup current generator.

One or more embodiments may provide various advantages such as, for instance:.

Briefly, one or more embodiments remove a limitation of conventional high-voltage startup regulators employing as a bypass component an electronic switch having a (high) bias current.

In that respect, it is again noted that, while a MOSFET transistor has been referred throughout as exemplary of the switch MBP, other types of electronic switches (for instance, JFET, BJT, GaN) may be used for the same purposes.

Such a limitation is related to the voltage drop on a pull-up resistor (RHV, for instance) having one end used to bias the bypass switch gate such a resistor and other end connected at the main input voltage.

One or more embodiments use a charge pump in order sustain the gate leakage current.

Such a charge pump can be supplied by the transistor source and its activation involves pulling-up the gate of the bypass switch MBP.

In one or more embodiments an electronic switch SW1 is interposed between the control node (gate, for instance) of the bypass switch MBP gate and the pull-up resistor. The switch is closed (by a latched comparator, for instance) as soon as the input voltage reaches a comparator threshold.

The charge stored in a capacitor (CHV, for instance) connected between RHV and GND facilitates gate pull-up with the capability of supplying (and sustaining) both the consumption of the comparator/latch consumption and gate leakage, before the charge pump turning-on.

Briefly, a circuit (for instance, <NUM>) as claimed herein comprises:
an electronic switch (for instance, MBP) having a current flow path (for instance, S, D) therethrough, the electronic switch configured to be coupled intermediate a high-voltage node (for instance, VHVIN) and low-voltage circuitry (for instance, LV), the electronic switch having a control node (for instance, G) configured to switch the electronic switch to a conductive state wherein the low-voltage circuitry is coupled to the high-voltage node.

A circuit as exemplified herein may further comprise a voltage-sensing node (for instance, HV) configured to be coupled to the high-voltage node via a pull-up resistor (for instance, RHV),.

A circuit as exemplified herein may comprise a charge capacitor (for instance, CHV) coupled (for instance, via HV) to the charge pump to be charged thereby via charge in excess of the charge pumped to the control node of the electronic switch.

In a circuit as exemplified herein, the charge pump is coupled to the voltage-sensing node and configured to pump therein charge sourced from the current flow-path of the electronic switch.

A circuit as exemplified herein may comprise a latch circuit (for instance, <NUM>) intermediate the comparator and the further electronic switch to latch therein said switch-on signal in response to the voltage at said voltage-sensing node reaching said threshold.

In a circuit as exemplified herein, the voltage-sensing node is configured to be coupled to the high-voltage node intermediate said pull-up resistor (for instance, RHV) and a voltage clamp (for instance, DZ) referred to ground (for instance, GND) configured to clamp to a limit value the voltage at the voltage-sensing node.

In a circuit as exemplified herein, the electronic switch (for instance, MBP) has a first node (for instance, D) configured to be coupled to the high-voltage node (for instance, VHVIN) and a second node (for instance, S) configured to be coupled to the low-voltage circuitry (for instance, LV), and wherein the charge pump is coupled to the current flow-path of the electronic switch at the second node (for instance, S).

In a circuit as exemplified herein, the electronic switch and the further electronic switch may comprise field-effect transistors, optionally MOSFET transistors.

A power supply system as exemplified herein comprises:.

A method as exemplified herein comprises supplying low-voltage circuitry (for instance, LV) a from a high-voltage source (for instance, BR, Cin) by:.

Claim 1:
A circuit (<NUM>), comprising:
an electronic switch (MBP) having a current flow path (S, D) therethrough, the electronic switch (MBP) configured to be coupled intermediate a high-voltage node (VHVIN) and low-voltage circuitry (LV), the electronic switch (MBP) having a control node (G) configured to switch the electronic switch (MBP) to a conductive state wherein the low-voltage circuitry (LV) is coupled to the high-voltage node (VHVIN),
a voltage-sensing node (HV) configured to be coupled to the high-voltage node (VHVIN) via a pull-up resistor (RHV),
a further electronic switch (SW1) intermediate the voltage-sensing node (HV) and the control node (G) of the electronic switch (MBP), the further electronic switch (SW1) switchable to a conductive state to couple the voltage-sensing node (HV) and the control node (G) of the electronic switch (MBP) in response to a switch-on signal (SW1_ON) being asserted,
a comparator (<NUM>) coupled to the voltage-sensing node (HV) and a threshold (VTH), the comparator (<NUM>) configured to compare a voltage at said voltage-sensing node (HV) with the threshold (VTH) and cause (<NUM>) the switch-on signal (SW1_ON) to be asserted in response to the voltage at said voltage-sensing node (HV) reaching said threshold (VTH), and
a charge pump (<NUM>) coupled to the current flow-path of the electronic switch (MBP) and configured to be activated with the further electronic switch (SW1) switched to the conductive state to pump electric charge (IGATE) from the current flow-path of the electronic switch (MBP) to the control node (G) of the electronic switch (MBP) via the further electronic switch (SW1) switched to the conductive state.