Patent Description:
An isolated transistor can be used as a control element in many circuit applications. For example, isolated transistors can be used in power electronics for switches, multiplexers (muxes), circuit breakers, ideal diodes, current or power limiters, hot swap functions, battery charger applications, etc. It would be desirable to have a measurement technique for controlling an isolated transistor that operates independent of the common-mode voltage of the application.

<CIT> discloses an integrated circuit (IC) for sensing a current flowing through a transistor device including a substrate and a current scaling circuit that includes first and second MOSFET devices.

<CIT> discloses integrated protection devices with monitoring of electrical characteristics. A switching mechanism is configured to have an on state or an off state, where the on state allows current to pass along a current path. A monitoring mechanism has one or more sensing inputs coupled to sense an electrical characteristic at the current path.

<CIT> discloses a switch device provided with a MOSFET (field effect transistor) connected between an input and an output, and an overcurrent detecting circuit for detecting the overcurrent when the current flowing in the MOSFET exceeds a predetermined value.

<CIT> discloses a system of providing a self-adjusting reference current for floating supply stages.

An isolated transistor can be used as a control element in integrated circuits. In a specific example, electronic circuit breaker (ECB) circuits can be used to connect a supply to a load if the current is below a pre-determined threshold, and to disconnect the supply from that load if the current is above a pre-determined threshold.

<FIG> is a block diagram of an example of an ECB circuit using ideal circuit elements. The ECB circuit includes a sense resistor RSNS, a switch SW1, a buffer and a comparator. Switch SW1 is ON or closed when ILOAD < VREF/RSNS and SW1 is OFF or open when ILOAD > VREF/RSNS. A practical implementation of the ECB circuit would replace switch SW1 with a transistor. However, consider an ECB where the input voltage VIN could be a high positive voltage, a voltage near ground, or a negative voltage. It would be desirable to have a measurement technique for an ECB that operates independent of the common-mode voltage of the switched current path (from IN to OUT in <FIG>). Furthermore, it would be advantageous to eliminate the voltage drop and power dissipation associated with the current sense resistor RSNS.

<FIG> is a circuit diagram of another example of an ECB circuit. The circuit performs a circuit-breaker function as in the example shown in <FIG> but switch SW1 between the input node IN and the output node OUT is replaced by a field effect transistor (FET) MPWR. The current sense resistor RSNS, buffer, and comparator in <FIG> are replaced with different monitoring circuitry in <FIG>. Transistor MPWR, switches X1, X2, X3, X4, and capacitors C1, C2 of the monitoring circuitry are located in a different voltage domain than the rest of the monitoring circuitry. In the example of <FIG>, transistor MPWR, switches X1, X2, X3, X4, and capacitors C1, C2 of the monitoring circuitry are located in a higher voltage (HV) region of the circuit, and the rest of the monitoring circuitry is located in a lower voltage (LV) region of the circuit. The HV domain and the LV domain are separated by an isolation structure.

Shown in the lower right corner of <FIG> is a timing diagram for the three phases of the switched capacitor circuitry. During the first phase, Phi1, the input common-mode of the output voltage VOUT is sampled on both of isolation capacitors C1 and C2. Simultaneously, all amplifiers (G1, G2, and A3) are auto-zeroed. The gained offset voltages of G1, G2 and A3 are summed and stored on auto zero (AZ) capacitors CAZ3 and CAZ4. The "output" common-mode of the AZ capacitors is set to common mode voltage VCM using switches X12 and X13. It should be noted that the combined gain of G1 and G2 into resistors R1, R2 and amplifier A3 is relatively small (e.g., gain is in a range of <NUM> - <NUM>, depending on expected input signal range). Capacitors C1, C2 and CAZ3, CAZ4 are matched capacitor pairs.

<FIG> also includes reference transistor MREF and a regenerator circuit. During the second phase, Phi2, the voltage on C2 does not change, but the Vds of MPWR is differentially presented to the inputs of G1 via C1 and C2. Simultaneously, the drain-to-source voltage (Vds) of MREF is connected to the inputs of G2. In an integrated circuit implementation, MPWR can be designed to be a multiple, α, of many MREF devices in parallel (e.g., α = <NUM>,<NUM>). A reference current (IREF) is forced through reference transistor MREF. To a first order, this ensures that the Vds of MREF equals the Vds of MPWR when the current of MPWR is α times IREF independent of temperature and process variations. In addition, G1 and G2 are designed to be matching transconductance amplifiers with equal transconductance.

The regenerator circuit includes a comparator circuit and circuitry that reduces noise at the output of the regenerator circuit. During Phi2, the gained up Vds of MREF is subtracted from the gained up Vds of MPWR and presented to the inputs of the regenerator circuit via CAZ3 and CAZ4. It should be noted that CAZ3 and CAZ4 previously stored the offset information from the autozeroing action of Phi1 providing high accuracy even when the Vds being measured is very small.

During the final phase, Phi <NUM>, the regenerator circuit is powered to decide whether the Vds of MPWR is smaller or larger than the Vds of MREF. This comparison decision is latched on the next rising edge of Phi1. Transistors MPWR and MREF are fabricated in a precise ratio (e.g., <NUM>,<NUM> to <NUM>). Because of the ratio between MREF and MPWR, this comparison of the Vds of the two devices is equivalent to comparing the current of MPWR to α times IREF (αIREF). The Vds of MPWR being greater than VDS of MREF indicates that the current in MPWR is higher than the desired trip current level. The output of the latch can then be used to turn transistor MPWR off in the event of a fault.

There are several benefits to the circuit topology of <FIG>. An obvious benefit is that no series sense resistor (RSNS) is required for current measurement. This eliminates the voltage drop and power dissipation associated with a sense resistor. Another primary benefit of this topology is that all circuitry can be implemented using low voltage devices except for MPWR, MREF, C1 and C2 (as well as a few level-shifting capacitors required to drive switches X1, X2, X3 and X4). In other words, if the signal processing circuitry is powered from a low voltage bias supply (e.g., supply VDD), but the input common mode voltage is high (e.g. <NUM> Volts (12V), or 40V, or 60V), then components MPWR, MREF, C1, and C2 need to handle the maximum input voltage, but all other circuitry can be low voltage consistent with VDD (e.g., <NUM>. 7V or <NUM>. The use of low voltage devices makes the circuit topology easily adaptable to many different voltage levels by simply changing the power device (MPWR), reference device (MREF), and isolation capacitors (C1, C2).

In addition, the circuit topology works equally well for high common-mode input voltages, very low common-mode input voltages (e.g. near 0V) or even negative common-mode voltages. Provided C1 and C2 have no polarity concerns, the level-translation of the Vds of MPWR to the low voltage (VDD) domain will work independent of input/output common-mode.

Another significant benefit of the topology is that this current measuring technique does not perturb the switched current path (IN to OUT) in a meaningful way. The average current drawn from IN or OUT can be well under <NUM> nano-Amperes (100nA) without affecting circuit operation. While operating the switched capacitor circuitry at very high frequency does increase current consumption, this current is provided by VDD and not from IN or OUT. Finally, the architecture can be modified to work for current flowing in either direction.

<FIG> is a circuit diagram of a drive circuit to drive the switches X1, X2, X3, X4, on the HV side of the circuit in <FIG>. The P-channel FET (PFET) M1 corresponds to any of switches X1, X2, X3, or X4 in <FIG>. Capacitor C1 in <FIG> corresponds to capacitor C1 in <FIG>. The drive circuit of <FIG> uses all low voltage devices except for isolation capacitor CDRV which needs to withstand the maximum input/output common-mode voltage. The average current consumption of the drive is determined by VDD and not by the IN or OUT nodes.

<FIG> is a circuit diagram of a drive circuit to drive switch transistor MPWR. Signal MPWRGATE in <FIG> is applied to the gate connection of transistor MPWR. A complication for the circuit topology of the drive circuit for MPWR is that, while the circuit of <FIG> will work with a variety of technologies for MPWR, it is often desirable to use an N-channel metal-oxide semiconductor field effect transistor (MOSFET) due to its high conductivity for MPWR (and MREF). This requires a boost circuit to generate a gate-to-source voltage (Vgs) for MPWR above the output voltage VOUT. Rather than boosting VDD to reach a suitable Vgs above VOUT, a better approach is to use charge pump techniques to stack VDD (or <NUM> times VDD) on top of VOUT to provide Vgs for MPWR.

Another complication for the circuit topology of the drive circuit for MPWR and for MREF is that in the current measurement technique of <FIG>, the Vgs of MREF and the Vgs of MPWR should be substantially the same, but the Vgs of MPWR will be referenced to node OUT while the Vgs of MREF will be referenced to ground. A technique to match the Vgs of MPWR to the Vgs of MREF is to use two regulated charge pumps - one for MPWR and one for MREF. The charge pump for MPWR would stack VDD (e.g., <NUM> times VDD) on top of VOUT while the charge pump for MREF would simply stack VDD (e.g., <NUM> times VDD) on top of ground. With a matching regulation network on each charge pump, the Vgs of each device MPWR, MREF can be the same.

Another technique to match technique Vgs of MPWR to Vgs of MREF is to use a single charge pump to stack VDD (e.g., <NUM> times VDD) on top of VOUT to provide Vgs for MPWR, and then, use a one-to-one (<NUM>:<NUM>) charge pump to level shift that same Vgs back down to ground to drive MREF. Other techniques to match Vgs of MPWR to the Vgs of MREF are possible. Once a voltage sufficient to drive MPWR is available, the technique for driving the circuit of <FIG> can be used to drive MPWR on and off.

<FIG> is a circuit diagram of another example of an ECB circuit. The circuit approach of <FIG> uses another pair of capacitors C3, C4 matched to capacitors C1, C2 to implement a fully differential circuit using the concepts of the example of <FIG>. Switches X1, X2, X3, X4 and X7, X8, X9, X10 act as Vds commutators for MPWR and MREF, respectively. The signal presented to amplifier A3 is doubled with respect to <FIG>, and the amplification stages are slightly simplified by removal of transconductance amplifiers G1 and G2. Shown in the upper right corner of <FIG> is a timing diagram for the two phases of the switched capacitor circuitry of <FIG>.

While the circuits of <FIG> and <FIG> are example of ECB circuits, the front-end signal processing techniques can be used in other applications to provide the advantages described previously herein.

<FIG> is a circuit diagram of an example of an analog current monitor circuit. The signal processing circuits of the example of <FIG> are used to drive a circuit loop to generate an analog current monitor output. The additional circuitry is shown in the box in the upper right of <FIG> and includes a differential to single-ended sample-hold circuit and an error amplifier. The differential to single-ended sample-hold and the error amplifier are used to drive the reference current in MREF so that the Vds of MPWR is substantially equal to the Vds of MREF, or Vds,MPWR = Vds,MREF. The analog monitor circuit includes two P-channel FETs M1 and M2. M2 generates the IREF applied to transistor MREF. PFET M1 matches PFET M2 and provides an analog current signal IMON proportional to the current of MPWR. The current can be converted to an analog voltage proportional to the current of MPWR. The analog voltage can be compared to a specified voltage threshold (e.g., VREF) using a comparator (ECB comparator). The output of the comparator can then be used to turn transistor MPWR off in the event of a fault.

<FIG> is a circuit diagram of an example of a digital current monitor circuit. The digital current monitor circuit digitizes the analog current signal IMON and provides a digital code proportionate to the current of MPWR. The digital current monitor circuit includes an analog-to-digital converter (ADC) circuit to convert the analog current signal IMON to the digital code which can be provided to a digital interface. The ADC circuit may be an integrating ADC, sigma-delta ADC, or a successive approximation register (SAR) ADC. The output of the ECB comparator or the digital code can be used to turn transistor MPWR off in the event of a fault.

The circuit examples described previously herein have included a fixed current threshold to implement the ECB function and the circuit monitor function. While scaling the ratio of the MPWR transistor and the matched MREF transistor using a fixed IREF in the example of <FIG> establishes a fixed electronic circuit breaker threshold, it is possible to make the electronic circuit breaker threshold variable by using a variable IREF. If the IREF current source is implemented as a current digital-to-analog (DAC) circuit, a variable threshold can be implemented by the corresponding variable Vds across the reference device MREF. This variable threshold modifies the trip point of the ECB and by adjusting the digital code controlling the IREF current DAC, the threshold of the ECB can be adjusted real-time in the end application.

The example of <FIG> includes a current monitor with the ECB. In <FIG>, the RIMON1 resistor sets the trip point of the ECB as well as the scaling of the analog output voltage proportional to the current through the MPWR switch. By replacing RIMON1 with an analog or digitally adjustable resistance, it is possible to implement a programmable trip threshold in the current monitor circuit of <FIG>.

Another approach to making the trip threshold of an ECB adjustable is to implement switch segmentation of one or both of the MPWR transistor and the MREF transistor. Segmentation is possible by dividing MPWR into an array of individual transistor devices and providing independent switch control of the independent gates of the individual transistor devices. For example, the segmented MPWR transistor can be composed of sixteen individual devices with the source and drain contacts connected to the IN and OUT nodes of the ECB. The gates of these individual devices can each be driven by a drive circuit shown in the example of <FIG>. The segmented transistors and portions of the drive circuits are included in the HV domain. Using individual drive circuits makes it possible to turn on fractions of MPWR transistor using the ENGATE signal to effectively make the drain-to-source on resistance (Rds-on) of the MPWR programmable. The enabling and disabling of the individual devices provides altering the Rds-on of the MPWR and altering the area ratio of MPWR/MREF which alters the trip threshold.

<FIG> and <FIG> show an example of segmentation of the MPWR transistor. In <FIG>, sixteen individual devices are connected between node IN and OUT as five segments of devices. The five segments are weighted. The first segment and the second segment each include one individually gated transistor device (gate<<NUM>> or gate<<NUM>>). The third segment includes two transistors devices controlled by one gate signal gate<<NUM>>, the fourth segment includes four transistors controlled by one gate signal gate<<NUM>>, and the fifth segment includes eight transistors controlled by one gate signal gate<<NUM>>.

<FIG> shows five instances of the drive circuit of <FIG>. One instance of the drive circuit is used to drive one of the five segments of the MPWR transistor of <FIG>. The switch control of <FIG> and <FIG> is as follows.

The switch control and weighted segmentation of MPWR of <FIG> and <FIG> provides a log-controlled weight of Rds-on of the segmented MPWR transistor to achieve a high dynamic range. It also reduces the number of instances of the drive circuit of <FIG> needed from sixteen if each of the <NUM> devices is individually controlled.

An additional benefit of reducing the area of the MPWR transistor (e.g., by segmenting) is to increase the imposed voltage drop for a given channel current. For smaller current thresholds, the corresponding Vds of MPWR is reduced which reduces the signal level to be compared with Vds of MREF. By engaging a fraction of MPWR, the Vds of MPWR is increased, which proportionately increases the voltage signal level at the input of amplifier G1 in the example of <FIG>. Thus, the accuracy is maintained at lower threshold trip currents. The ECB architecture that uses a combination of a programmable current DAC for IREF and programmable switch segmentation is capable of a much larger operating dynamic range while maintaining accuracy over the operating range. Segmentation of the MREF transistor is possible as well, which may be used as an additional means of adjustment of the trip threshold.

For completeness, <FIG> is a flow diagram of an example of a method <NUM> of controlling operation of an ECB circuit. At <NUM>, a switch transistor disposed in a first voltage domain is activated. The first voltage domain can be the HV domain in the example circuits of the Figures and the switch transistor can be an isolated NFET. The transistor is activated by applying a gate voltage to the gate region of the switch transistor. At <NUM>, a reference transistor is activated by applying a gate voltage to the gate region of the reference transistor. The reference transistor is disposed in a second voltage domain isolated from the first voltage domain. The reference transistor may be of a size that is a predetermined ratio of the switch transistor.

At <NUM>, a current in the reference transistor is monitored to determine when a current of the switch transistor is greater than a specified current. The current may be a fault trip current that is associated with a circuit fault. At <NUM>, the switch transistor is turned off in response to determining that the current in the reference transistor indicates that the current of the switch transistor is greater than the specified current.

According to some aspects, the Vds of the switch transistor and the Vds of the reference transistor are monitored as in the circuit examples of <FIG> and <FIG>. A reference current is applied to reference transistor. The reference current is the same ratio <NUM>/α to the specified trip point current as the size ratio of the two transistors. The Vds of the reference transistor equals the Vds of the switch transistor when the current of the switch transistor is α times the reference current.

According to some aspects, an error amplifier is used to set the reference current in the reference transistor so that the Vds of the reference transistor tracks the Vds of the switch transistor. The Vds of the reference transistor is monitored and compared to trip point voltage corresponding to the trip point current of the switch transistor. When the monitored voltage exceeds the trip point voltage, the current in the switch transistor exceeds the specified current, and the switch transistor is turned off.

Claim 1:
An electronic circuit breaker, ECB, circuit comprising:
a sense field effect transistor, FET, (MPWR) having a drain and a source coupled in between an input (IN) and an output (OUT) of the ECB, wherein the sense FET is disposed in a first voltage domain (HV) of the ECB;
a reference FET (MREF) disposed in a second voltage domain (LV) of the ECB, wherein the reference FET is configured to conduct a reference current (IREF) between a drain and a source of the reference FET; and
a sensing circuitry disposed in the first voltage domain and in the second voltage domain, wherein the sensing circuitry includes:
a switched capacitor circuit configured to sample a drain-to-source voltage, Vds, of the sense FET, the switched capacitor circuit comprising a first isolation capacitor (C1) and a second isolation capacitor (C2), wherein the first isolation capacitor (C1) and the second isolation capacitor are arranged to isolate the second voltage domain from the first voltage domain and comprise, respectively, a first terminal disposed in the first voltage domain and a second terminal disposed in the second voltage domain, and
a comparison circuitry disposed in the second voltage domain comprising a first input terminal and a second input terminal and configured to determine if the sampled Vds of the sense FET is greater than or less than a Vds of the reference FET in order to determine if a drain-source current of the sense FET is greater than or less than a specified current, wherein the first and second input terminals of the comparison circuitry are coupled to the second terminal of the first isolation capacitor (C1), to the second terminal of the second isolation capacitor (C2), and to the drain and source source of the reference FET (MREF).