Patent Description:
The present invention, in some embodiments thereof, relates to cloaking and deception of a detection system and, more particularly, but not exclusively, to cloaking and/or deception of a Doppler radar. Embodiments of the present invention can generally be applied to against any detection system, which contains a moving target indicator (MTI), based on phase information.

The invention of radars was soon followed by extensive research and development of counter measures. By employing special geometric designs and carefully selected materials, a reduction of the target's radar cross section and the resulting backscattered energy was successfully achieved, substantially reducing distances between the radar and the target for successful detection. In addition to stealth technology, numerous active jamming countermeasures have been developed. In this case, signals are transmitted to the investigating radar systems, either to cause the radar to wrongly conclude estimation parameters, or to blind the radar by degrading the signal to nose ratio.

The field of meta-materials has further advanced the ability to cloak objects of interest from investigation by introducing man-made materials to tailor electromagnetic scattering [<NPL>); <NPL>), <NPL>)].

Chinese Patent Application No. <CIT> discloses an electromagnetic wave frequency conversion time domain super surface. Basic units are arranged periodically and are controlled by signal generated by the same control circuit. When the super surface is irradiated by the electromagnetic wave, real-time control of each characteristic parameter of the reflected wave can be realized.

Published Application No. <CIT> discloses a method for controlling radiation scattering. Radiation is directed to a metamaterial having an array of individually tunable and near field coupled resonators. An individual resonance frequency of at least one of the resonators is varied to provide a collective resonance scattering of the radiation by the metamaterial. A scattering signal of the collective resonance scattering of the radiation is measured, and the metamaterial is identified based on the scattering signal.

Saikia et al disclose an electromagnetic wave modulation technique using actively tuned frequency-selective surfaces to shift the frequency of electromagnetic waves reflected from the surface. The reflected electromagnetic wave contains significant components at multiple sideband frequencies along with the original incident frequency. The technique applies time-modulation to reflection coefficient to suppress the original incident frequency component and significantly boost the sidebands.

Liu et al disclose a technique for making an object invisible for detectors, using a broadband switchable metasurface integrated with p-i-n diodes, where the reflection phase of the metasurface can be changed. By modulating the metasurface quasi-randomly in the time domain, the incident narrow-band signal is spread into a white-noiselike spectrum upon reflection, creating a spectral camouflage.

The present invention provides cloaking system according to claim <NUM>, a vehicle according to claim <NUM>, and a method of cloaking or deception a detection system according to claim <NUM>.

The Inventors found that achieving requirements that are desired to provide a countermeasure against real radar systems is challenging. The Inventors found that some of the factors to consider are: (i) all angle of incidence operation, (ii) dial polarization operation, (iii) bandwidth, and (iv) conformity to geometries (including and not limited to mechanic rigidity, weigh and other system parameters), dictated by a real object, subject to cloaking.

According to some embodiments of the invention the present invention there is provided a cloaking system. The system comprises: a structure having a plurality of resonators having a controllable resonance frequency, wherein the resonators are arranged to collectively ensure that variation of the resonance frequency over a predetermined range of resonance frequencies generates a phase shift between the an electromagnetic wave incident on the structure and an electromagnetic wave scattered off the structure; and a controller configured for controlling the resonance frequency to provide a time-varying resonance frequency having a temporal function which comprises a linear time-dependence;
characterized in that said controller (<NUM>) is configured to receive velocity data characterizing a motion of a vehicle (<NUM>) and to select said time-varying resonance frequency based on said velocity data.

According to some embodiments of the present invention the system is configured for a central frequency of the incident wave, wherein at least one of the resonators has a dispersive response to the incident wave, the dispersive response being selected to ensure that the phase shift range is effective for any frequency within a frequency band of at least <NUM>% of the central frequency.

According to some embodiments of the invention at least one of the resonators is configured to maintain, within a predetermined tolerance, equality between a frequency of the scattered wave and a frequency of the incident wave.

According to some embodiments of the invention at least one of the resonators comprises an electronic element having a controllable impedance, wherein the controlling the resonance frequency comprises varying the impedance.

According to some embodiments of the invention at least one of the resonators comprises an electric dipole defining an airgap, and an electronic element having a controllable impedance at the airgap.

According to some embodiments of the invention at least one of the resonators comprises a magnetic dipole, and an electronic element having a controllable impedance.

According to some embodiments of the invention the system comprises a metal screen and dielectric structure, between the resonators and the metal screen, wherein the resonators are mounted on the dielectric structure.

According to the claimed invention the controller is configured to receive velocity data characterizing a motion of a vehicle and to select the time-varying resonance frequency based on the velocity data.

According to an aspect of some embodiments of the present invention there is provided a vehicle. The vehicle comprises a propulsion system carried by a vehicle body; and the system as delineated above and optionally and preferably as further detailed below, mounted on an external surface of the vehicle body.

According to some embodiments of the invention the vehicle is a manned vehicle.

According to some embodiments of the invention the vehicle is an unmanned vehicle. According to some embodiments of the invention the vehicle is a controllable vehicle. According to some embodiments of the invention the vehicle is an autonomous vehicle.

According to the claimed invention the controller is configured to receive velocity data characterizing a motion of the vehicle and to select the time-varying resonance frequency based on the velocity data.

According to an aspect of some embodiments of the present invention there is provided a method of cloaking or deception a detection system transmitting an electromagnetic wave having a central frequency. The method comprises: scattering the detection system's wave off a structure having a plurality of resonators having a controllable resonance frequency, wherein the resonators are arranged to collectively ensure that variation of the resonance frequency over a predetermined range of resonance frequencies generates a phase shift between the detection system's wave and an electromagnetic wave scattered off the structure; and controlling the resonance frequency to provide a time-varying resonance frequency having a temporal function;
characterized in that the method comprises receiving (<NUM>) velocity data characterizing a motion of a vehicle and selecting said time-varying resonance frequency based on said velocity data.

According to some embodiments of the invention at least one of the resonators has a dispersive response to the detection system's wave, the dispersive response being selected to ensure that the phase shift range is effective for any frequency within a frequency band of at least <NUM>% of the central frequency.

According to some embodiments of the invention the dispersive response comprises a frequency-dependent impedance.

According to some embodiments of the invention an associated reactance of the frequency-dependent impedance is a decreasing function of the frequency.

According to some embodiments of the invention the electronic element has a voltage-dependent impedance, and wherein the varying the impedance comprises varying a voltage applied to the electronic element.

According to some embodiments of the invention a time-dependence of the variation of the voltage is nonlinear and selected to at least partially cancel nonlinearities in a voltage-dependence of the impedance.

According to some embodiments of the invention the temporal function comprises a linear time-dependence.

According to some embodiments of the invention the resonators are mounted on a dielectric structure which is between a metal screen and the resonators.

According to the claimed invention the method comprises receiving velocity data characterizing a motion of a vehicle and selecting the time-varying resonance frequency based on the velocity data.

According to some embodiments of the invention the phase shift is over a respective range of at least <NUM>.

Referring now to the drawings, <FIG> is a schematic illustration of a system <NUM> suitable for cloaking a detection system, according to some embodiments of the present invention. The detection system can be any detection system. Preferably, the detection system contains a moving target indicator (MTI) based on phase information. In some embodiments of the present invention the detection system is a radar.

System <NUM> comprises a structure <NUM> having a controllable resonance frequency, and a controller <NUM>, having a circuit configured for activating the structure <NUM>. Controller <NUM> and structure <NUM> typically communicate via one or more communication lines <NUM>, which can be wired, as illustrated, or wireless.

Structure <NUM> is a synthetic cellular structure that scatters a wave interacting therewith. The wave is typically an electromagnetic wave transmitted by a detection system (not shown) so as to interrogate an object (e.g., vehicle) carrying system <NUM>. Preferably, but not necessarily, the scattered wave has at a frequency of from about <NUM> to about <NUM>.

As used herein, "cellular" is used to indicate that the structure defines a network of generally repeating and inter-coupled cells <NUM>. The coupling between the cells <NUM> is preferably near field coupling.

As used herein "near field coupling" refers to interaction by exchanging a non-radiative physical field (e.g., electric field, magnetic field, electromagnetic field).

Structure <NUM> can be in any known form that has controllable resonance frequency. Representative examples including, without limitation, a metamaterial a metasurface, a time-dependent mask and the like.

Preferably, but not necessarily, the cells <NUM> of structure <NUM> are arranged as an array. The array shown in <FIG> is defined over a rectangular grid, but this not necessarily be the case, since, for some applications, it may be desired to define a non-rectangular grid (e.g., triangular, hexagonal, etc).

Each of the cells <NUM> optionally and preferably comprises a resonator <NUM>, which is a circuit that is configured to electromagnetically resonate at a frequency referred to as a resonance frequency. In various exemplary embodiments of the invention the resonance frequency of the resonator <NUM> is controllable, and the circuit of controller <NUM> is configured to control this frequency as further detailed below. Controller <NUM> can control each of the resonator circuits individually, or it can be configured to control one or more (e.g., all) the resonator circuits collectively. When two or more resonator circuits are controlled individually, the controller can set different resonance frequencies to different individually-controlled circuits. When two or more resonator circuits are controlled collectively, the controller can apply the same change to the resonance frequencies of all the collectively-controlled circuits (e.g., the controller can set the same resonance frequencies to the collectively-controlled circuits). Adjustable control over resonant frequencies can optionally and preferably also provide a countermeasure against frequency hopping interrogating systems.

Resonators <NUM> are typically mounted (e.g., soldered, glued, printed, or otherwise connected) on a dielectric structure <NUM>, serving for supporting the array. Dielectric structure <NUM>, is optionally and preferably conformal to surface of the object to be concealed from the detection system.

In use, the side of dielectric structure <NUM> which is opposite to the resonator array is mounted on an external surface of an object to be concealed from the detection system, thereby also serving as a spacer between the surface of the object and the resonators. The thickness of dielectric structure <NUM> is typically several millimeters but other thicknesses are also contemplated. Dielectric structure <NUM> is optionally and preferably made of a material that is transparent to the wave for which system <NUM> is designed, which is typically the frequency band of the electromagnetic radiation which is expected to be transmitted by the detection system. The dielectric losses can degrade the resonant behaviors of the structure <NUM>, but can be compensated by additional elements. In some embodiments of the present invention a metal screen <NUM>, can be introduced between the surface of the object to be concealed and the structure <NUM> to uncouple electromagnetic properties of the object's surface from structure <NUM>. The thickness of screen <NUM> is optionally and preferably several skin depths of the incident wave.

In some embodiments of the present invention the circuit of controller <NUM> is configured for controlling the resonance frequency of resonators <NUM> to provide a time-varying resonance frequency. Preferably, the controller ensures that the resonance frequency variation is characterized by a temporal function which comprises a linear time-dependence, more preferably a temporal function which is dominated by a linear time-dependence, more preferably a linear temporal function. In some embodiments of the present invention the entire resonance of the structure <NUM> can be shifted to a desired frequency to cope with, for example, frequency hopping radars. The Examples section that follows demonstrates broadband operation by a circuit, such as, but not limited to, the circuit shown in <FIG>.

A temporal function is said to be dominated by a linear time-dependence, if a ratio between the nonlinear part and linear parts of the temporal function is less than <NUM>% for any time during the variation applied by the controller.

Preferably, the linear time-dependence is linear modulo 2π. Mathematically, a time-dependence which is linear modulo 2π can be written as f(t) = a + b t (mod 2π), where f is an observable (e.g., resonance frequency) which varies according to the time dependence, a is a constant offset parameter, b is a constant slope parameter, t is the time variable of the time-dependence, and mod is a function which returns the modulus of the operation, which in this case is the remainder of the division of t by 2π.

The inventors found that sleeting a temporal function which comprises a linear time-dependence, is advantageous for cloaking, in particular when the incident wave is transmitted by a detection system employing a moving target indicator (MTI) filter, as will now be explained with reference to <FIG>.

<FIG> schematically illustrates a moving vehicle <NUM> having a body <NUM> carrying a propulsion system <NUM>. Cloaking system <NUM> is mounted on an external surface of body <NUM>. The system of the present embodiments can be mounted on any type of manned or unmanned vehicle, either controllable, or autonomous. Representative examples of vehicles suitable for the present embodiments include, without limitation, an aerial vehicle (e.g., a drone, an aircraft, a jet airplane, a helicopter, an unmanned aerial vehicle, a passenger aircraft, a cargo aircraft), a ground vehicle (e.g., an automobile, a motorcycle, a truck, a tank, a train, a bus, an unmanned ground vehicle), an aqueous vehicle (e.g., a boat, a raft, a battleship), an amphibious vehicle and a semi-amphibious vehicle.

Each of the gray squares that are shown in <FIG> on body <NUM> can enact system <NUM>, including its own frequency-controllable structure and its own controller, or, more preferably, at least some of (e.g., all) the gray squares can be sub-systems of system <NUM>, each comprising its own frequency-controllable structure but uses a controller that is shared among the sub-systems. In a preferred embodiment, the frequency-controllable structure(s) of system <NUM> cover a majority of the external surface area of body <NUM>. <FIG> also illustrates a radar <NUM> which transmits a wave <NUM>. A portion of wave <NUM> incidents on vehicle <NUM>, and a portion of wave <NUM> incidents on static objects in the field-of-view of radar <NUM>. A representative example of such a static object is mountain <NUM>. Backscattered waves from vehicle <NUM> and mountain <NUM> are shown at <NUM> and <NUM>, respectively.

Radar <NUM> receives the backscattered wave <NUM> and <NUM> and analyzes phase variations caused by the Doppler effect. Radar <NUM> considers echoes that do not exhibit phase variations as originating from static object, and filters out signals corresponding to those echoes, so as to reduce clutter. Such a filter is known as an MTI filter (see, e.g., <NPL>). Thus, signals corresponding to wave <NUM> are filter out by the MTI filter, because mountain <NUM> is static and so wave <NUM> does not exhibits phase variation.

The inventors found that variation of the resonance frequency of the resonators of system <NUM> causes backscattered wave <NUM> to be phase shifted relative to transmitted wave <NUM>. The inventors have therefore postulated, and showed experimentally, that by controlling the resonance frequency of the resonators according to a temporal function which comprises a linear time-dependence the change of the phase of wave <NUM> due to the Doppler effect can be at least partially compensated. Such a compensation or partial compensation conceals the Doppler signature of vehicle <NUM>, making it appear to radar <NUM> as stationary as, e.g., mountain <NUM>, or any other object, e.g., a tree a cloud a ground or the like. This causes the MTI filter of radar <NUM> to filter out also the signals corresponding to wave <NUM>, thus significantly reducing the visibility of vehicle <NUM>, or, more preferably, rendering it invisible to radar <NUM>.

The circuit of controller <NUM> (see <FIG>) receives velocity data characterizing the motion of vehicle <NUM> and selects the time-varying resonance frequency based on the velocity data. For example, controller <NUM> can derive from the velocity data a linear time-dependence characterized by a slope parameter that is linearly proportional to the velocity of vehicle <NUM>, and vary the resonance frequency according to the derived time-dependence, preferably modulo 2π, thus compensating for the Doppler phase shift due to this velocity. The controller can also include a detection of arrival (DoA) detector to define the interrogation direction. The time-variation optionally and preferably compensates the radial velocity in respect to the antenna of the detection system. Angular reflectivity of the device can also be adjusted accordingly.

In some embodiments of the present invention the circuit of controller <NUM> is configured to not significantly modulate the frequency of the transmitted wave <NUM>. In other words, in these embodiments the circuit of controller <NUM> maintains, within a predetermined tolerance (e. g, ±<NUM>%, or ±<NUM>%, or±<NUM>%), equality between the frequency of scattered wave <NUM> wave and the frequency of transmitted wave <NUM>.

The resonators <NUM> of system <NUM> are preferably arranged to collectively ensure that when the resonance frequency is varied over a predetermined range of resonance frequencies, the resulted phase shift between the wave incident on structure <NUM> and the wave scattered off structure <NUM>, is over a respective range of at least <NUM>. 5π, more preferably at least <NUM>. 8π, more preferably at least <NUM>. 9π, e.g., about 2π or more. Specifically, characterizing the predetermined range of resonance frequencies by a lower frequency threshold fL and an upper frequency threshold fU, and the resonators <NUM> of system <NUM> are preferably arranged to collectively ensure that there is a one-to-one mapping between the range [fL,fU] and the phase shift range [<NUM>,φMAX], where φMAX is at least <NUM>. 8π, more preferably at least <NUM>. 9π, most preferable about 2π or more. For example, a resonance frequency of fL can be mapped to a zero phase shift, a resonance frequency of fU can be mapped to a phase shift of φMAX, and any resonance frequency f satisfying fL < f < fU can be mapped to a unique phase shift φ satisfying <NUM> < φ < φMAX.

With reference to <FIG>, the controllability of the resonance frequency of resonator <NUM> can in some embodiments of the present invention be achieved by providing each resonator with an electronic element <NUM> having a controllable capacitance or any other resonant shifting element, such as, but not limited to, an inductor. In this case controller <NUM> controls the resonance frequency of the resonator by varying the impedance (e.g., capacitance) of element <NUM>. For example, electronic element <NUM> can have a voltage-dependent impedance, and controller <NUM> can control the resonance frequency by varying the voltage applied to the electronic element. A representative example of an electronic element with a voltage-dependent impedance, and which is suitable for the present embodiment is a varactor. The dependence of the impedance on the voltage need not to be linear. For example, in varactors the impedance typically varies nonlinearly with the applied bias voltage. When the dependence of the impedance on the voltage is nonlinear, controller <NUM> preferably varies the voltage nonlinearly with the time according to a nonlinear time-dependence selected to at least partially cancel the nonlinearities of the voltage-dependence of the impedance. A representative procedure for canceling nonlinearities of the voltage-dependence is described in the Examples section that follows.

Resonator <NUM> typically also comprises an antenna <NUM> that interacts with the incident wave and resonate responsively to this interaction. In the schematic illustration of <FIG>, which is not to be considered as limiting, the antenna <NUM> is a dipole antenna defining an airgap <NUM>, wherein electronic element <NUM> is at the airgap <NUM>. It is to be understood that other shapes for the antenna <NUM> are also contemplated.

While the embodiments below are described with a particular emphasis to electric dipoles, it is to be understood that the present embodiments also contemplate use of magnetic dipoles instead of, or in addition to, electric dipoles.

In some embodiments of the present invention one or more of resonators <NUM> has a dispersive response to the incident wave. This is advantageous since the dispersive response of resonator <NUM> can be selected to increase the bandwidth over which the aforementioned one-to-one mapping between the range of resonance frequencies and the phase shift range is effective. Preferably, the dispersive response is selected such that the phase shift range is effective for any frequency within a frequency band of at least <NUM>% or at least <NUM>% or at least <NUM>% or of the central frequency of the incident wave.

A dispersive response of resonator <NUM> can be achieved by constructing the electronic element <NUM> as a dispersive element. For example, the dispersive response can be a dispersive impedance (e.g., capacitance), in which case electronic element <NUM> can be constructed to exhibit a dispersive impedance property, e.g., a frequency-dependent impedance. When the controllability of electronic element <NUM> is embodied as a voltage-dependent impedance, the dispersive impedance of electronic element <NUM> can be achieved by combining two or more frequency responsive elements (e.g., capacitive elements, inductive elements), where at least one of these capacitive elements has a voltage-dependent impedance and at least one these frequency responsive elements has a frequency-dependent impedance, thereby providing an electronic element in which the impedance varies both with the voltage and with the frequency. The use of dispersive element is advantageous since it increases the operation bandwidth. However, the inventors found that the system of the present embodiments is useful against many detection systems even without dispersive elements, wherein element <NUM> is a nondispersive element. A schematic illustration of an equivalent circuit describing the electronic property of electronic element <NUM> suitable for these embodiments is provided in <FIG>. Shown in <FIG> are a controllable voltage-dependent capacitor 24a (e.g., a varactor) and a frequency-dependent capacitor 24b, connected in parallel to each other so that the effective capacitance C of element <NUM> is C=CV+Cw, where CV is the capacitance of capacitor 24a and Cw is the capacitance of capacitor 24b. While <FIG> illustrators to additional elements in the equivalent circuit, it is to be understood that the equivalent circuit may include additional or other elements, provided these elements aid in tuning the resonance frequency of the cells <NUM>.

In various exemplary embodiments of the invention the frequency-dependent capacitor 24b is not controlled by controller <NUM>, so that any variation in the capacitance Cω of capacitor 24b is in response to the incident wave <NUM>. In some embodiments of the present invention the associated reactance of the frequency-dependent capacitance Cω of capacitor 24b is a decreasing function of the frequency. It is to be understood that capacitor 24b need not be a capacitor per se, and that active electronic circuitry can be designed to enact capacitor 24b.

Reference is now made to <FIG>, which is a flowchart diagram describing a method suitable for cloaking a detection system transmitting an electromagnetic wave characterized by a central frequency, according to some embodiments of the present invention. Selected operations of the method can be executed using system <NUM>.

The method begins at <NUM> and continues to <NUM> at which the method receives velocity data characterizing a motion of a vehicle. The method continues to <NUM> at which the detection system's wave is scattered off a frequency-controllable structure, such as, but not limited to, structure<NUM>. The method continues to <NUM> at which resonance frequencies of the structure are dynamically control to provide a time-varying resonance frequency characterized by a temporal function which comprises a linear time-dependence, as further detailed hereinabove. The temporal function is selected based on the velocity data as further detailed hereinabove.

Concealing objects from interrogation has been a primary objective since the integration of radars into surveillance systems. Metamaterial-based invisibility cloaking, which was considered a promising solution, did not yet succeed in delivering reliable performance against real radar systems, mainly due to its narrow operational bandwidth. This Example demonstrates an approach, which addresses the issue from a signal-processing standpoint and, as a result, is capable of coping with the vast majority of unclassified radar systems by exploiting vulnerabilities in their design. In particular, this Example demonstrates complete concealment of a <NUM> square meter moving metal plate from an investigating radar system, operating in a broad frequency range approaching <NUM>% bandwidth around the carrier of <NUM>. The radar countermeasure is based on a temporally modulated coating. This auxiliary structure is designed to dynamically and controllably adjust the reflected phase of the impinging radar signal, which acquires a user-defined Doppler shift. A particular case discussed herein imposes a frequency shift that compensates for the real Doppler signatures originating from the motion of the target. In this case the radar considers the target static, even though it is moving. As a result, the reflected echo is discarded by the clutter removal filter, which is a part of any modern radar system that is designed to operate in real conditions. This allows rendering the target invisible to the radar even though it scatters electromagnetic radiation.

Modern radar systems can simultaneously measure the location and radial velocity of investigated targets. In the simplest terms, their method of operation is based on transmitting modulated (for example pulsed) electromagnetic radiation towards a target and recording the reflected echoes <NUM>-<NUM>. From the delay between the transmitted and received signals (time of flight) the range to the target can be deduced, while the phase difference between consecutive pulses, produced by the Doppler effect, allows the measurement of the instantaneous radial velocity.

The semi-passive approach to radar invisibility described in this Example does not require transmitting signals to confuse or jam the radar, nor does it require a lot of a priori knowledge about the type of radar at hand. Instead, a temporally modulated reflecting coating is suggested, which can control the time dependent phase of the electromagnetic field as it is backscattered towards the interrogating radar.

Owing to the fact that the Doppler information is extracted from the difference in the phase of consecutive pulses, dynamically controlling the reflected phase from the target produces backscattered echoes, which contain fake Doppler signatures that are indistinguishable from the ones created by genuine motion. It is therefore possible to deceive a radar system into concluding it is observing a moving target when the target is in fact stationary. The method described in this Example makes it possible to compensate the real phase difference between consecutive pulses, which originates from the movement of the target, thereby cloaking the signatures of motion and making the target appear stationary to the radar.

The Doppler-cloaked target still scatters a lot of energy since it is not employing any method of scattering suppression. The method of the present embodiments serves to deny the interrogating system information about the target's instantaneous velocity, which is useful for the proper operation of numerous radar systems relying on clutter removal methods, such as the moving target indicator (MTI) filter <NUM>-<NUM>. The absence of Doppler information originating from the target makes it indistinguishable from the surrounding clouds, mountains and ground, which can backscatter much more energy with very small Doppler shifts. This means that the MTI filter removes the energy related to the target along with the rest of the clutter, rendering it effectively invisible to the radar.

For narrowband radars, typically defined as those having less than <NUM>% bandwidth around the central frequency, it is sufficient to cloak the Doppler signature around the central frequency. For broadband signals, on the other hand, the Doppler shift in the entire range should be cloaked, requiring broadband phase matching as will be discussed ahead.

In order to understand the operation of the metasurface Doppler cloak, it is instructive to consider a single dipole, which allows gaining physical insight into the phenomenon. This insight will be used to understand the basic operation principle behind the suggested broadband invisibility concept.

The polarizability of a dipole has a Lorenzian shape in the frequency domain <NUM>, where near the resonance the phase is approximately linear in frequency. The dipole is excited by an incident radiation, which is partially reflected back into the source (e.g. a radar antenna). If the resonant frequency of the dipole it temporally modulated, the scattered field acquires an additional time-dependent phase shift. Note that radar systems almost never rely on the amplitude of scattered echoes for detection, mainly due to its unpredictability in real environments and unidentified targets. Temporal modulation of the dipole is realized by incorporating a voltage-controlled capacitor (varactor) within the structure. <FIG> demonstrates a lumped elements scheme for the scattering scenario containing the dipole and varactor. The impinging wave is represented by the voltage source Vin, while the resistance R, capacitance C, and inductance L characterize the dipole and depend on the material composition and geometric shape of the resonator. Placing a voltage dependant varactor in the feeding gap of the dipole can serve as a resonance-shifting element, allowing control over the scattered phase, shown in <FIG>. The varactor is represented on the scheme as an additional capacitor Cv in parallel to the dipole's natural one (C). The current flowing through the resistor is related to the scattered electromagnetic field and its phase is the goal of the following derivation. The current in the frequency domain is given by: <MAT> where ω is the angular frequency of the impinging radiation. It is important to note that a time scale separation method is used to derive Eq. <NUM>. The assumption is that any time-dependent changes in the varactor's capacitance are far slower than the carrier frequency of the exciting radiation. In this case, the dipole may be considered as stationary at any particular time, solving the fast scattering problem while keeping the capacitance Cv(t) as a parameter. As it will be shown ahead, the required modulation frequency of the varactor is of no more than a few kHz, while typical radar systems transmit above <NUM>, making this approximation perfectly justifiable.

The phase of the current in Eq. <NUM> is given by: <MAT>.

<FIG> summarizes the results of Eq. <NUM>, demonstrating that the phase undergoes rapid change from π to <NUM> around the resonant frequency, which is controlled by the varactor capacitance. Other system parameters used for the plot are: C = <NUM>pF, L = <NUM>µH and R = 50Ω. Note that these values were chosen in order to obtain a response similar to the one found in experiment, yet this combination is obviously not unique. Since radar systems are not sensitive to absolute phase, but rather to the phase difference between consecutive pulses, it is more instructive to inspect the change in phase as a function of varactor capacitance: <MAT> where Cv(<NUM>) is assumed to be <NUM> for simplicity. Plotting Eq.<NUM> versus the incident frequency f = ω/<NUM>π produces <FIG>, where a knife-like image may be seen. Any horizontal cut-line of the knife represents a phase shift versus frequency, similar to the plots shown on <FIG> (specifically subtraction between curves (vi) and (vi)). On the other hand, controlling the bias of the voltage drop on the varactor diode while keeping the incident frequency constant, is equivalent to taking a vertical cut-line in <FIG>. It can be seen that with increasing varactor capacitance the phase goes from <NUM> to π abruptly in an almost step-like manner, with the transition capacitance depending on the incident frequency. This transition capacitance is located along the knife's edge, and its accurate conditions will be derived later on. The conclusion from the above discussion is that for any frequency of incident radiation, it is possible to continuously induce up to a π phase shift in the reflected field by carefully tuning the bias voltage of the varactor diode. While this π-shift can severely hamper the ability of an investigating system to deduce instantaneous velocity, full control over <NUM>π phase is desirable for achieving complete invisibility, as will be discussed ahead.

As demonstrated above, a single dipolar scatterer is not sufficient to provide full phase control over the reflected wave, motivating the development of more advanced configurations. This Example shows that resonator-based reflect arrays, often termed as metasurfaces, indeed can allow controllable <NUM>π phase shift of the reflected waves. A typical example is a structure with a switchable characteristic impedance, which has properties resembling either perfect electric or a perfect magnetic conductor. In this case the reflection coefficient varies from '-<NUM>' to '<NUM>' respectively and thereby allowing to obtain full control over the reflected phase <NUM>,<NUM>. While analytic models for arrays of scattering dipoles do exist <NUM>,<NUM>, they might be quite cumbersome for obtaining immediate physical insights. In addition, these models tend to neglect higher-order multipolar interaction, edge effects in finite sized systems and several other aspects, which might be important in practical realizations. Instead, it is frequently preferable to use full wave numeric simulations in order to optimize the metasurface and obtain the desired results. This is the approach undertaken ahead using the time domain FDTD method implemented in CST Studio. <FIG> shows a single unit cell of a 9x9 array of dipoles on a dielectric substrate (FR-<NUM>, εr = <NUM>), which is located above a metallic surface, assumed to be a perfect electric conductor. The metallic surface represents the target that is to be cloaked. A biasing network, made out of thin wires, provides the temporal modulation of the voltage drop, which is used to control the capacitance of the varactor. An additional capacitor Cω(ω) is shown in parallel to the varactor, however it will only be required later on and is assumed to be disconnected in the following discussion. <FIG> shows a color map of the phase shift of the reflected field as a function of substrate thickness (distance between the dipole and the metal surface, denoted as 'h' on <FIG>) and incident frequency. The map is obtained by repeating the full wave simulation for different substrate thicknesses, while discretely switching the capacitance between the maximum and minimum values of <NUM>. 6pF and <NUM> pF, in accordance with the datasheet of the varactor diode that is used in the experimental set up. A full <NUM>π phase shift is thus clearly achievable for substrate thicknesses between about <NUM> and <NUM> millimeters, when the other dimensions are fixed at d = <NUM>, <NUM> = <NUM>, ls=<NUM>, ws=<NUM>, w=<NUM>, the dipole width is <NUM> and the width of the biasing network wires is much smaller than <NUM>. It is worth noting that this numerical optimization of the entire structure allows approaching experimental realization quite closely, as will be seen later.

<FIG> shows the phase shift of the metasurface as a function of the incident wave frequency and the varactor's capacitance, bearing a remarkable resemblance to <FIG>. Indeed, a knife-like image can be seen on both plots. The difference lies in the fact that a maximal controllable phase shift of <NUM>π can be obtained with this array, unlike the π phase shift attainable with a single dipole. The resulting array is similar to the phase switched screen, which was previously used to redistribute the reflected energy outside of the receiving radar band, therefore making it less visible to the interrogator. This was achieved by fast switching of reflectivity between two values, causing broadband modulation of the incident field.

This Example investigates a perturbative and quasi-static approach, which does not significantly modulate the frequency of the incident wave - this is useful in the case of wide-band radar systems and provides significantly better performances in passive deception applications, since the low frequency modulation does not radiate at the switching frequency. The purpose of the modulation is to produce a linear time dependent phase shift of the backscattered field, which exactly compensates for the linear phase shift produced by the motion of a target via the Doppler effect. The linearity of the phase can be achieved by modulating the bias voltage in time with the inverse function of the capacitance-phase relation shown on the vertical cuts of <FIG>, which serves to "straighten out" the phase dependence on time at the frequency of interest. However, linear phase response is not retained across the entire band, seeing as the threshold varactor capacitance, which is the edge of the knife in <FIG>, varies from frequency to frequency. Additional correction is optionally and preferably be taken to achieve broad phase matching.

It is preferred to control the phase change across a range of frequencies with an identical (shared along the band) driving varactor bias voltage. The time dependant varactor biasing approach, summarized in <FIG>, shows that the discussed metasurface might not cover the entire bandwidth of a wideband radar system (typically defined as above <NUM>% around the carrier). The reason is that for a range of frequencies the phase difference transitions from <NUM> to <NUM>π occurs at different varactor capacitance values. In order to achieve a broadband response, the knife-shape of <FIG> is optionally and preferably be transformed into a rectangular form, where the transition from <NUM> to <NUM>π occurs at the same cutoff values of the varactor capacitance, leading to broadband phase matching. To accomplish this goal, an additional frequency-dependent capacitor is optionally and preferably introduced within the circuit in parallel with the varactor, as shown on <FIG>. The goal of this new element is to 'straighten out the knife' by shifting the frequency dependent threshold capacitance toward lower varactor values. In the case of the single dipole discussed earlier, this transitional cutoff capacitance Cω(ω) may be derived from Eq. <NUM>. by updating the lumped elements scheme to include the new dispersive element Cv → Cv + Cω and setting <MAT> as a convenient but arbitrary value for the threshold, after which the phase difference transitions from <NUM> to 2π. It is then possible to solve and obtain an expression for Cω: <MAT>.

Analysis of Eq. <NUM> shows that Cω is a decaying function of frequency (for the values of L, C and R that were used before) and its plot versus frequency is in fact the edge of the knife shown on <FIG>. For the array of dipoles, on the other hand, no simple formula exists and a numerical approach is undertaken in a similar fashion by using the color map of <FIG> in order to obtain the threshold capacitance. <FIG> shows a simulation, which is identical to the one performed in FIG. 3C, with the exception of an additional frequency dependent capacitor that was placed in the gap of each dipole in the array. The frequency dependant capacitor was modelled as two subwavelength metallic plates with a frequency dependant dielectric material in between. The material demonstrates anomalous dispersion, which resembles the leading edge of the "knife's edge", and provides, in fact, the required rectifying capacitance Cω. It can be seen that after adding this capacitor, around <NUM> of bandwidth become available for simultaneous phase switching. This represents more than <NUM>% of bandwidth around the carrier (assuming the carrier is in the middle of the band). An additional shift of the rectified knife in comparison with the knife shape in <FIG> is observed. This is due to the addition of the new capacitor, and is expected due to the fact the dispersive capacitance values do not vanish anywhere in the observed frequency band, serving to lower the resonant frequency of the array. It is noted that -<NUM>/(ωCω) is decreasing with frequency. This behaviour can be achieved by using dispersive dielectric materials between the capacitor's electrodes, or alternatively using active circuits <NUM>.

In order to demonstrate the capabilities of the described metasurface at concealing a large target from an investigating radar, an experimental device was fabricated and is shown on <FIG>.

The array of 9x9 dipoles was manufactured according to the simulation presented in the previous section with the same dimensions. The dipoles were chemically etched from a copper surface that was deposited on top of a dielectric FR-<NUM> substrate. SMV1405 varactors were soldiered in the dipole gaps, while the edges of the dipoles were soldiered to thin wires forming the biasing network. The array of dipoles was glued on top of a supporting structure, which was transparent to centimetre waves and served as a spacer of <NUM>, altogether forming the metasurface.

This metasurface was designed to be placed in front of the metal plate, which was the target to be hidden from the radar. The target covered by the metasurface was placed on a polyester structure that connected it to a motorized conveyor belt, which enabled moving it forward and backward with a controllable speed reaching up to about <NUM>/s. A stepped frequency continuous wave (SFCW) radar system was implemented with a Network Analyzer, which is capable of sweeping the entire band of interest (<NUM>-<NUM>) while recording the amplitude and phase of the received echoes from the target. This type of radar is typically used in ultra-wideband applications since it is able to transmit carriers sequentially, while the receiver is locked on the transmitted frequency in a predefined time window.

This architecture allows avoiding expensive high frequency samplers that would otherwise be needed for sampling extremely short pulses <NUM>. The radar's antenna was placed directly in front of the moving conveyor inside an anechoic chamber and served both for transmitting and receiving the radiation (monostatic radar scheme), linearly polarized in the direction of the dipoles (horizontal).

Moving the concealed target without modulating the bias voltage produced linear phase shifts in time as can be seen on <FIG>, which show forward and backward motion correspondingly at constant velocity of <NUM>/s and under <NUM> radar illumination. It can be observed that the full span of <NUM>π phase was traced in the process of the movement as expected.

In the following experiments the target was moved at a constant velocity of -<NUM>/s (negative sign indicating motion away from the radar) while various bias voltages were applied to the metasurface. Since the relationship between the modulating voltage and scattered phase is not linear, a calibration procedure was first performed. By applying a linear voltage modulation to the biasing network while keeping the target stationary, the nonlinear temporal phase profile was recorded. Applying the inverse of that function back into the input of the biasing network produced the calibrated linear phase shown on <FIG>, where a maximal phase shift of Δφ = <NUM>° is achieved, close to the theoretical maximum. It is noted that the phase modulation produced by the metasurface is very similar to the one produced by actual motion, as seen on <FIG>. Controlling the modulation frequency fm of the metasurface (which is the frequency of the applied periodic voltage drop on the biasing network) while moving the target leads to plots on <FIG>. The sign of the frequency indicates the direction of phase modulation (positive and negative signs correspond to increasing and decreasing phase in time, accordingly). The modulation frequency controls the slope of the phase shift φMetasurface(t) which is added together with the phase shift produced by the motion of the target via the Doppler Effect as follows: <MAT>.

<FIG> show the results for the moving target while different modulation frequencies are applied to the metasurface. In <FIG> the modulation is of the same polarity as the real motion of the target, leading to their phases adding up as in Eq.<NUM>. The result is an overall faster changing slope, corresponding to faster motion than that of the real target. In contrast, <FIG> has the modulation polarity in the opposite direction to that of the real motion, causing the target to appear slower than it is, as well making it appear to be moving towards the radar while in reality it is moving away from it. In <FIG>, the modulation slope is very close to the one created by the real motion of the target, but with opposite polarity, causing almost complete flattening of the phase as a function of time. This corresponds closely to the case φDoppler(t) = -φMetasurface(t), however slight oscillations of the moving platform in the experiment prevented perfect cancelation. This modulation frequency causes the structure to appear almost stationary to the radar, which means the MTI filter considers it as clutter, concealing the presence of the target as will be shown ahead. Ut is noted that the amplitude of the reflected signal is modulated as well, as shows in <FIG>. Since radar systems tend to rely on the phase of reflected echoes and not their intensity, this amplitude modulation does not affect the results ahead.

One of the most popular methods of extracting Doppler information from the phase difference of consecutive pulses is by using a fast Fourier transform (FFT) filter bank. This method, as many other alternatives, serves to average out the signal, which is reminiscent of finding the fittest linear approximation to the function. For real targets in field conditions the phase difference is unlikely to be perfectly linear, partially owing to the fact that the target may fluctuate, e.g. change direction rapidly, enter an area that degrades SNR conditions, or have different moving parts that add additional modulation to the reflected echoes, termed micro-Doppler <NUM>-<NUM>.

The phases in the plots shown on <FIG> were multiplied by the imaginary unit j, exponentiated and put through an FFT filter bank in order to estimate the velocity of the target from the peak of the FFT output, as shown on <FIG> shows the output of the FFT for various modulation frequencies with the green dashed line being the output for the moving target without any modulation of the metasurface. The two orange-shade lines [curves (ii) and (iii)] correspond to modulation frequencies of ±<NUM>. For the positive modulation, the output estimates a larger velocity than the ground truth, as shown by the orange dashed outline and in correspondence with the phase in <FIG>. Conversely, for the negative modulation the target appears slower to the radar as well as heading in the opposite direction (towards it), in correspondence with the phase profile of <FIG>. The red curve [curve (iv)] of <FIG> shows that a modulation frequency of -<NUM> shifts the perceived velocity of the target to <NUM>, making it appear as stationary to the radar as any of the surrounding clutter. This result is in correspondence with <FIG>. <FIG> shows the same results as <FIG> but with the additional processing of an MTI filter implemented as a two-pulse canceller, performed before the FFT processing. <MAT> where xk = eiφk and φk are samples of the phase as shown on <FIG>. Intuitively, the above filter removes any static contributions of the signal, leaving only time-dependent components that change in between the samples. This can be seen more rigorously by taking the Z-transform of Eq.<NUM>. The transfer function of the two pulse canceler MTI is <MAT> which has zeros at normalized discrete frequencies of <NUM>πn, where n=<NUM>,<NUM>,<NUM> and so on. This means that the DC contribution of the received signal, i.e. the clutter, is removed by the filter. The maximal passband is achieved at normalized discrete frequencies of π(<NUM>n + <NUM>), meaning it is preferable to down-sample the phases in xk in a way that would allow the expected Doppler frequencies to pass without significant attenuation. This approach was undertaken to produce <FIG>, where the output of the FFT, corresponding to metasurface modulation frequency of -<NUM>, is completely attenuated. The curves corresponding to modulation frequencies of -<NUM> and <NUM> are also slightly attenuated by the filter due to their relatively low velocity, while the output of the FFT remains virtually unchanged for the modulation frequency of <NUM>. Small side-lobes remain due to the fact the phase is not completely flat, as shown in <FIG>. It is observable that the output of the FFT is somewhat coarse, which is the result of processing only a relatively small time window (and therefore a small amount of samples as per the transfer function restrictions discussed above). This constrain solely relates to the experimental setup, since a short conveyor belt was used and the observation time was limited. It is noted that typical settings for airborne target detection are tuned to remove anything moving slowly in the scene, since this can be associated with wind, birds and other nonstationary clutter that will otherwise produce false alarms.

While <FIG> show the output of the FFT for a select group of metasurface modulation frequencies, it is desirable to demonstrate full control over the velocity measured by the radar. To do so, the experiments mentioned earlier were repeated for <NUM> different real velocities of the cloaked target, as well as <NUM> different modulation frequencies, producing <FIG>. Two lines are drawn and marked "real velocity line" and "invisibility line". The first indicates the velocity of the target without applying any modulation to the metasurface. The second reveals the required modulation frequency of the metasurface in order to conceal the target (by shifting its effective Doppler signature to <NUM>). A linear relation between the modulation frequency and the measured velocity is observed, in accordance with Eq. <NUM>. Some of the data points represented by the green x's on <FIG> are the same ones used in <FIG>, as well as in <FIG>.

The problem of radar invisibility was revisited from a signal processing point of view. While quite a few efforts in the field concentrate on the scattering suppression approach, this Example demonstrates how to use loopholes in radar's post processing for concealing a target. Specifically, any real radar system uses a filter bank to improve SNR and cope with extremely small echoes from the targets of interest. One of the most dominant sources of noise in the receiver originates from static or slowly moving clutter, which causes substantial backscattered electromagnetic energy. Moving target indicators, regardless of their particular implementation, rely on Doppler information to isolate an object of interest from a clutter. The approach of the present embodiments uses this extremely powerful technique as a weakness. Temporally modulated metasurface covers were shown to be capable of imprinting arbitrary Doppler shifts on to the backscattered echoes. More importantly, they were shown to be able to compensate for real Doppler shifts caused by genuine motion of the target, causing a moving target to look like a stationary one. Any unclassified radar system, operating under real outdoor conditions filters out zero-Doppler targets, even though they reflect quite a substantial amount of energy. This phase-based realization has significant advantages over amplitude approaches, which aim on suppression of reflected energy. For example, reducing the reflection by a factor of <NUM> leads to only 3dB SNR reduction, which is quite negligible for most radar systems, operating with 90dB and even higher dynamic ranges. Only dramatic reduction of target's reflection coefficient for a broadband, mixed polarization, all angle of incidence causes successful concealment of the target. Meanwhile, the phase approach of the present embodiments already demonstrates perfect cloaking of macroscopic objects.

Claim 1:
A cloaking system (<NUM>), comprising:
a structure (<NUM>) having a plurality of resonators (<NUM>) having a controllable resonance frequency, wherein said resonators (<NUM>) are arranged to collectively ensure that variation of said resonance frequency over a predetermined range of resonance frequencies generates a phase shift between an electromagnetic wave (<NUM>) incident on said structure (<NUM>) and an electromagnetic wave (<NUM>) scattered off said structure (<NUM>); and
a controller (<NUM>) configured for controlling said resonance frequency to provide a time-varying resonance frequency having a temporal function which comprises a linear time-dependence;
characterized in that said controller (<NUM>) is configured to receive velocity data characterizing a motion of a vehicle (<NUM>) and to select said time-varying resonance frequency based on said velocity data.