Patent Description:
A Global Navigation Satellite System (GNSS) is a satellite navigation system that provides the geographic location of a user's receiver anywhere in the world. The Global Positioning System (GPS) is an example of a GNSS. It is owned and operated by the US Air Force. GNSS/GPS systems may coexist with terrestrial networks in adjacent bands where satellite frequencies are used and/or reused terrestrially. It may be necessary to reduce or prevent radiation by the terrestrial network and the user equipments (UEs) from interfering with the GNSS/GPS communications. Specifically, GNSS/GPS receivers may need to exhibit tolerance to strong adjacent band signals from terrestrial networks.

The invention is defined by the appended independent claim <NUM>. Further refinements are defined by the dependent claims <NUM>-<NUM>.

The figures are useful to understand the invention which is described in detail by the blocks <NUM>, <NUM>, <NUM>, <NUM>, <NUM>, <NUM> of <FIG>.

The Global Navigation Satellite System (GNSS) is a satellite navigation system that provides the geographic location of a user's receiver anywhere in the world. In the United States, the Global Positioning System (GPS) is a space-based navigation system that provides location information of receivers. As used herein, the terms GNSS and GPS will be used interchangeably to represent any satellite navigation system that provides the geographic location.

GPS receivers have become ubiquitous in a variety of applications in airplanes, trains, ships, industrial equipment, farm equipment, and/or personal communication devices such as smart phones. GPS receivers may be standalone devices or integrated in another device such as a smartphone, camera, vehicle information and/or entertainment system. Conventional receivers for GPS are often not designed to be able to withstand high levels of adjacent band powers, resulting in spectrum close to the GPS frequency band(s) being underutilized. Modern communication applications such as voice, video, and data applications are increasingly hungry for spectrum additional spectrum. Furthermore, GPS systems may coexist with terrestrial networks in adjacent bands where satellite frequencies are used and/or reused terrestrially. It may be necessary to mitigate overload interference to GPS receivers from relatively high power radiation by the terrestrial network and the user equipment in the adjacent bands. Specifically, GPS receivers may need to exhibit tolerance to strong adjacent band signals from terrestrial networks. This tolerance may be achieved by highly frequency selective filters, which may be subject to frequency drift and a consequent degradation in the rejection of the adjacent band terrestrial signals. Therefore, it may be advantageous for the highly frequency selective filters to have methods for mitigating, or compensating for, the frequency drift.

Various embodiments described herein may arise from the recognition that underutilization of frequency bands adjacent to GPS frequency bands is an undesirable side effect of poor conventional GPS receiver designs that offer limited radio frequency (RF) filtering in the front end. Conventional GPS receiver designs thus may not sufficiently filter GPS frequency sidebands. Frequency drift due to temperature variation of RF front end filters exasperates poor filtering of the GPS frequency sidebands. A frequency drift compensation system/method for the GPS receiver, according to various embodiments described herein, may enable GPS receivers to provide greater rejection of adjacent band powers. In some embodiments, the frequency drift compensation system/method may include a self-calibrating closed loop producing a GPS receiver design that is more tolerant of strong adjacent band signals. This additional robustness stems from the ability in the design to use filters with greater frequency selectivity, which might otherwise be impractical owing to frequency response drifts caused by temperature and manufacturing variations. Additionally, Intermediate Frequency (IF) filter center frequencies may drift due to the environmental effects and may be compensated based on detecting shifts in detected pilot signal powers. For example, the frequency drift compensation system/method may include a pilot signal injector circuit that is configured to inject a pilot signal into the intermediate frequency signal to obtain a composite signal, and a drift frequency compensator that is configured to provide a frequency drift control signal to the local oscillator based on frequency drift identified in the composite signal. The local oscillator changes the local oscillator frequency responsive to the frequency drift control signal.

A basic approach is described first, with specific embodiments described subsequently. One or more pilot signals are added locally to the received signal inside the receiver so that the pilot signals are located (spectrally) on either the lower slope, or the upper slope or both, of the cumulative bandpass frequency response to the receiver. Drift of the cumulative frequency response due to temperature or manufacturing variations is sensed as follows. The time-averaged received power level, or levels, of the one or more pilot signals are measured by a processor in the receiver. The said measured values are operationally processed to optimally estimate the frequency response drift. The estimated drift is used to adjust the receiver's local oscillators' frequencies, responsive to which, an Intermediate Frequency (IF) of the receiver is adjusted. As a result of adjusting the IF, the drifted cumulative frequency response of the receiver once again optimally spans, or overlaps with, the spectrum of the desired signal.

Referring now to <FIG>, a block diagram of a frequency drift compensation system/method <NUM> for a GPS receiver is illustrated, according to some embodiments described herein. A received GPS signal <NUM> is input to a first mixer <NUM>. The first mixer <NUM> is responsive to the received GPS signal <NUM> and to a first local oscillator <NUM> at a first local oscillator frequency to down-convert the received GPS signal <NUM> into an intermediate frequency signal <NUM>. The intermediate frequency signal <NUM> is filtered by filter <NUM>, which may be, in some embodiments, a bandpass filter. The filter <NUM> may have a center frequency that drifts as function of temperature. In some embodiments, a pilot signal generator <NUM> is configured to generate a first pilot signal <NUM> at a first pilot signal frequency and a second pilot signal <NUM> at a second pilot signal frequency. The frequency drift compensation system <NUM> includes a second local oscillator <NUM> that is configured to generate a second local oscillator signal <NUM> at a second local oscillator frequency. A mixer <NUM> that is responsive to the first pilot signal <NUM> and to the second local oscillator signal <NUM> may be filtered by filter <NUM> to generate a first offset pilot signal <NUM> at a first offset pilot frequency. A mixer <NUM> that is responsive to the second pilot signal <NUM> and to the second local oscillator signal <NUM> may be filtered by filter <NUM> to generate a second offset pilot signal <NUM> at a second offset pilot frequency. A summer <NUM> is configured to add the first offset pilot signal <NUM> and the second offset pilot signal <NUM> to the intermediate frequency signal <NUM> or to a filtered version of the intermediate frequency signal <NUM> output by filter <NUM> to obtain a composite signal <NUM>.

Still referring to <FIG>, in some embodiments, a processor <NUM> is configured to detect frequency drift in the first offset pilot signal <NUM> and the second offset pilot signal <NUM> responsive to the composite signal <NUM> and to generate a frequency drift control signal <NUM> to compensate for the frequency drift. The processor <NUM> may be coupled to a memory <NUM> that stores information related to the various signals described herein for use in computation of the frequency drift control signal <NUM>. The frequency drift control signal <NUM> is configured to compensate for the temperature-based drift associated with the IF filters in the processor <NUM>. In some embodiments, filter <NUM> may include a surface acoustic wave (SAW) filter. Processor <NUM> detects frequency drift by determining a first average pilot power associated with the first pilot signal <NUM> based on the composite signal <NUM> and the first pilot signal <NUM>, and/or by determining a second average pilot power associated with the second pilot signal <NUM> based on the composite signal <NUM> and the second pilot signal <NUM>. A difference between the second average pilot power and the first average pilot power may be used to determine frequency drift by the processor <NUM>.

Still referring to <FIG>, the first local oscillator <NUM> adjusts the first local oscillator frequency and/or the second local oscillator <NUM> adjusts the second local oscillator frequency, responsive to the frequency drift control signal <NUM>. The first local oscillator <NUM> may adjust the first local oscillator frequency by performing operations such as incrementing and/or decrementing the first local oscillator frequency based on the frequency drift control signal <NUM>. The second local oscillator <NUM> may adjust the second local oscillator frequency by incrementing and/or decrementing the second local oscillator frequency based on the frequency drift control signal <NUM>.

Further referring to <FIG>, the processor <NUM> detects frequency drift and generates the frequency drift control signal <NUM> based on various criteria. In some embodiments, the second local oscillator signal <NUM> may be fed back to the processor <NUM>. The processor <NUM> may generate the frequency drift control signal <NUM> to indicate increasing the first local oscillator frequency and/or increasing the second local oscillator frequency, when the difference between the second average pilot power and the first average pilot power is higher than a fixed or variable threshold value. The processor <NUM> may generate the frequency drift control signal <NUM> to indicate decreasing the first local oscillator frequency and/or decreasing the second local oscillator frequency, when the difference between the second average pilot power and the first average pilot power is less than a fixed or variable threshold value. The processor <NUM> may generate the frequency drift control signal <NUM> to indicate no change to the first local oscillator frequency and/or no change to the second local oscillator frequency, when the difference between the second average pilot power and the first average pilot power is less than a fixed or variable threshold value and the difference between the second average pilot power and the first average pilot power is higher than a fixed or variable threshold value. In some embodiments, the fixed or variable threshold values may be the same. In some embodiments, the first local oscillator <NUM> increases the first local oscillator frequency by a first fixed frequency increment when the frequency drift control signal <NUM> indicates increasing the first local oscillator frequency. The second local oscillator <NUM> increases the second local oscillator frequency by a second fixed frequency increment when the frequency drift control signal <NUM> indicates increasing the second local oscillator frequency. Similarly in some embodiments, the first local oscillator <NUM> decreases the first local oscillator frequency by a third fixed frequency decrement when the frequency drift control signal <NUM> indicates decreasing the first local oscillator frequency. The second local oscillator <NUM> decreases the second local oscillator frequency by a fourth fixed frequency decrement in response to the frequency drift control signal <NUM> indicating decreasing the second local oscillator frequency.

Still referring to <FIG>, in some embodiments, the local oscillators <NUM>, <NUM> are adjusted based on the slope of a transition band of one or more filters in the receiver. The first local oscillator <NUM> increases the first local oscillator frequency by a first variable frequency increment of ΔP/(Q1+Q2) when the frequency drift control signal <NUM> indicates increasing the first local oscillator frequency. The second local oscillator <NUM> increases the second local oscillator frequency by a second variable frequency increment of ΔP/(Q1+Q2) when the frequency drift control signal <NUM> indicates increasing the second local oscillator frequency. In some embodiments, the first local oscillator <NUM> decreases the first local oscillator frequency by a third variable frequency decrement of ΔP/(Q1+Q2) when the frequency drift control signal <NUM> indicates decreasing the first local oscillator frequency. The second local oscillator <NUM> decreases the second local oscillator frequency by a fourth variable frequency decrement of ΔP/(Q1+Q2) when the frequency drift control signal <NUM> indicates decreasing the second local oscillator frequency. As used above, ΔP equals the difference between the second average pilot power and the first average pilot power, Q1 indicates the slope for a lower transition band of one or more filters in the receiver, and Q2 indicates the slope for an upper transition band of one or more filters in the receiver. The first average pilot power may be averaged over a time period that is at least twice a period of the intermediate frequency signal <NUM>. The second average pilot power may be averaged over a time period that is at least twice the period of the intermediate frequency signal <NUM>.

To better explain the concept of the above frequency plan, example frequencies are shown on <FIG>. These frequencies are not unique to the present inventive concepts and are provided merely as examples. Other frequency plans are feasible, conforming to the methods and systems taught here, and are covered by this invention disclosure.

<FIG> is a flowchart of operations for a GPS system/method. The GPS receiver may include a first mixer that is responsive to a received GPS signal. The GPS receiver may include a first local oscillator at a first local oscillator frequency to down-convert the received GPS signal into an intermediate frequency signal. The GPS receiver may include a filter that is configured to filter the intermediate frequency signal. Referring now to <FIG>, a GPS signal may be received at block <NUM>. At block <NUM>, a first pilot signal and a second pilot signal are generated that are at pilot signal frequencies such as, for example, <NUM> and <NUM>. At block <NUM>, a second local oscillator frequency signal is generated at a second local oscillator frequency. At block <NUM>, the first local oscillator frequency signal is offset by the first pilot signal to obtain a first offset pilot frequency signal at a first offset pilot frequency. For example, the first local oscillator frequency signal may be at <NUM> and the first pilot signal may be at <NUM> to produce the first offset pilot frequency signal at <NUM>. At block <NUM>, the second local oscillator frequency signal is offset by the second pilot signal to obtain a second offset pilot frequency signal at a second offset pilot frequency. For example, the second local oscillator frequency signal may be at <NUM> and the second pilot signal may be at <NUM> to produce the second offset pilot frequency signal at <NUM>. At block <NUM>, the filtered offset pilot frequency signal is added to a down-converted GPS signal that is based on the GPS frequency signal in the receiver of the GPS system to produce a composite signal. At block <NUM>, the frequency drift is determined based on the composite signal. At block <NUM>, a frequency drift control signal is generated based on the detected pilot signal power to control the local oscillators and to compensate the frequency drift of one or more IF filters in the receiver for the GPS system.

Conventional L band GPS/GNSS receivers are not usually designed to tolerate high power level adjacent band carriers. As a result, receiver performance may be severely degraded in the presence of high power level adjacent band carriers. Some receiver designs have considered the adjacent band carriers/interferers. However, the assumed spectrum location of these carriers are tens, if not hundreds of megahertz from the GNSS band edges. Due to these limitations with respect to adjacent band carriers, the use of spectrum close to the GNSS band is typically under-utilized in terms of power and bandwidth, resulting in inefficient use of valuable L band spectrum.

Furthermore, received GNSS signals from satellites are of very low power. Therefore, GNSS receivers are designed with emphasis on minimizing the Noise Figure (NF) by not incurring excessive losses in front end bandpass filters. Hence, conventional GPS receivers offer limited use of adjacent band carriers. Conventional receiver architectures are illustrated in <FIG>. Although <FIG> illustrate single IF stages, these conventional receiver architectures may also utilize multiple IF stages.

<FIG> is a block diagram of a conventional single intermediate frequency (IF) Global Navigation Satellite System (GNSS) receiver architecture. Referring to <FIG>, the GNSS signal received at the antenna <NUM> through the antenna interface <NUM> is amplified by the Low Noise Amplifier (LNA) <NUM>. The signal is filtered by a surface acoustic wave (SAW) filter <NUM> and down-converted by a mixer <NUM> to a non-zero Intermediate frequency (IF). The down-conversion uses a local oscillator <NUM> that may provide a signal at a frequency at an offset from the GNSS center frequency <NUM> (i.e. <NUM> + XX MHz). The IF signals are further amplified by one or more amplifiers <NUM>, filtered by additional SAW filters <NUM> and <NUM> and then may be down-converted by a Complex IF down-converter <NUM> to baseband (zero IF) in-phase "I" and quadrature-phase "Q" analog signals. The I and Q components of the baseband signal, when viewed together, may be referred to as a complex baseband signal. The complex baseband signal is filtered once again by a low pass filter <NUM>. The filtered complex baseband signal is digitized by the Analog to Digital Converter (ADC) <NUM> for further processing by a baseband Digital Signal Processor (DSP) <NUM>. The dual down-conversion by in-phase and quadrature-phase local oscillators may be referred to as complex down-conversion.

Regarding front end design drivers, key factors determining the receiver's inband performance may include: (<NUM>) minimizing the Noise Figure (NF), (<NUM>) achieving an optimum RF/IF gain and (<NUM>) maximizing the linearity of amplifiers. These key factors may be prioritized in the indicated order. RF bandpass filters are used to reject strong adjacent band signals so as not to overload nonlinear devices, such as amplifiers and mixers, in subsequent stages. However, because of NF considerations, the bulk of the frequency selective components are typically placed after the LNA <NUM>. In some designs, limited frequency selectivity is usually assigned to filters before the LNA <NUM> as the frequency selectivity of a bandpass filter may be directly related to its insertion loss. Sometimes, in conventional designs, no bandpass filter precedes the first LNA, as illustrated in <FIG>, although a low selectivity preselector filter may be included.

<FIG> is a block diagram of a conventional direct down conversion (zero IF) GNSS receiver architecture. Referring to <FIG>, the GNSS signal received at the antenna <NUM> through the antenna interface <NUM> is amplified by the Low Noise Amplifier (LNA) <NUM>. The signal is filtered by a surface acoustic wave (SAW) filter <NUM>. Amplifiers <NUM>, <NUM>, and/or <NUM> may provide the necessary gain of the GNSS signal. Filters <NUM> and/or <NUM> attenuate the adjacent band signals to protect the amplifiers <NUM>, <NUM>, and/or <NUM> from overload. The output of amplifier <NUM> may be filtered by an additional SAW filter <NUM>, after which the signal may be down-converted by a Complex IF down-converter <NUM> to baseband (zero IF) in-phase "I" and quadrature-phase "Q" analog signals. The I and Q components of the baseband signal, when viewed together, may be referred to as a complex baseband signal. The complex baseband signal may be filtered once again by a low pass filter <NUM>. The filtered complex baseband signal may be digitized by the Analog to Digital Converter (ADC) <NUM> for further processing by a baseband Digital Signal Processor (DSP) <NUM>. The filter <NUM> before the ADC determines the noise power bandwidth since filters <NUM>, <NUM>, and <NUM> usually have successively lower bandwidth. The sampling frequency of a Sampling & Hold circuit in the ADC <NUM> is selected to avoid spectral overlapping (i.e. aliasing) of the complex baseband signals after sampling. The sampled I and Q signals are processed by the DSP <NUM> to perform pseudo range measurements and other functions.

The conventional GPS/GNSS receiver design of <FIG> may have several shortcomings. These conventional design approaches may offer limited filtering in the front end of the receiver. When high level adjacent band carriers/interference are present at the antenna input, the nonlinear components in the receiver may cause gain compression and generate in-band intermodulation products as shown in <FIG> illustrates the gain compression and intermodulation noise in the RF front end of the conventional receiver design of <FIG>. The nonlinearity is contributed mainly by amplifiers such as LNA <NUM> in <FIG> or LNA <NUM> in <FIG> that are overdriven into their non-linear operating region. However, mixer <NUM> of <FIG> may also make contributions to the nonlinearity of the receiver. Receiver gain compression reduces the power of the desired GNSS signals relative to the background noise and interference. The intermodulation noise generated in the front end of a receiver, if it falls in GNSS band, may not be easily filtered out in the subsequent stages.

Furthermore, in conventional designs, the selectivity of the RF filters, especially those using surface acoustic wave (SAW) technology, is limited by temperature stabilities. The temperature instability of the frequency response of SAW filters can presently be as high as -<NUM> ppm/<NUM>C to -<NUM> ppm/°C, causing <NUM> to <NUM> frequency drift over the typical environmental temperature ranges. Hence, there has not been much interest in substantially increasing the frequency selectivity of RF/IF SAW filters. Therefore, the aggregate selectivity of filters beyond the first few elements of the RF stage of a receiver have traditionally not determined how close a strong adjacent band signal could be to the edge of the GNSS band. As a result, traditional GNSS receivers are generally incapable of tolerating the high levels of adjacent band carriers that they would have tolerated had the frequency drift of the filters been compensated.

One way to compensate for SAW filter drift due to temperature variation, as used in conventional receiver designs, is to down-convert GNSS signals to a lower Intermediate Frequency (IF), implement an IF bandpass filter by a cascade of SAW/ceramic filters, and then up-convert the IF back to the original GNSS frequencies. One motivation for this design approach is that the temperature drift of filters at lower IFs is smaller than at higher RFs. Additionally, at lower frequencies, it may be easier to realize practical bandpass filters with improved selectivity than at higher frequencies. This type of filter drift compensation may be performed by selecting Local Oscillator (LO) frequencies of up/down-converters based on a measured temperature inside the GNSS receiver and corresponding values in a stored look-up table. The look-up table may define the LO frequency as a function of temperature. This use of a temperature dependent look-up table arrangement adjusts the IF so that it is approximately at the center of the shifted frequency response of the filters over the expected range of ambient temperatures. This solution is an open loop solution and may have one or more of the following limitations: (<NUM>) the manufacturer has to create an accurate calibration table for "filter selectivity shift versus temperature" for each GNSS receiver during the testing process. A single table may not suffice for all receivers built according to a particular design as different samples of the same filter may have different temperature coefficients; and (<NUM>) this solution requires accurate ambient temperature measurement inside the GNSS receiver.

In order to address the above described limitations of conventional GPS receiver designs, frequency drift compensation systems are proposed, according to various embodiments described herein. According to various embodiments described herein, the GNSS receiver design can enable the reception of navigation signals in the L1 GNSS band, including GPS, Galileo, BeiDou and/or GLONASS, in the presence of strong adjacent band signals with a guard band relative to the edge of the GNSS band of less than <NUM>. The adjacent band signals may be terrestrial LTE signals from base stations operating with an Equivalent Isotropically Radiated Power (EIRP) of <NUM> dBW, and/or handsets operating with an EIRP of -<NUM> dBW.

According to various embodiments described herein, with judicious selection of commercially available Off-The-Shelf (COTS) amplifiers and highly frequency-selective but low loss RF filters, placement of these amplifiers and RF filters in the receiver chain, and/or using self-calibrating closed loop methods, a wideband GNSS receiver may be designed which is more tolerant of strong, close in frequency, adjacent band signals than was possible with conventional designs.

<FIG> is a block diagram of an RF front end of a wide band GNSS receiver, according to various embodiments described herein. Referring to <FIG>, the GNSS signal received at the antenna <NUM> through the antenna interface <NUM> are first filtered by a low insertion loss, highly selective, preselect RF band pass filter <NUM> implemented with Ceramic, SAW, and/or bulk acoustic wave (BAW) technology. Even with a superior bandpass filter, relatively moderate frequency selectivity can be achieved by this filter to produce minimize insertion loss. More filtering may be provided down the receiver chain. The filtered signals are then amplified by a highly linear Low Noise Amplifier <NUM>, with a <NUM> dB Gain Compression Point (<NUM> dB GCP) and with low Noise Figure (NF) specifications. The LNA <NUM> may have sufficiently small signal gain to minimize the effect of thermal noise contribution from the subsequent RF/IF stages. The amplified signal may then pass though a post-select filter <NUM>, which may be a ceramic filter. The post-select filtered signal may then be amplified by LNA <NUM> before providing the signal to RF and/or IF stages of the receiver. For example, the output of LNA <NUM> may be provide as an input to mixer <NUM> of <FIG> or as an input to mixer <NUM> of <FIG>. These signals may be coupled directly to the subsequent stages of the receiver, or through an RF cable if the front end circuitry is implemented in an active antenna assembly.

<FIG> illustrates the gain compression and intermodulation noise in the RF front end of the wide band GNSS receiver design of <FIG>. Referring to <FIG>, reduced in-band intermodulation products are generated in the front end, even with the presence of strong adjacent band carriers at the antenna input. The gain compression in LNA <NUM> of <FIG> is almost non-existent and a negligible level of intermodulation products are generated in the GNSS band. The post-select filter <NUM> of <FIG> serves to further increase the frequency selectivity at close-in interference frequencies, with moderate insertion loss but with steeper frequency response when placed after LNA <NUM>. The post select filter <NUM> may be implemented with ceramic technology that offers superior frequency selectivity and center frequency stability with temperature variation. The post-select filter <NUM> and LNA <NUM>, along with characteristics of LNA <NUM> of <FIG>, produce the substantially reduced in-band intermodulation products, as illustrated in <FIG>.

<FIG> is a block diagram of the IF and baseband processing sections of a frequency drift compensation system for a GNSS receiver, according to various embodiments described herein. As illustrated in <FIG>, partially filtered GNSS signals from the RF front end of <FIG> are again filtered by filter <NUM> to provide additional rejection at the close-in frequencies and enhance the image frequency response of the receiver. As an example, the selected IF for this receiver may be <NUM> +/-<NUM>. High side LO mixing is used (LO Frequency = <NUM>). Filter <NUM> may also reduce the level of interference signal at the input of amplifier <NUM>, resulting in lower intermodulation noise generated at the output of amplifier <NUM>. Amplifier <NUM> also may provide amplification to in-band signals before down conversion to an IF by mixer <NUM>, using the local oscillator <NUM> having a frequency for "High side" injection in the mixer <NUM> of (FIF = FLO - FRF, where FIF is the intermediate frequency, FLO is the local oscillator frequency and FRF is the GNSS radio frequency). An unmodulated Continuous Wave (CW) or Direct Sequence modulated Spread Spectrum (DSSS) pilot signal generator <NUM>, which may correspond to the pilot signal generator <NUM> of <FIG>, may produce pilot signals F1 and F2 that are converted by mixer <NUM> and mixer <NUM>, which may correspond to mixer <NUM> or mixer <NUM> of <FIG>, using local oscillator LO-<NUM><NUM> to produce signals S1 and S2. S1 and S2 are bandpass filtered by bandpass filters <NUM> and <NUM>, the results of which are added by summers <NUM> and/or <NUM>, which may correspond to summer <NUM> of <FIG>, to the IF signal.

Still referring to <FIG>, the IF chain following mixer <NUM> and summers <NUM> and <NUM> includes a series of IF amplifier/SAW filter combination such as filters <NUM>, <NUM>, <NUM>, and/or <NUM>, and amplifiers <NUM>, <NUM>, <NUM>, <NUM>, and/or <NUM>. In some embodiments, the amplifiers may be identical and/or the SAW filters may be identical. The combination of a bandpass filter sandwiched by two amplifiers (for example, filter <NUM> sandwiched by amplifier <NUM> and amplifier <NUM>) may be referred to as a prototype active bandpass filter. In <FIG>, the prototype active bandpass filter is repeated two times but in other embodiments, depending on use-case specific requirements, the number of repetitions may be different. As described herein, the repeating prototype filter may increase or maximize both frequency selectivity and linearity of a composite, active bandpass filter. The prototype filter components is chosen to increase adjacent band rejection while ensuring that the amplifier operates well below its <NUM> dB GCP in the presence of strong adjacent band signals that are likely to be encountered. Selection of prototype filter components may be applied to each prototype filter separately or in combination. Constructing a composite filter from a number of atomic prototypes may help streamline the design process and facilitate component procurement.

A SAW filter's center frequency and frequency response may drift depending on the ambient temperature. If this temperature-based drift is not compensated, a guard band may need to be included in the design. The guard band translates to a minimum frequency separation for an adjacent band signal. In practice, these guard bands may be in the range of <NUM> to <NUM> relative to upper and lower band edges of the IF filter. As mentioned earlier, conventional receivers compensate for SAW filter temperature drift by an open loop approach. According to various embodiments described herein, a more robust, closed loop approach for compensating for SAW filter frequency drift is presented.

Still referring to <FIG>, the output of amplifier <NUM> is input into a gain control amplifier <NUM> to produce an amplified signal at the intermediate frequency. A mixer <NUM> responsive to local oscillator <NUM> uses the intermediate frequency signal to produce a baseband I/Q channel signal. The baseband I/Q channel signal is filtered by low pass filter <NUM> and digitized by ADC(s) <NUM> to produce, for example, a <NUM>-bit encoded signal. The digitized signal is input to a baseband processor <NUM>, that may correspond to processor <NUM> of <FIG>.

<FIG> is a block diagram of a closed loop feedback system/method for frequency drift compensation that may be used in conjunction with the IF and baseband processing sections of <FIG>. As illustrated in <FIG>, a cumulative shift in the frequency response of IF SAW filters due to temperature change is compensated by alteration of Local Oscillator frequencies from local oscillators <NUM> and <NUM>, in a closed loop feedback system. Referring to <FIG>, an intermediate frequency (IF) signal FIF is output from mixer <NUM> response to the RF front end signal FRF and a signal from a local oscillator <NUM>. The combined frequency drift of IF SAW filters is sensed by measuring two out of band, pilot signals generated by, for example, a CW or Direct Sequence Spread Spectrum (DSSS) pilot signal generator <NUM>. The pilot signals of <FIG> may correspond, for example to the pilot signals at F1=<NUM> and F2=<NUM>, as discussed with respect to <FIG>. These pilot signals are low frequency, frequency stable reference signals, up-converted to IF band by mixer <NUM> and mixer <NUM>. Mixers <NUM> and <NUM> use the local oscillator <NUM> frequency FLO2 for "high side" injection (S1 = FLO2 + F1, S2 = FLO2 - F2). SAW bandpass filters <NUM> and <NUM> at the mixer <NUM> and <NUM> outputs select proper (upper or lower side band) pilot signals for linear addition to the IF signals. Due to the previously described techniques of generating pilot signals, these signals S1 and S2 remain aligned to the IF. As used herein, "aligned" may mean that a signal maintains a fixed frequency-offset relative to another signal.

Still referring to <FIG>, the IF signal FIF is summed with signals S1 and S2 using summers <NUM> and <NUM>. The resulting output of summer <NUM> is input to a composite of IF filters <NUM>. In some embodiments, IF filters <NUM> is a cumulative representation of the frequency response of one or more IF filters in the receiver. For example, if a temperature variation causes the frequency response of the composite IF filter <NUM> to shift down in frequency by <NUM>, the nominal IF of <NUM> is also shifted down by <NUM> to <NUM> due to the LO-<NUM> frequency change. The pilot signals are also shifted down by exactly <NUM>. After shifting, the frequency offsets of S1 and S2 relative to the IF remain unchanged.

The pilot signals F1 and F2 may be Continuous Wave (CW), and/or Direct Sequence Spread Spectrum (DSSS) modulated with PN code signals S1 and S2 (for example, with a chip rate: <NUM> chips/s), injected before the first IF filter of the composite IF filters <NUM>, with a duty cycle of, for example, <NUM>/<NUM>, for an active cycle of <NUM> for every second. Other duty cycles and pilot signal repetition periods, designed for specific applications and use cases, may be chosen without departing from the teachings herein.

The spread spectrum modulation of the pilot signals F1 and F2 helps to mitigate against received, adjacent band interference that is co-channel with one or more of the pilot signals. After despreading, the pilot signal may be compressed to a narrow bandwidth, which will pass relatively little of the interference signal power. The DSSS signal may be spectrally shaped to further reduce its spectrum occupancy.

<FIG> illustrates placement of pilot signals in the receivers of <FIG>, <FIG>, and/or <FIG>. Referring to <FIG>, pilot signals are placed on the lower slope of the cumulative filter selectivity transition band and on the upper slope of the cumulative filter selectivity transition band. The pilots signals of <FIG> may correspond, for example, to first and second pilot signals <NUM> and <NUM> of <FIG>. A filter frequency shift occurs due to characteristics of the receiver. The pilot signals are received and processed by the baseband signal processor, also referred to as a DSP (Digital Signal Processor). This may be applicable to direct quadrature down conversion to digital complex baseband, as illustrated in <FIG>. However, the methods and systems described herein are equally applicable to other embodiments described herein. For the first pilot signal, <NUM> indicates power level P1 before the filter shift occurs whereas <NUM> indicates the power level after the filter shift occurs. For the second pilot signal, <NUM> indicates power level P2 before the filter shift occurs whereas <NUM> indicates the power level after the filter shift occurs.

The received time averaged power levels P1 and P2 are computed in the baseband processor, which may include a microcomputer and/or circuits running signal and protocol processing tasks. An algorithm described below and illustrated in <FIG> and/or <NUM> may determine the direction (i.e. increment or decrement) of the local oscillator frequency shifts. N may be defined as the number of signal power measurements over pre-determined time such as, for example, a few seconds. As illustrated in <FIG>, ΔP = P2 - P1 - Z (dB), where Z is a design parameter used to normalize ΔP at room temperature (i.e. ΔP = <NUM> dB at room temperature). More specifically, Z represents the value of ΔP, at the factory at the time of manufacture, with the pilot signals set at their nominal frequencies. Typically, Z would be measured in the factory at the time of manufacture and utilized by the receiver over its lifetime. Z would be specific to each receiver. By being specific to a receiver, Z represents manufacturing tolerances in the cumulative frequency responses of the filters in the receiver chain.

<FIG> and <FIG> are flowcharts for frequency correction in the closed loop frequency drift compensation system of <FIG>, <FIG>, and/or <NUM>. As illustrated in <FIG>, if ΔP is higher than or equal to a threshold value, ΔPth, (where ΔPth is a positive value) such as, for example, <NUM> dB, at a given temperature, the local oscillator LO-<NUM> frequency and/or of the local oscillator LO-<NUM> frequency may be increased by a predetermined, arbitrary frequency increment ΔF, such as <NUM>. If ΔP </= - ΔPth at a given temperature, the local oscillator LO-<NUM> frequency and/or of the local oscillator LO-<NUM> frequency may be decreased by ΔF. This process may be repeated until the residual ΔP has a magnitude that is less than ΔPth, resulting in no further local oscillator frequency correction being required.

The purpose of a non-zero ΔPth is to introduce some hysteresis in the feedback loop. Without this hysteresis, there may be excessive jitter in the local oscillator frequencies, which may introduce harmful phase/frequency modulation to the GNSS signals. To further reduce the impact on the GNSS signals, the frequency correction loop may be designed to work intermittently, such as every few seconds, depending on the expected rate of change of the ambient temperature. Alternatively, the threshold may be varied over time to reduce jitter and/or for other purposes.

The sign of ΔP (positive or negative) indicates the direction of drift (increment or decrement) of the composite IF filter's frequency response. By shifting local oscillator LO-<NUM> in the same direction as the drift, the IF is also shifted in the same direction, ultimately, after the process has converged, placing the GNSS IF signals at the approximate center of the drifted composite IF filter response. By shifting the local oscillator LO-<NUM> frequency by the same value and in the same direction as the IF shift, the zero-IF (or other low IF) center frequency of the complex baseband signal is preserved. The pilot signals are shifted by the same value and in the same direction as the IF shift, thus preserving their relative spectral position with respect to shifted IF center frequency.

Referring to <FIG>, the frequency shift, ΔF, may not be fixed and arbitrary as in <FIG>, but based on the known nominal slopes of the composite IF bandpass filter. At each step of the process, a calculated ΔF is used which is designed to take the local oscillator frequencies towards their optimum values. For example, if the slopes are defined as Q1 dB/MHz for the lower transition band, and Q2 dB/MHz for the upper transition band, as illustrated in <FIG>, ΔF = ΔP/(Q1+Q2) MHz. For example, if Q1 = <NUM> dB/MHz, Q2 = <NUM> dB/MHz, and ΔP is <NUM> dB, the ΔF used will be (<NUM>/<NUM>) = <NUM>. This will enable the final local oscillator frequencies to be achieved, ideally, in a single step, compared to <NUM> or <NUM> steps, as may be the case in the embodiment of <FIG>. If, however, due to inaccurate knowledge of the frequency response slopes of the composite IF bandpass filter, a ΔP value may remain, whose magnitude is higher than ΔPth, then the above step is repeated. This process is repeated until the residual ΔP has a magnitude that is less than ΔPth. The advantage of the embodiment of <FIG> over the embodiment of <FIG> may be more rapid convergence to the final local oscillator frequencies. The techniques described above may also compensate for variations in the frequency response of the composite IF bandpass filter due to manufacturing tolerances of the components of the filters. The various embodiments described herein may apply equally for mitigating such variations.

The flowcharts of <FIG> and <FIG> will now be described in greater detail. It is noted that blocks <NUM> to <NUM> of <FIG> and <FIG> are identical so the description of these blocks will not be repeated for each of <FIG> and <FIG>. Referring now to <FIG>, at block <NUM>, N received samples of P1 for pilot S1 are read, i.e. [P1i] = [Xi], i = <NUM> to N. At block <NUM>, P1 is calculated as P1 = (<NUM>/N)*(ΣXi). Other forms of averaging, such as with a digital lowpass filtering, may be used without departing from the essential elements of the invention. At block <NUM>, N received samples of P2 for pilot S2 are read, i.e. [P2i] = [Xi], i = <NUM> to N. At block <NUM>, P2 is calculated as P2 = (<NUM>/N)*(ΣXi). At block <NUM>, ΔP is calculated as ΔP = P2 - P1 (dB). At block <NUM>, ΔP is compared to <NUM> and if ΔP><NUM>, flow proceeds to block <NUM>. If ΔP is not ><NUM> then flow proceeds to block <NUM>. At block <NUM>, ΔP is compared to +ΔPth. If ΔP >/= +ΔPth, the flow proceeds to block <NUM>, where the local oscillator frequency is increased by ΔF. At block <NUM>, ΔP is compared to -ΔPth. If ΔP </= -ΔPth, flow proceeds to block <NUM>, where the local oscillator frequency is decreased by ΔF. If the comparisons of blocks <NUM> and <NUM> are both untrue, no frequency correction occurs, at block <NUM>. Referring now to <FIG>, at block <NUM>, ΔP is compared to +ΔPth. If ΔP >/= +ΔPth, flow proceeds to block <NUM>, where the local oscillator frequency is increased by ΔF=ΔP/(Q1+Q2). At block <NUM>, ΔP is compared to -ΔPth. If ΔP </= - ΔPth, flow proceeds to block <NUM>, where the local oscillator frequency is decreased by ΔF=ΔP/(Q1+Q2). If the comparisons of blocks <NUM> and <NUM> are both untrue, no frequency correction occurs, at block <NUM>.

Additional embodiments that may include a single pilot signal will now be discussed. Embodiments related to a single pilot signal may be feasible in cases of low IF amplifiers gain drift, such as <+/1dB gain drift. Although IF amplifiers with such a low gain drift may not be widely available today, they be envisioned in the future, and/or with the judicious use of Automatic Gain Control (AGC) loop independent of frequency control loop. The single pilot signal case may be applied to GPS systems/methods that include stable amplifier gain or a system with AGC that does not vary much over the operational temperature range. Although GPS system with stable amplifier gain may not be widely available today, they be envisioned in the future. As in the case of the dual pilot system described above, the single pilot system may measure the received level of the pilot signal at the time of manufacture, at room temperature, and utilize it over the lifetime of the receiver.

Referring now to <FIG>, a block diagram of a frequency drift compensation system/method <NUM> for a GPS receiver is illustrated, according to various embodiments described herein. A received GPS signal <NUM> is input to a first mixer <NUM>. The GPS signal <NUM> is in a GPS frequency range of <NUM>-<NUM>. The first mixer <NUM> is responsive to the received GPS signal <NUM> and to a first local oscillator <NUM> at a first local oscillator frequency to down-convert the received GPS signal <NUM> into an intermediate frequency signal <NUM>. In one example embodiment, the GPS signal <NUM> may be at a center frequency of <NUM>, the local oscillator frequency may be at a frequency of <NUM> such that the intermediate frequency signal <NUM> is at a frequency of <NUM>. The intermediate frequency signal <NUM> is filtered by filter <NUM>. In some embodiments, filter <NUM> may be a bandpass filter. In some embodiments, filter <NUM> represents a cumulative frequency response of one or more intermediate frequency filters in the system.

Still referring to <FIG>, the frequency drift compensation system/method <NUM> includes a pilot signal generator <NUM> that is configured to generate a pilot signal <NUM> at a pilot signal frequency. The frequency drift compensation system/method includes a second local oscillator <NUM> that is configured to generate a second local oscillator frequency signal <NUM> at a second local oscillator frequency. A signal from a second mixer <NUM> that is responsive to the pilot signal <NUM> and to the second local oscillator signal <NUM> may be filtered by filter <NUM> to generate an offset pilot signal <NUM> at an offset pilot frequency. A summer <NUM> is configured to add the offset pilot signal <NUM> to the intermediate frequency signal <NUM> or to a filtered version of the intermediate frequency signal <NUM> to obtain a composite signal <NUM>. In some embodiments, a processor <NUM> is configured to detect the pilot signal <NUM> power drift due to IF filter center frequency drift. The processor <NUM> generates a frequency drift control signal <NUM> to compensate for the frequency drift, based on an average value of the pilot signal power. The processor <NUM> is coupled to a memory <NUM> that stores information related to the various calibrated pilot signal powers at the drifted IF filter center frequencies described herein for use in computation of the frequency drift control signal <NUM>. The first local oscillator <NUM> is configured to adjust the first local oscillator frequency and/or the second local oscillator <NUM> is configured to adjust the second local oscillator frequency, responsive to the frequency drift control signal <NUM>.

Referring now to <FIG>, a block diagram of a frequency drift compensation system/method <NUM> for a GPS receiver is illustrated, according to other embodiments described herein. A received GPS signal <NUM> is input to a mixer <NUM>. The mixer <NUM> is responsive to the received GPS signal <NUM> and to a local oscillator <NUM> at a local oscillator frequency to down-convert the received GPS signal <NUM> into an intermediate frequency signal <NUM>. In some embodiments, a pilot signal injector <NUM> generates a pilot signal <NUM> at a pilot signal frequency. A summer <NUM> adds the pilot signal <NUM> to the intermediate frequency signal <NUM> or to a filtered version of the intermediate frequency signal <NUM> output by filter <NUM> to obtain a composite signal <NUM>. A drift frequency compensator <NUM> detects the pilot signal power and compares it to the stored values in memory (<NUM> of <FIG>) to determine the frequency shift of IF filters. The drift frequency compensator <NUM> generates a frequency drift control signal <NUM> to compensate for the frequency drift. The local oscillator <NUM> is configured to adjust the local oscillator frequency, responsive to the frequency drift control signal <NUM>.

Still referring to <FIG>, the frequency drift compensation system/method <NUM> for a radio receiver for processing of a radio signal comprising a radio frequency spectrum, the radio receiver having a cumulative frequency response, is illustrated. The frequency drift compensation system/method <NUM> includes a pilot signal injector circuit <NUM> that is configured to inject one or more local pilot signals <NUM> that sense a frequency drift of the cumulative frequency response of the radio receiver. A drift frequency compensator <NUM> is configured to adjust a local oscillator frequency of the radio receiver to compensate for the frequency drift such that the cumulative frequency response is substantially centered on the radio frequency spectrum. In some embodiments, the drift frequency compensator <NUM> may be configured to perform closed loop adjustments that do not utilize a look up table of temperature versus characteristics of filters and/or other radio receiver components.

The frequency drift compensation system/method attempts to keep the cumulative frequency response relatively centered on the desired radio signal spectrum. In order to accomplish this, intermediate frequency (IF) is adaptive, i. e the IF is a variable IF in this radio architecture. The cumulative frequency response is tracking the radio signal spectrum. In other words, if the filter's frequency response moves in a given direction on the radio signal spectrum, the variable IF is also moved in the same direction on the radio signal spectrum such that the cumulative frequency response is tracking the radio signal spectrum. A goal of the frequency drift compensation system/method is to try to track the movement of the filter by changing the IF frequency.

Referring now to <FIG>, a block diagram of a frequency drift compensation system/method <NUM> for a GPS receiver is illustrated, according to other embodiments described herein. The frequency drift compensation system/method <NUM> of <FIG> may include various components of the frequency drift compensation system of <FIG>, and like numbers will be used to designate these components. A received GPS signal <NUM> is input to a mixer <NUM>. The mixer <NUM> is responsive to the received GPS signal <NUM> and to a first local oscillator <NUM> at a first local oscillator frequency to down-convert the received GPS signal <NUM> into an intermediate frequency signal <NUM>. In some embodiments, the pilot signal injector <NUM> of <FIG> is configured to generate an offset pilot signal <NUM> at an offset pilot signal frequency. The pilot signal injector <NUM> of <FIG> includes a pilot signal generator <NUM> that is configured to generate a pilot signal <NUM>. A mixer <NUM> is responsive to the pilot signal <NUM> and to a second local oscillator <NUM> at a second local oscillator frequency to down-convert the pilot signal <NUM>. The signal output from mixer <NUM> may be filtered by filter <NUM> to generate the offset pilot signal <NUM>. A summer <NUM> is configured to add the offset pilot signal <NUM> to the intermediate frequency signal <NUM> or to a filtered version of the intermediate frequency signal <NUM> output by filter <NUM> to obtain a composite signal <NUM>. A drift frequency compensator <NUM> detects frequency drift in the pilot signal <NUM> responsive to the composite signal <NUM>. The drift frequency compensator <NUM> generates a frequency drift control signal <NUM> to compensate for the frequency drift. The first local oscillator <NUM> is configured to adjust the first local oscillator frequency and/or the second local oscillator <NUM> is configured to adjust the second local oscillator frequency, responsive to the frequency drift control signal <NUM>. The drift frequency compensator <NUM> includes a processor <NUM> coupled to a memory <NUM> that stores information related to the various signals described herein for use in computation by the processor <NUM> of the frequency drift control signal <NUM>.

<FIG> illustrates further details of the pilot signal injector <NUM> of <FIG>. Referring now to <FIG>, the pilot signal injector <NUM> of <FIG> and/or <FIG> include a pilot signal generator <NUM> that is configured to generate a first pilot signal <NUM> and a second pilot signal <NUM>. A mixer <NUM> is responsive to the first pilot signal <NUM> and to a second local oscillator <NUM> at a second local oscillator frequency to down-convert the first pilot signal <NUM>. The down-converted first pilot signal may be filtered by filter <NUM> to generate a first offset pilot signal <NUM>. A mixer <NUM> is responsive to the second pilot signal <NUM> and to the second local oscillator <NUM> at the second local oscillator frequency to down-convert the second pilot signal <NUM>. The down-converted second pilot signal may be filtered by filter <NUM> to generate a second offset pilot signal <NUM>. Summer <NUM> adds the intermediate frequency signal <NUM> or to a filtered version of the intermediate frequency signal <NUM> output by filter <NUM> to the first offset pilot signal <NUM> and/or to the second offset pilot signal <NUM> to obtain a composite signal <NUM>.

Referring again to <FIG> and <FIG>, processor <NUM> of the drift frequency compensator <NUM> determines a first average pilot power associated with the first pilot signal <NUM> based on the composite signal <NUM>. Processor <NUM> determines a second average pilot power associated with the second pilot signal <NUM> based on the composite signal <NUM>. Processor <NUM> may then determine a difference between the second average pilot power and the first average pilot power. The difference between the second average pilot power and the first average pilot power is used to generate the frequency drift control signal <NUM>. The frequency drift control signal <NUM> indicates increasing the first local oscillator frequency and/or the second local oscillator frequency, when the difference between the second average pilot power and the first average pilot power is higher than a threshold value. The frequency drift control signal <NUM> indicates decreasing the first local oscillator frequency and/or the second local oscillator frequency, when the difference between the second average pilot power and the first average pilot power is less than the threshold value. The frequency drift control signal <NUM> indicates no change to the first local oscillator frequency and/or the second local oscillator frequency, when the difference between the second average pilot power and the first average pilot power is less than the threshold value and the difference between the second average pilot power and the first average pilot power is higher than the threshold value.

Referring now to <FIG>, a block diagram of a frequency drift compensation system <NUM> for a GPS receiver is illustrated, according to other embodiments described herein. A received GPS signal <NUM> is input to a wideband filter <NUM>. The wideband filter <NUM> filters the received GPS signal <NUM> to output a filtered GPS signal <NUM>. The filter GPS signal <NUM> is down-converted by an intermediate frequency (IF) converter <NUM> to produce a down-converted GPS signal <NUM>. The frequency drift compensation system includes a pilot signal generator <NUM> that produces a pilot signal <NUM> at a pilot signal frequency. A mixer <NUM> that is responsive to the pilot signal <NUM> and a first local oscillator frequency signal <NUM> that is generated by a first local oscillator <NUM> produces an offset pilot frequency signal <NUM>. The offset pilot frequency signal <NUM> is input to a bandpass filter <NUM> that produces a filtered offset pilot frequency signal <NUM>. An adder and/or summer <NUM> sums the filtered offset pilot frequency signal <NUM> and the down-converted GPS signal <NUM> to produce a composite signal <NUM>. The composite signal <NUM> is filtered by one or more bandpass filters <NUM>. The one or more active bandpass filters <NUM> may be a series of serially connected filters that each may include a first amplifier, a second amplifier, and a frequency selective filter that is between the first amplifier and the second amplifier. The one or more active band pass filters filter the composite signal <NUM> to obtain a filtered composite signal <NUM>. A mixer <NUM> is responsive to the filter composite signal <NUM> and second local oscillator frequency signal <NUM> from a second local oscillator <NUM> to produce a baseband signal <NUM>. The baseband signal <NUM> is processed by a baseband processor <NUM> to obtain a frequency drift control signal <NUM>. The first local oscillator <NUM> is configured to adjust the first local oscillator frequency and/or the second local oscillator <NUM> is configured to adjust the second local oscillator frequency, responsive to the frequency drift control signal <NUM>.

<FIG> is a flowchart of operations for a GPS system of <FIG>, <FIG>, <FIG> and/or <NUM>. Referring now to <FIG>, a GPS signal may be received at block <NUM>. At block <NUM>, a pilot signal is generated that is at a pilot signal frequency such as, for example, <NUM>. At block <NUM> a local oscillator frequency signal is offset by the pilot signal to obtain an offset pilot frequency signal at an offset pilot frequency. For example, the local oscillator frequency signal may be at <NUM> to produce an offset pilot frequency signal at <NUM>. At block <NUM>, the offset pilot frequency signal is bandpass filtered to obtain a filtered offset pilot frequency signal. At block <NUM>, the filtered offset pilot frequency signal is added to a down-converted GPS signal that is based on the GPS frequency signal in the receiver of the GPS system to produce a composite signal. At block <NUM>, the frequency drift is determined based on the composite signal. At block <NUM>, a frequency drift control signal is generated to control the local oscillator. The frequency drift control signal is based on the oscillator drift that was determined to compensate for the frequency drift of one or more filters in the receiver for the GPS system.

Various embodiments described herein may enable compatible operation of strong, adjacent-band terrestrial signals and weak GNSS satellite signals without overloading the GNSS receivers. Strong, adjacent-band terrestrial signals may be handled with a small frequency offset of the terrestrial signals from the GNSS band (<NUM> to <NUM>) than conventional receivers. These GNSS receivers may be realized using highly frequency selective, active, one or more IF bandpass filters, a composite active bandpass filter with a cascade of a plurality of identical, prototype, and/or active bandpass filters, each using commercially available, off the shelf (COTS) components. As described herein, the frequency response drift of the cascaded prototype filters caused by temperature variations and/or manufacturing variations, may be adaptively compensated utilizing a closed loop feedback system that adjusts the IF relative to the frequency response of the composite, active, IF bandpass filter. Unlike conventional receivers, the adaptively compensated adjustment of the IF may performed without a priori knowledge (i.e. look up table) of the dependence of filter characteristics on ambient temperature.

According to various embodiments described herein, the adaptive adjustment of the IF are performed with the aide of two pilot signals at the two edges of the composite, active, IF bandpass filter. The pilot signals may be Continuous Wave (CW) or Direct Sequence Spread Spectrum (DSSS) types. The pilot signals that are used may be intermittent in time with a selected duty cycle. The IF may be adjusted when the local oscillator frequencies are in the receiver chain, by adjusting the IF immediately before the composite, active, and/or IF bandpass filters, are adjusted. The frequencies of the pilot signals may be adjusted such that they maintain constant frequency offsets relative to the variable IF. The adjustment of the local oscillator frequencies may be based on the differential levels of the received pilot signals. In some embodiments, the local oscillator frequencies may be adjusted recursively in relatively small, fixed, arbitrary steps. The recursions may be continued until the differential pilot signal level reduces to a value lower than a predetermined threshold value. In some embodiments, the local oscillator frequencies may be adjusted based on a known functional relationship between a delta-LO-frequency and differential pilot signal level, where the functional relationship depends on the slopes of frequency response rolloffs of the composite, active, and/or IF bandpass filters. This process may be repeated until the differential pilot signal level reduces to a value lower than a predetermined threshold value. A highly linear [high <NUM> dB Gain Compression Point (<NUM> dB GCP)] front-end Low Noise Amplifier with low Noise Figure (NF) specifications may be utilized in order to minimize the possibility of front-end overload and intermodulation product generation.

Various embodiments were described herein with reference to the accompanying drawings, in which embodiments of the invention are shown.

It will be understood that, when an element is referred to as being "connected", "coupled", "responsive", or variants thereof to another element, it can be directly connected, coupled, or responsive to the other element or intervening elements may be present.

Moreover, as used herein, the term "and/or" includes any and all combinations of one or more of the associated listed items.

It will be further understood that terms, such as those defined in commonly used dictionaries, should be interpreted as having a meaning that is consistent with their meaning in the context of this specification and the relevant art and will not be interpreted in an idealized or overly formal sense expressly so defined herein.

For purposes of illustration and explanation only, various embodiments of the present invention were described herein in the context of receivers that are configured to receive GPS signals. It will be understood, however, that the present invention is not limited to such embodiments and may be embodied generally in any wireless communication terminal that is configured to transmit and receive according to one or more radio access technologies.

As used herein, the term "user equipment" includes cellular and/or satellite radiotelephone(s) with or without a display (text/graphical); Personal Communications System (PCS) terminal(s) that may combine a radiotelephone with data processing, facsimile and/or data communications capabilities; Personal Digital Assistant(s) (PDA) or smart phone(s) that can include a radio frequency transceiver and a pager, Internet/Intranet access, Web browser, organizer, calendar and/or a global positioning system (GPS) receiver; and/or conventional laptop (notebook) and/or palmtop (netbook) computer(s) or other appliance(s), which include a radio frequency transceiver. As used herein, the term "user equipment" also includes any other radiating user device that may have time-varying or fixed geographic coordinates and/or may be portable, transportable, installed in a vehicle (aeronautical, maritime, or land-based) and/or situated and/or configured to operate locally and/or in a distributed fashion over one or more terrestrial and/or extra-terrestrial location(s). Finally, the term "node" includes any fixed, portable and/or transportable device that is configured to communicate with one or more user equipment and a core network, and includes, for example, terrestrial cellular base stations (including microcell, picocell, wireless access point and/or ad hoc communications access points) and satellites, that may be located terrestrially and/or that have a trajectory above the earth at any altitude.

Example embodiments were described herein with reference to block diagrams and/or flowchart illustrations of computer-implemented methods, apparatus (systems and/or devices) and/or computer program products. It is understood that a block of the block diagrams and/or flowchart illustrations, and combinations of blocks in the block diagrams and/or flowchart illustrations, can be implemented by computer program instructions that are performed by processor circuitry. These computer program instructions may be provided to processor circuitry of a general purpose computer circuit, special purpose computer circuit such as a digital processor, and/or other programmable data processor circuit to produce a machine, such that the instructions, which execute via the processor circuitry of the computer and/or other programmable data processing apparatus, transform and control transistors, values stored in memory locations, and other hardware components within such circuitry to implement the functions/acts specified in the block diagrams and/or flowchart block or blocks, and thereby create means (functionality) and/or structure for implementing the functions/acts specified in the block diagrams and/or flowchart block(s). These computer program instructions may also be stored in a computer-readable medium that can direct a computer or other programmable data processing apparatus to function in a particular manner, such that the instructions stored in the computer-readable medium produce an article of manufacture including instructions which implement the functions/acts specified in the block diagrams and/or flowchart block or blocks.

A tangible, non-transitory computer-readable medium may include an electronic, magnetic, optical, electromagnetic, or semiconductor data storage system, apparatus, or device. More specific examples of the computer-readable medium would include the following: a portable computer diskette, a random access memory (RAM) circuit, a read-only memory (ROM) circuit, an erasable programmable read-only memory (EPROM or Flash memory) circuit, a portable compact disc read-only memory (CD-ROM), and a portable digital video disc read-only memory (DVD/BlueRay).

The computer program instructions may also be loaded onto a computer and/or other programmable data processing apparatus to cause a series of operational steps to be performed on the computer and/or other programmable apparatus to produce a computer-implemented process such that the instructions which execute on the computer or other programmable apparatus provide steps for implementing the functions/acts specified in the block diagrams and/or flowchart block or blocks.

Accordingly, embodiments of the present invention may be embodied in hardware and/or in software (including firmware, resident software, micro-code, etc.) that runs on a processor such as a digital signal processor, which may collectively be referred to as "processor circuitry," "a module" or variants thereof.

Finally, other blocks may be added/inserted between the blocks that are illustrated.

Many different embodiments were disclosed herein, in connection with the following description and the drawings. Accordingly, the present specification, including the drawings, shall be construed to constitute a complete written description of all combinations and subcombinations of the embodiments described herein, and of the manner and process of making and using them, and shall support claims to any such combination or subcombination. Although the embodiments taught here are applicable to many classes of radio receivers, GPS receivers are of particular interest to this application and are used as an example embodiment in order to explain the inventive concepts. Notwithstanding the above, the embodiments are applicable to any radio receiver. There is no dependence of the embodiments on specific characteristics of the GPS signal or GPS receivers.

Claim 1:
An active filter for a radio receiver, the active filter comprising: a first prototype active band pass filter comprising
a first (<NUM>) low noise amplifier, hereinafter referred to as LNA;
a second LNA (<NUM>); and
a first bandpass filter (<NUM>) connected between the first LNA (<NUM>) and the second LNA (<NUM>);
wherein an input terminal of the first LNA (<NUM>) is responsive to a GPS signal in a GPS frequency band that has been down-converted to an intermediate frequency;
the active filter further comprising:
a second prototype active passband filter comprising
a third LNA (<NUM>) that receives a signal corresponding to the output of the second LNA (<NUM>);
a fourth LNA (<NUM>);
a second bandpass filter (<NUM>) connected between the third LNA (<NUM>) and the fourth LNA (<NUM>).