Patent Description:
The present document relates to audio coding systems which make use of a harmonic transposition method for high frequency reconstruction (HFR), and to digital effect processors, e.g. so-called exciters, where generation of harmonic distortion adds brightness to the processed signal. In particular, the present document relates to low complexity methods for implementing high frequency reconstruction.

In the patent document <CIT> the concept of transposition was established as a method to recreate a high frequency band from a lower frequency band of an audio signal. A substantial saving in bitrate can be obtained by using this concept in audio coding. In an HFR based audio coding system, a low bandwidth signal, also referred to as the low frequency component of a signal, is presented to a core waveform coder, and the higher frequencies, also referred to as the high frequency component of the signal, are regenerated using signal transposition and additional side information of very low bitrate describing the target spectral shape of the high frequency component at the decoder side. For low bitrates, where the bandwidth of the core coded signal, i.e. the low band signal or low frequency component, is narrow, it becomes increasingly important to recreate a high band signal, i.e. a high frequency component, with perceptually pleasant characteristics. The harmonic transposition defined in the patent document <CIT> performs well for complex musical material in a situation with low cross over frequency, i.e. in a situation of a low upper frequency of the low band signal. The principle of a harmonic transposition is that a sinusoid with frequency ω is mapped to a sinusoid with frequency Tω, where T > <NUM> is an integer defining the order of the transposition, i.e. the transposition order. In contrast to this, a single sideband modulation (SSB) based HFR maps a sinusoid with frequency ω to a sinusoid with frequency ω + Δω, where Δωis a fixed frequency shift. Given a core signal with low bandwidth, i.e. a low band signal with a low upper frequency, a dissonant ringing artifact will typically result from the SSB transposition, which may therefore be disadvantageous compared to harmonic transposition.

In order to reach improved audio quality and in order to synthesize the required bandwidth of the high band signal, harmonic HFR methods typically employ several orders of transposition. In order to implement a plurality of transpositions of different transposition order, prior art solutions require a plurality of filter banks either in the analysis stage or the synthesis stage or in both stages. Typically, a different filter bank is required for each different transposition order. Moreover, in situations where the core waveform coder operates at a lower sampling rate than the sampling rate of the final output signal, there is typically an additional need to convert the core signal to the sampling rate of the output signal, and this upsampling of the core signal is usually achieved by adding yet another filter bank. All in all, the computationally complexity increases significantly with an increasing number of different transposition orders.

The present invention provides a method for reducing the complexity of harmonic HFR methods by means of enabling the sharing of an analysis and synthesis filter bank pair by several harmonic transposers, or by one or several harmonic transposers and an upsampler. The proposed frequency domain transposition may comprise the mapping of nonlinearly modified subband signals from an analysis filter bank into selected subbands of a synthesis filter bank. The nonlinear operation on the subband signals may comprise a multiplicative phase modification. Furthermore, the present invention provides various low complexity designs of HFR systems.

The invention is as defined in appended claims <NUM>-<NUM>.

It should be noted that the embodiments and aspects described in this document may be arbitrarily combined. In particular, it should be noted that the aspects and features outlined in the context of a system are also applicable in the context of the corresponding method and vice versa.

The present invention will now be described by way of illustrative examples, not limiting the scope of the invention, with reference to the accompanying drawings, in which:.

The below-described embodiments are merely illustrative for the principles of the present invention for efficient combined harmonic transposition. It is understood that modifications and variations of the arrangements and the details described herein will be apparent to others skilled in the art. It is the intent, therefore, to be limited only by the scope of the impending patent claims and not by the specific details presented by way of description and explanation of the embodiments herein.

<FIG> illustrates the operation of a frequency domain (FD) harmonic transposer <NUM>. In a basic form, a Tth order harmonic transposer is theoretically a unit that shifts all signal components of the input signal to a T times higher frequency. In order to implement such transposition in the frequency domain, an analysis filter bank (or transform) <NUM> transforms the input signal from the time-domain to the frequency domain and outputs complex subbands or subband signals, also referred to as the analysis subbands or analysis subband signals. The analysis subband signals are submitted to nonlinear processing <NUM> modifying the phase and/or the amplitude according to the chosen transposition order T. Typically, the nonlinear processing outputs a number of subband signals which is equal to the number of input subband signals, i.e. equal to the number of analysis subband signals. However, it is proposed in the context of an advanced nonlinear processing to output a number of subband signals which is different from the number of input subband signals. In particular, two input subband signals may be processed in a nonlinear manner in order to generate one output subband signal. This will be outlined in further detail below. The modified subbands or subband signals, which are also referred to as the synthesis subbands or synthesis subband signals, are fed to a synthesis filter bank (or transform) <NUM> which transforms the subband signals from the frequency domain into the time domain and outputs the transposed time domain signal.

Typically, each filter bank has a physical frequency resolution measured in Hertz and a time stride parameter measured in seconds. These two parameters, i.e. the frequency resolution and the time stride, define the discrete-time parameters of the filter bank given the chosen sampling rate. By choosing the physical time stride parameters, i.e. the time stride parameter measured in time units e.g. seconds, of the analysis and synthesis filter banks to be identical, an output signal of the transposer <NUM> may be obtained which has the same sampling rate as the input signal. Furthermore, by omitting the nonlinear processing <NUM> a perfect reconstruction of the input signal at the output may be achieved. This requires a careful design of the analysis and synthesis filter banks. On the other hand, if the output sampling rate is chosen to be different from the input sampling rate, a sampling rate conversion may be obtained. This mode of operation may be necessary , e.g. when applying signal transposition where the desired output bandwidth is larger than the half of the input sampling rate, i.e. when the desired output bandwidth exceeds the Nyqvist frequency of the input signal.

<FIG> illustrates the operation of a multiple transposer or multiple transposer system <NUM> comprising several harmonic transposers <NUM>-<NUM>,. , <NUM>-P of different orders. The input signal which is to be transposed is passed to a bank of Pindividual transposers <NUM>-<NUM>, <NUM>-<NUM>,. The individual transposers <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P perform a harmonic transposition of the input signal as outlined in the context of <FIG>. Typically, each of the individual transposers <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P performs a harmonic transposition of a different transposition order T. By way of example, transposer <NUM>-<NUM> may perform a transposition of order T=<NUM>, transposer <NUM>-<NUM> may perform a transposition of order T=<NUM>,. , and transposer <NUM>-P may perform a transposition of order T=P. The contributions, i.e. the output signals of the individual transposers <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P may be summed in the combiner <NUM> to yield the combined transposer output.

It should be noted that each transposer <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P requires an analysis and a synthesis filter bank as depicted in <FIG>. Moreover, the usual implementation of the individual transposers <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P will typically change the sampling rate of the processed input signal by different amounts. By way of example, the sampling rate of the output signal of the transposer <NUM>-P may be P times higher than the sampling rate of the input signal to the transposer <NUM>-P. This may be due to a bandwidth expansion factor of P used within the transposer <NUM>-P, i.e. due to the use of a synthesis filter bank which has P times more subband channels than the analysis filter bank. By doing this the sampling rate and the Nyqvist frequency is increased by a factor P. As a consequence, the individual time domain signals may need to be resampled in order to allow for combining of the different output signals in the combiner <NUM>. The resampling of the time domain signals can be carried out on the input signal or the output signal to each individual transposer <NUM>-<NUM>, <NUM>-<NUM>,.

<FIG> illustrates an exemplary configuration of a multiple harmonic transposer or multiple transposer system <NUM> performing several orders of transposition and using a common analysis filter bank <NUM>. A starting point for the design of the multiple transposer <NUM> may be to design the individual transposers <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P of <FIG> such that the analysis filter banks (reference sign <NUM> in <FIG>) of all transposers <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P are identical and can be replaced by a single analysis filter bank <NUM>. As a consequence, the time domain input signal is transformed into a single set of frequency domain subband signals, i.e. a single set of analysis subband signals. These subband signals are submitted to different nonlinear processing units <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P for different orders of transposition. As outlined above in the context of <FIG>, nonlinear processing comprises a modification of the phase and/or amplitude of the subband signals and this modification differs for different orders of transposition. Subsequently, the differently modified subband signals or subbands have to be submitted to different synthesis filter banks <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P corresponding to the different nonlinear processing <NUM>-<NUM>, <NUM>-<NUM>,. As an outcome, P differently transposed time domain output signals are obtained which are summed in the combiner <NUM> to yield the combined transposer output.

It should be noted that if the synthesis filter banks <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P corresponding to the different transposition orders operate at different sampling rates, e.g. by using different degrees of bandwidth expansion, the time domain output signals of the different synthesis filter banks <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P need to be differently resampled in order to align the P output signals to the same time grid, prior to their summation in combiner <NUM>.

<FIG> illustrates an example configuration of a multiple harmonic transposer system <NUM> using several orders of transposition, while using a common synthesis filter bank <NUM>. The starting point for the design of such a multiple transposer <NUM> may be the design of the individual transposers <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P of <FIG> such that the synthesis filter banks of all transposers are identical and can be replaced by a single synthesis filter bank <NUM>. It should be noted that in an analogous manner as in the situation shown in <FIG>, the nonlinear processing <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P is different for each transposition order. Furthermore, the analysis filter banks <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P are different for the different transposition orders. As such, a set of P analysis filter banks <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P determines P sets of analysis subband signals. These P sets of analysis subband signals are submitted to corresponding nonlinear processing units <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P to yield P sets of modified subband signals. These P sets of subband signals may be combined in the frequency domain in the combiner <NUM> to yield a combined set of subband signals as an input to the single synthesis filter bank <NUM>. This signal combination in combiner <NUM> may comprise the feeding of differently processed subband signals into different subband ranges and/or the superposing of contributions of subband signals to overlapping subband ranges. In other words, different analysis subband signals which have been processed with different transposition orders may cover overlapping frequency ranges. In such cases, the superposing contributions may be combined, e.g. added and/or averaged, by the combiner <NUM>. The time domain output signal of the multiple transposer <NUM> is obtained from the common synthesis filter bank <NUM>. In a similar manner as outlined above, if the analysis filter banks <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P operate at different sampling rates, the time domain signals input to the different analysis filter banks <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P may need to be resampled in order to align the output signals of the different nonlinear processing units <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P to the same time grid.

<FIG> illustrates the operation of a multiple harmonic transposer system <NUM> using several orders of transposition and comprising a single common analysis filter bank <NUM> and a single common synthesis filter bank <NUM>. In this case, the individual transposers <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P of <FIG> should be designed such that both, the analysis filter banks and the synthesis filter banks of all the P harmonic transposers are identical. If the condition of identical analysis and synthesis filter banks for the different P harmonic transposers is met, then the identical filter banks can be replaced by a single analysis filter bank <NUM> and a single synthesis filter bank <NUM>. The advanced nonlinear processing units <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P output different contributions that are combined in the combiner <NUM> to yield a combined input to the respective subbands of the synthesis filter bank <NUM>. Similarly to the multiple harmonic transposer <NUM> depicted in <FIG>, the signal combination in the combiner <NUM> may comprise the feeding of differently processed outputs of the nonlinear processing units <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P into different subband ranges, and the superposing of multiple contributing outputs to overlapping subband ranges.

As already indicated above, the nonlinear processing <NUM> typically provides a number of subbands at the output which corresponds to the number of subbands at the input. The non-linear processing <NUM> typically modifies the phase and/or the amplitude of the subband or the subband signal according to the underlying transposition order T. By way of example a subband at the input is converted to a subband at the output with T times higher frequency, i.e. a subband at the input to the nonlinear processing <NUM>, i.e. the analysis subband, [(k - <NUM>/<NUM>)Δf, (k + <NUM>/<NUM>)Δf] may be transposed to a subband at the output of the nonlinear processing <NUM>, i.e. the synthesis subband, [(k - <NUM>/<NUM>)TΔf, (k + <NUM>/<NUM>)TΔf], wherein k is a subband index number and Δfif the frequency resolution of the analysis filter bank. In order to allow for the use of common analysis filter banks <NUM> and common synthesis filter banks <NUM>, one or more of the advanced processing units <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P may be configured to provide a number of output subbands which is different from the number of input subbands. In an embodiment, the number of input subbands into an advanced processing unit <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P may be roughly F/T times the number of output subbands, where T is the transposition order of the advanced processing unit and F is a filter bank resolution factor introduced below.

In the following, the principles of advanced nonlinear processing in the nonlinear processing units <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P will be outlined. For this purpose, it is assumed that.

Furthermore, it is assumed that the filter banks are evenly stacked, i.e. the subband with index zero is centered around the zero frequency, such that the analysis filter bank center frequencies are given by kΔfwhere the analysis subband index k = <NUM>,<NUM>,. LA - <NUM> and LA is the number of subbands of the analysis filter bank. The synthesis filter bank center frequencies are given by n F Δfwhere the synthesis subband index n = <NUM>,<NUM>,. LS - <NUM> and LS is the number of subbands of the synthesis filter bank.

When performing a conventional transposition of integer order T ≥ <NUM> as shown in <FIG>, the resolution factor F is selected as F = T, and the nonlinearly processed analysis subband kis mapped into the synthesis subband with the same index n = k. The nonlinear processing <NUM> typically comprises multiplying the phase of a subband or subband signal by the factor T. for each sample of the filter bank subbands one may write <MAT> where θA(k) is the phase of a sample of the analysis subband kand θS(k) is the phase of a sample of the synthesis subband k. The magnitude or amplitude of a sample of the subband may be kept unmodified or may be increased or decreased by a constant gain factor. Due to the fact that Tis an integer, the operation of equation (<NUM>) is independent of the definition of the phase angle.

If the resolution factor F is selected to be equal to the transposition order T, i.e. F = T, then the frequency resolution of the synthesis filter bank, i.e. F Δf, depends on the transposition order T. Consequently, it is necessary to use different filter banks for different transposition orders T either in the analysis or synthesis stage. This is due to the fact that the transposition order T defines the quotient of physical frequency resolutions, i.e. the quotient of the frequency resolution Δfof the analysis filter bank and the frequency resolution F Δfof the synthesis filter bank.

In order to be able to use a common analysis filter bank <NUM> and a common synthesis filter bank <NUM> for a plurality of different transposition orders T, it is proposed to set the frequency resolution of the synthesis filter bank <NUM> to F Δf, i.e. it is proposed to make the frequency resolution of the synthesis filter bank <NUM> independent of the transposition order T. Then the question arises of how to implement a transposition of order T when the resolution factor F, i.e. the quotient Fof the physical frequency resolution of the analysis and synthesis filter bank, does not necessarily obey the relation F = T.

As outlined above, a principle of harmonic transposition is that the input to the synthesis filter bank subband n with center frequency n F Δfis determined from an analysis subband at a T time lower center frequency, i.e. at the center frequency n F Δf/T. The center frequencies of the analysis subbands are identified through the analysis subband index k as kΔf. Both expressions for the center frequency of the analysis subband index, i.e. n F Δf/Tand kΔf, may be equated. Taking into account that the index n is an integer value, the expression <MAT> is a rational number which can be expressed as the sum of an integer analysis subband index kand a remainder r ∈ {<NUM>,<NUM>/T, <NUM>/T,. (T - <NUM>)/T}such that <MAT>.

As such, it may be stipulated that the input to a synthesis subband with synthesis subband index n may be derived, using a transposition of order T, from the analysis subband or subbands kwith the index given by equation (<NUM>). In view of the fact that <MAT> is a rational number, the remainder r may be unequal to <NUM> and the value k+r may be greater than the analysis subband index k and smaller than the analysis subband index k+<NUM>. Consequently, the input to a synthesis subband with synthesis subband index n should be derived, using a transposition of order T, from the analysis subbands with the analysis subband index k and k+<NUM>, wherein k is given by equation (<NUM>).

As an outcome of the above analysis, the advanced nonlinear processing performed in a nonlinear processing unit <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P may comprise, in general, the step of considering two neighboring analysis subbands with index kandk + <NUM> in order to provide the output for synthesis subband n. For a transposition order T, the phase modification performed by the nonlinear processing unit <NUM>-<NUM>, <NUM>-<NUM>,. , <NUM>-P may therefore be defined by the linear interpolation rule, <MAT> where θA(k) is the phase of a sample of the analysis subband k, θA(k + <NUM>) is the phase of a sample of the analysis subband k+<NUM>, and θS(k) is the phase of a sample of the synthesis subband n. if the remainder r is close to zero, i.e. if the value k+r is close to k, then the main contribution of the phase of the synthesis subband sample is derived from the phase of the analysis subband sample of subband k. On the other hand, if the remainder r is close to one, i.e. if the value k+r is close to k+<NUM>, then the main contribution of the phase of the synthesis subband sample is derived from the phase of the analysis subband sample of subband k+<NUM>. It should be noted that the phase multipliers T (<NUM> - r) and T r are both integers such that the phase modifications of equation (<NUM>) are well defined and independent of the definition of the phase angle.

Concerning the magnitudes of the subband samples, the following geometrical mean value may be selected for the determination of the magnitude of the synthesis subband samples, <MAT> where aS(n) denotes the magnitude of a sample of the synthesis subband n, aA(k) denotes the magnitude of a sample of the analysis subband kand aA(k + <NUM>) denotes the magnitude of a sample of the analysis subband k+<NUM>.

For the case of an oddly stacked filter bank where the analysis filter bank center frequencies are given by (k + <NUM>/<NUM>)Δfwith k = <NUM>,<NUM>,. LA - <NUM> and the synthesis filter bank center frequencies are given by (n + <NUM>/<NUM>) F Δfwith n = <NUM>,<NUM>,. LS - <NUM>, a corresponding equation to equation (<NUM>) may derived by equating the transposed synthesis filter bank center frequency (n + <NUM>/<NUM>) <MAT> and the analysis filter bank center frequency (k + <NUM>/<NUM>)Δf. Assuming an integer index kand a remainder r ∈ [<NUM>,<NUM>[the following equation for oddly stacked filter banks can be derived: <MAT>.

It can be seen that if T - F, i.e. the difference between the transposition order and the resolution factor, is even, T (<NUM> - r) and T r are both integers and the interpolation rules of equations (<NUM>) and (<NUM>) can be used.

The mapping of analysis subbands into synthesis subbands is illustrated in <FIG> shows four diagrams for different transposition orders T=<NUM> to T=<NUM>. Each diagram illustrates how the source bins <NUM>, i.e. the analysis subbands, are mapped into target bins <NUM>, i.e. synthesis subbands. For ease of illustration, it is assumed that the resolution factor F is equal to one. In other words, <FIG> illustrates the mapping of analysis subband signals to synthesis subband signals using Eq. (<NUM>) and (<NUM>). In the illustrated example the analysis/synthesis filter bank is evenly stacked, with F = <NUM> and the maximum transposition order P = <NUM>.

In the illustrated case, equation (<NUM>) may be written as n/T = k + r. Consequently, for a transposition order T=<NUM>, an analysis subband with an index k is mapped to a corresponding synthesis subband n and the remainder r is always zero. This can be seen in <FIG> where a source bin <NUM> is mapped one to one to a target bin <NUM>.

In case of a transposition order T=<NUM>, the remainder r takes on the values <NUM> and <NUM>/<NUM> and a source bin is mapped to a plurality of target bins. When reversing the perspective, it may be stated that each target bin <NUM>, <NUM> receives a contribution from up to two source bins. This can be seen in <FIG>, where the target bin <NUM> receives a contribution from source bins <NUM> and <NUM>. However, the target bin <NUM> receives a contribution from source bin <NUM> only. If it is assumed that target bin <NUM> has an even index n, e.g. n=<NUM>, then equation (<NUM>) specifies that target bin <NUM> receives a contribution from the source bin <NUM> with an index k=n/<NUM>, e.g. k=<NUM>. The remainder r is zero in this case, i.e. there is no contribution from the source bin <NUM> with index k+<NUM>, e.g. k+<NUM>=<NUM>. This changes for target bin <NUM> with an odd index n, e.g. n=<NUM>. In this case, equation (<NUM>) specifies that target bin <NUM> receives contributions from the source bin <NUM> (index k=<NUM>) and source bin <NUM> (index k+<NUM>=<NUM>). This applies in a similar manner to higher transposition order T, e.g. T=<NUM> and T=<NUM>, as shown in <FIG>.

The similar situation for the case of F = <NUM>, where equation (<NUM>) may be written as 2n/T = k + r is depicted in <FIG>. For a transposition order T=<NUM>, an analysis subband with an index k is mapped to a corresponding synthesis subband n and the remainder r is always zero. This can be seen in <FIG> where a source bin <NUM> is mapped one to one to a target bin <NUM>.

In case of a transposition order T=<NUM>, the remainder r takes on the values <NUM>, <NUM>/<NUM>, and <NUM>/<NUM> and a source bin is mapped to a plurality of target bins. When reversing the perspective, it may be stated that each target bin <NUM>, <NUM> receives a contribution from up to two source bins. This can be seen in <FIG>, where the target bin <NUM> receives a contribution from source bins <NUM> and <NUM>. If it is assumed that target bin <NUM> has index e.g. n=<NUM>, then equation (<NUM>) specifies that k=<NUM> and r = <NUM>/<NUM>, so target bin <NUM> receives contributions from the source bin <NUM> (index k=<NUM>) and source bin <NUM> (index k+<NUM>=<NUM>). However, for target bin <NUM> with index n=<NUM>, the remainder r is zero such that the target bin <NUM> receives a contribution from source bin <NUM> only. This applies in a similar manner to higher transposition order T, e.g. T=<NUM>, as shown in <FIG>.

A further interpretation of the above advanced nonlinear processing may be as follows. The advanced nonlinear processing may be understood as a combination of a transposition of a given order T and a subsequent mapping of the transposed subband signals to a frequency grid defined by the common synthesis filter bank, i.e. by a frequency gridF Δf. In order to illustrate this interpretation, reference is made again to <FIG> or <FIG>. However, in the present case, the source bins <NUM> or <NUM> are considered to be synthesis subbands derived from the analysis subbands using an order of transposition T. These synthesis subbands have a frequency grid given by TΔf. In order to generate synthesis subband signals on a pre-defined frequency gridF Δfgiven by the target bins <NUM> or <NUM>, the source bins <NUM> or <NUM>, i.e. the synthesis subbands having the frequency grid TΔf, need to be mapped onto the pre-defined frequency grid F Δf. This can be performed determining a target bin <NUM> or <NUM>, i.e. a synthesis subband signal on the frequency grid F Δf, by interpolating one or two source bins <NUM> or <NUM>, i.e. synthesis subband signals on the frequency grid TΔf. In a preferred embodiment, linear interpolation is used, wherein the weights of the interpolation are inversely proportional to the difference between the center frequency of the target bin <NUM> or <NUM> and the corresponding source bin <NUM> or <NUM>. By way of example, if the difference is zero, then the weight is <NUM>, and if the difference is TΔf then the weight is <NUM>.

In summary, a nonlinear processing method has been described which allows the determination of contributions to a synthesis subband by means of the transposition of several analysis subbands. The nonlinear processing method enables the use of single common analysis and synthesis subband filter banks for different transposition orders, thereby significantly reducing the computational complexity of multiple harmonic transposers.

In the following various embodiments of multiple harmonic transposers or multiple harmonic transposer systems are described. In audio source coding/decoding systems employing HFR (high frequency reconstruction), such as SBR (spectral band replication) specified e.g. in <CIT> which is incorporated by reference, a typical scenario is that the core decoder, i.e. the decoder of a low frequency component of an audio signal, outputs a time domain signal to the HFR module or HFR system, i.e. the module or system performing the reconstruction of the high frequency component of the audio signal. The low frequency component may have a bandwidth which is lower than half the bandwidth of the original audio signal comprising the low frequency component and the high frequency component. Consequently, the time domain signal comprising the low frequency component, also referred to as the low band signal, may be sampled at half the sampling rate of the final output signal of the audio coding/decoding system. In such cases, the HFR module will have to effectively resample the core signal, i.e. the low band signal, to twice the sampling frequency in order to facilitate the core signal to be added to the output signal. Hence, the so-called bandwidth extension factor applied by the HFR module equals <NUM>.

After generation of a high frequency component, also referred to as the HFR generated signal, the HFR generated signal is dynamically adjusted to match the HFR generated signal as close as possible to the high frequency component of the original signal, i.e. to the high frequency component of the originally encoded signal. This adjustment is typically performed by a so-called HFR processor by means of transmitted side information. The transmitted side information may comprise information on the spectral envelope of the high frequency component of the original signal and the adjustment of the HFR generated signal may comprise the adjustment of the spectral envelope of the HRF generated signal.

In order to perform the adjustment of the HFR generated signal according to the transmitted side information, the HFR generated signal is analyzed by a multichannel QMF (Quadrature Mirror Filter) bank which provides spectral QMF subband signals of the HFR generated signal. Subsequently, the HFR processor performs the adjustment of the HFR generated signal on the spectral QMF subband signals obtained from analysis QMF banks. Eventually, the adjusted QMF subband signals are synthesized in a synthesis QMF bank. In order to perform a modification of the sampling frequency, e.g. in order to double the sampling frequency from the sampling frequency of the low band signal to the sampling frequency of the output signal of the audio coding/decoding system, the number of analysis QMF bands may be different from the number of synthesis QMF bands. In an embodiment, the analysis QMF bank generates <NUM> QMF subband signals and the synthesis QMF bank processes <NUM> QMF subbands, thereby providing a doubling of the sampling frequency. It should be noted that typically the analysis and/or synthesis filter banks of the transposer generate several hundred analysis and/or synthesis subbands, thereby providing a significantly higher frequency resolution than the QMF banks.

An example of a process for the generation of a high frequency component of a signal is illustrated in the HFR system <NUM> of <FIG>. A transmitted bit-stream is received at the core decoder <NUM>, which provides a low frequency component of the decoded output signal at a sampling frequency fs. The low frequency component at sampling frequency fs is input into the different individual transposers <NUM>-<NUM>,. , <NUM>-P, wherein each single transposer corresponds to a single transposer of transposition order T = <NUM>,. , P as illustrated in <FIG>. The individually transposed signals for T = <NUM>, <NUM>,. , P are separately fed to specific instances of separate analysis QMF banks <NUM>-<NUM>,. It should be noted that the low frequency component is considered to be the transposed signal of order T = <NUM>. The resampling of the core signal, i.e. the resampling of the low frequency component at sampling frequency fs, is achieved by filtering the low frequency component using a downsampled QMF bank <NUM>-<NUM>, typically having <NUM> channels instead of <NUM> channels. As an outcome, <NUM> QMF subband signals are generated, wherein each QMF subband signal has a sampling frequency fs/<NUM>.

The impact of transposition by an order T=<NUM> on a signal at a sampling frequency fs is shown in the frequency diagrams illustrated in <FIG>. The frequency diagram <NUM> shows an input signal to the transposer <NUM>-<NUM> with a bandwidth B Hz. The input signal is segmented into analysis subband signals using an analysis filter bank. This is represented by the segmentation into frequency bands <NUM>. The analysis subband signals are transposed to a T=<NUM> times higher frequency range and the sampling frequency is doubled. The resulting frequency domain signal is illustrated in frequency diagram <NUM>, wherein frequency diagram <NUM> has the same frequency scale as frequency diagram <NUM>. It can be seen that the subbands <NUM> have been transposed to the subbands <NUM>. The transposition operation is illustrated by the dotted arrows. Furthermore, the periodic spectrum <NUM> of the transposed subband signals is illustrated in the frequency diagram <NUM>. Alternatively, the process of transposition can be illustrated as in frequency diagram <NUM>, where the frequency axis has been scaled, i.e. multiplied by the transposition factor T=<NUM>. In other words, the frequency diagram <NUM> corresponds to the frequency diagram <NUM> at a T=<NUM> time higher scale. The subband segments <NUM> each have bandwidths twice that of the segments <NUM>. This results in an output signal of the transposer <NUM>-<NUM> which has a T=<NUM> times higher sampling rate than the input signal, i.e. a sampling rate of 2fs, while the time duration of the signal remains unchanged.

As can be seen in <FIG> and as has been outlined above, the output signal of the individual transposer <NUM>-<NUM> with transposition order T = <NUM> has a sampling frequency of 2fs. In order to generate QMF subband signals at a sampling frequency fs/<NUM>, an analysis QMF bank <NUM>-<NUM> having <NUM> channels should be used. In a similar manner, the output signal of the individual transposer <NUM>-P with transposition order T = P has a sampling frequency of Pfs. In order to generate QMF subband signals at a sampling frequency fs/<NUM>, an analysis QMF bank <NUM>-<NUM> having <NUM> · Pchannels should be used. In other words, the subband outputs from all the instances of the analysis QMF banks <NUM>-<NUM>,. , <NUM>-P will have equal sampling frequencies if the size, i.e. the number of channels for each of the analysis QMF banks <NUM>-<NUM>,. , <NUM>-P is adapted to the signal originating from the corresponding transposer <NUM>-<NUM>,. The sets of QMF subband signals at the sampling frequency fs/<NUM> are fed to the HFR processing module <NUM>, where the spectral adjustment of the high frequency components is performed according to the transmitted side information. Finally the adjusted subband signals are synthesized to a time domain signal by a <NUM> channel inverse or synthesis QMF bank <NUM>, thereby effectively producing a decoded output signal at sampling frequency 2fs from the QMF subband signals sampled at fs/<NUM>.

As has been outlined above, the transposer modules <NUM>-<NUM>,. , <NUM>-P produce time domain signals of different sampling rates, i.e. sampling rates <NUM>fs,. , Pfs, respectively. The resampling of the output signals of the transposer modules <NUM>-<NUM>,. , <NUM>-P is achieved by "inserting" or discarding subband channels in the following corresponding QMF analysis banks <NUM>-<NUM>,. In other words, the resampling of the output signals of the transposer modules <NUM>-<NUM>,. , <NUM>-P may be achieved by using a different number of QMF subbands in the subsequent respective analysis QMF banks <NUM>-<NUM>,. , <NUM>-P and the synthesis QMF bank <NUM>. Hence, the output QMF subband signals from the QMF banks <NUM>-<NUM>,. , <NUM>-P may need to be fitted into the <NUM> channels finally being transmitted to the synthesis QMF bank <NUM>. This fitting or mapping may be achieved by mapping or adding the <NUM> QMF subband signals coming from the <NUM> channel analysis QMF bank <NUM>-<NUM> to the first <NUM> channels, i.e. the <NUM> lower frequency channels, of the synthesis or inverse QMF bank <NUM>. This effectively results in a signal which is filtered by the analysis QMF bank <NUM>-<NUM> to be upsampled by a factor <NUM>. All the subband signals coming from the <NUM> channel analysis QMF bank <NUM>-<NUM> may be mapped or added directly to the <NUM> channels of the inverse QMF bank <NUM>. In view of the fact that the analysis QMF bank <NUM>-<NUM> is of exactly the same size as the synthesis QMF bank <NUM>, the respective transposed signal will not be resampled. The QMF banks <NUM>-<NUM>,. , <NUM>-P have a number of output QMF subband signals which exceeds <NUM> subband signals. In such cases, the lower <NUM> channels may be mapped to or added to the <NUM> channels of the synthesis QMF bank <NUM>. The upper remaining channels may be discarded. As an outcome of the use of a <NUM> · Pchannel analysis QMF bank <NUM>-P, the signal which is filtered by QMF bank <NUM>-P will be downsampled a factor P/<NUM>. Consequently, this resampling depending on the transposition order P will result in all transposed signals having the same sampling frequency.

In other words, it is desirable that the subband signals have the same sampling rates when fed to the HFR processing module <NUM>, even though the transposer modules <NUM>-<NUM>,. , <NUM>-P produce time domain signals of different sampling rates. This may be achieved by using different sizes of the analysis QMF banks <NUM>-<NUM>,. , <NUM>-P, where the size typically is 32T, with T being the transposition factor or transposition order. Since the HFR processing module <NUM> and the synthesis QMF bank <NUM> typically operate on <NUM> subband signals, i.e. twice the size of analysis QMF bank <NUM>-<NUM>, all subband signals from the analysis QMF banks <NUM>-<NUM>,. , <NUM>-P with subband indices exceeding this number may be discarded. This can be done since the output signals of the transposers <NUM>-<NUM>,. , <NUM>-P may actually cover frequency ranges above the Nyqvist frequency fs of the output signal. The remaining subband signals, i.e. the subband signals that have been mapped to the subbands of the synthesis QMF bank <NUM>, may be added to generate frequency overlapping transposed signals (see <FIG> b discussed below) or combined in some other way, e.g. to obtain non-overlapping transposed signals as depicted in <FIG> (discussed below). In case of non-overlapping transposed signals, a transposer <NUM>-T of transposition order T, wherein T=<NUM>,. ,P, is typically assigned a particular frequency range for which the transposer <NUM>-T exclusively generates a frequency component. In an embodiment, the dedicated frequency range of the transposer <NUM>-T may be [(T-<NUM>)B,TB] where B is the bandwidth of the input signal to the transposer <NUM>-T. In such cases, synthesis subband signals of the transposer <NUM>-T which are outside the dedicated frequency range are ignored or discarded. On the other hand, a transposer <NUM>-T may generate frequency components which overlap with frequency components of other transposers <NUM>-<NUM>,. In such cases, these overlapping frequency components are superposed in the QMF subband domain.

As indicated above, in typical embodiments, a plurality of transposers <NUM>-<NUM>,. , <NUM>-P are used to generate the high frequency component of the output signal of the HFR module <NUM>. It is assumed that the input signal to the transposers <NUM>-<NUM>,. , <NUM>-P, i.e. the low frequency component of the output signal, has a bandwidth of B Hz and a sampling rate fs and the output signal of the HRF module <NUM> has a sampling rate 2fs. Consequently, the high frequency component may cover the frequency range [B,fs]. Each of the transposers <NUM>-<NUM>,. , <NUM>-P may provide a contribution to the high frequency component, wherein the contributions may be overlapping and/or non-overlapping. <FIG> illustrates the case, where the high frequency component is generated from overlapping contributions of the different transposers <NUM>-<NUM>,. The frequency diagram <NUM> illustrates the low frequency component, i.e. the input signal to the transposers <NUM>-<NUM>,. Frequency diagram <NUM> illustrates the output signal of the <NUM>nd order transposer <NUM>-<NUM> comprising subbands in the frequency range [B,2B] which is indicated by the hatched frequency range. The frequency range [<NUM>,B] generated by the transposer is typically ignored or discarded, since this range is covered by the low frequency input signal. This is indicated by the white frequency range. Frequency diagram <NUM> illustrates the output signal of the <NUM>rd order transposer <NUM>-<NUM> covering the frequency range [B,<NUM>B] which is indicated by the hatched frequency range. In a similar manner, the transposer <NUM>-P generates an output signal covering the frequency range [B,PB] shown in frequency diagram <NUM>. Eventually, the output signals of the different transposers <NUM>-<NUM>,. , <NUM>-P and the low frequency component are mapped to the QMF subbands using analysis QMF banks <NUM>-<NUM>,. , <NUM>-P, thereby generating P sets of QMF subbands. As can be seen in frequency diagram <NUM>, the QMF subbands covering the frequency range [<NUM>,B] , reference sign <NUM>, receive a contribution only from the low frequency component, i.e. from the signal obtained from <NUM>st order transposition. The QMF subbands covering the frequency range [B,<NUM>B] , reference sign <NUM>, receive a contribution from the output signals of the transposers of order T=<NUM>,. The QMF subbands covering the frequency range [<NUM>B,<NUM>B] , reference sign <NUM>, receive a contribution from the output signals of the transposers of order T=<NUM>,. ,P, and so on. The QMF subbands covering the frequency range [(P-<NUM>)B,PB] , reference sign <NUM>, receive a contribution from the output signal of the transposer of order T=P.

<FIG> illustrates a similar scenario to <FIG>, however, the transposers <NUM>-<NUM>,. , <NUM>-P are configured such that the frequency ranges of their output signals do not overlap. Frequency diagram <NUM> illustrates the low frequency component. Frequency diagram <NUM> illustrates the output signal of the <NUM>nd order transposer <NUM>-<NUM> covering the frequency range [B,<NUM>B]. Frequency diagram <NUM> illustrates the output signal of the <NUM>rd order transposer <NUM>-<NUM> covering the frequency range [<NUM>B,<NUM>B] and frequency diagram <NUM> illustrates the output signal of the Pth order transposer <NUM>-P covering the frequency range [(P-<NUM>)B,PB]. The low frequency component and the output signals of the transposers <NUM>-<NUM>,. , <NUM>-P are fed to respective analysis QMF banks <NUM>-<NUM>,. , <NUM>-P which provide P sets of QMF subbands. Typically, these QMF subbands do not comprise contributions in overlapping frequency ranges. This is illustrated in frequency diagram <NUM>. The QMF subbands covering the frequency range [<NUM>,B] , reference sign <NUM>, receive a contribution only from the low frequency component, i.e. from the signal obtained from <NUM>st order transposition. The QMF subbands covering the frequency range [B,<NUM>B] , reference sign <NUM>, receive a contribution from the output signal of the transposer of order T=<NUM>. The QMF subbands covering the frequency range [<NUM>B,<NUM>B] , reference sign <NUM>, receive a contribution from the output signal of the transposer of order T=<NUM>, and so on. The QMF subbands covering the frequency range [(P-<NUM>)B,PB] , reference sign <NUM>, receive a contribution from the output signal of the transposer of order T=P.

<FIG> and <FIG> illustrate the extreme scenarios of completely overlapping output signals of the transposers <NUM>-<NUM>,. , <NUM>-P and of completely non-overlapping output signals of the transposers <NUM>-<NUM>,. It should be noted that mixed scenarios with partly overlapping output signals are possible. Moreover, it should be noted that the two scenarios of <FIG> and <FIG> describe systems where the transposers <NUM>-<NUM>,. , <NUM>-P are configured such that the frequency ranges of their output signals do or do not overlap. This may be achieved by applying windowing in the spectral domain of the transposers, e.g. by setting selected subband signals to zero. An alternative is to let the transposers <NUM>-<NUM>,. , <NUM>-P, in both scenarios of <FIG> and <FIG> generate wideband signals and perform the filtering of the transposed signals in the QMF subband domain by combining the subband signals obtained from the analysis QMF banks <NUM>-<NUM>,. , <NUM>-P in an appropriate manner. in the non-overlapping case, only one of the analysis QMF banks <NUM>-<NUM>,. , <NUM>-P contributes to the subband signals fed to the HFR processor <NUM> in each transposer output frequency range. For the overlapping case, pluralities of the subband signals are added before entering the HFR processor <NUM>.

A more efficient implementation of the system of <FIG> is obtained if some or all of the signals of the HRF system <NUM> are (close to) critically sampled, as shown in <FIG> and <FIG> for the HFR system <NUM>. This means that the output signal of the core decoder <NUM> and preferably also other intermediate signals of the HRF system <NUM>, e.g. the output signals of the transposers <NUM>-<NUM>,. , <NUM>-P are critically downsampled. For example, the core decoded signal at the output of the core decoder <NUM> is downsampled by a rational factor Q=M<NUM>/M<NUM>, where M<NUM> and M<NUM> are appropriately chosen integer values. The downsampling factor Q should be the largest factor that forces the input signal of bandwidth B to be close to critically sampled. At the same time, Q should be selected such that the size (<NUM>/Q) of the QMF bank <NUM>-<NUM> remains an integer. The downsampling by a rational factor Q is performed in downsampler <NUM> and yields an output signal at the sampling frequency fs/Q. In order to provide transposed signals which are also critically sampled, the transposers <NUM>-<NUM>,. , <NUM>-P preferably only output the part of the transposed signal that is relevant, i.e. the frequency range that is actually used by the HFR processor <NUM>. The relevant frequency range for a transposer <NUM>-T of transposition order T may be the range [(T-<NUM>)B,TB] for an input signal having a bandwidth B Hz in the non-overlapping case.

This means that the output from the downsampler <NUM> and the output from the transposers <NUM>-<NUM>,. , <NUM>-P are critically sampled. The output signal of the <NUM>nd order transposer <NUM>-<NUM> would have a sampling frequency fs/Q which is identical to the output signal of the downsampler <NUM>. However, it should be noted that the signal from the <NUM>nd order transposer <NUM>-<NUM> is actually a highpass signal with a bandwidth of fs/(<NUM>Q) which is modulated to the baseband, since the transposer <NUM>-<NUM> is configured such that it only synthesizes a transposed frequency range from approximately B to 2B Hz.

For transposers of larger order, e.g. transposer <NUM>-P, at least two likely scenarios are possible. The first scenario is that the transposed signals are overlapping, i.e. the lower frequency part of the Pth order transposed signal is overlapping with the frequency range of the transposed signal of order P-<NUM> (see <FIG>). In this case, the output from the critically sampled transposer <NUM>-P has the sampling frequency Sfs/Q, where S = min(P-<NUM>, 2Q-<NUM>). When S=P-<NUM>, the uppermost frequency of the Pth order transposed signal is still below the Nyqvist frequency fs of the output signal of the HFR system <NUM>, and when S=2Q-<NUM>, the Pth order transposed signal is bandwidth limited by the Nyqvist frequency fs of the output signal of the HFR system <NUM>. the sampling frequency of the output signal of the transposer <NUM>-P is never larger than (<NUM> - <NUM>/Q)fs , which corresponds to a signal covering the frequency interval from fs/(<NUM>Q) (highest frequency of lowband signal) up to the Nyqvist frequency fs. The other scenario is that the transposed signals are non-overlapping. In this case S = <NUM>, and all transposed signals have identical sampling frequencies, albeit covering different non-overlapping frequency ranges in the output signal of the inverse QMF bank <NUM>, i.e. in the output signal of the HFR system <NUM> (see <FIG>).

The effect of the described subsampling or downsampling on an output signal of the core decoder <NUM> having a bandwidth B Hz is illustrated in <FIG>. <FIG> schematically illustrates the transition of the signal from the output of the core decoder <NUM> to the output of the transposer <NUM>-<NUM> of transposition order T=<NUM>. The frequency diagram <NUM> shows the output signal of the core decoder <NUM> with bandwidth B Hz. This signal is critically downsampled in downsampler <NUM>. The downsampling factor Q is a rational value which ensures that the analysis QMF band <NUM>-<NUM> has an integer number <NUM>/Q of subbands. Furthermore, the downsampler <NUM> should provide a critically sampled output signal, i.e. an output signal having a sampling frequency fs/Q which is as close as possible to two times the bandwidth B of the core decoded signal, i.e. Q < fs/2B. Such a critically sampled signal is illustrated in the frequency diagram <NUM>. This critically sampled signal with sampling frequency fs/Q is passed to the transposer <NUM>-<NUM> where it is segmented into analysis subbands. Such a segmented signal is illustrated in frequency diagram <NUM>. Subsequently, nonlinear processing is performed on the analysis subband signals which results in a stretching of the analysis subbands to T=<NUM> times higher frequency ranges and a sampling frequency <NUM>fs/Q. This is illustrated in frequency diagram <NUM>, which alternatively may be viewed as the frequency diagram <NUM> with scaled frequency axis. It should be noted that only a subset of the transposed subbands will typically be considered in the HFR processing module <NUM>. These relevant transposed subbands are indicated in frequency diagram <NUM> as the hatched subbands which cover the frequency range [B,2B]. Only the hatched subbands may need to be considered in the transposer synthesis filter bank, and hence the relevant range can be modulated down to the baseband and the signal may be downsampled by a factor <NUM> to a sampling frequency of fs/Q. This is illustrated in frequency diagram <NUM>, where it can be seen that the signal covering a frequency range [B,2B] has been modulated into the baseband range [<NUM>,B]. The fact that the modulated signal actually covers the higher frequency range [B,<NUM>B] is illustrated by the reference signs "B" and "<NUM>B".

It should be noted that the illustrated steps of transposition (shown in frequency diagram <NUM>) and the subsequent modulation into the baseband (shown in frequency diagram <NUM>) are only shown for illustrative purposes. Both operations may be performed by assigning the hatched subbands (shown in frequency diagram <NUM>) to the synthesis subbands of a synthesis filter bank having half the number of subbands as the analysis filter bank. As an outcome of such mapping operation, the output signal shown in frequency diagram <NUM>, which is modulated into the baseband, i.e. which is centered around the zero frequency, may be obtained. In the non-overlapping scenario, the synthesis filter bank size is reduced with respect to the analysis filter bank in order to enable the achievable downsampling factor which is given by the ratio between the full frequency range [<NUM>,PB] which may be covered by the output signal of a Pth order transposer <NUM>-P and the actual frequency range [(P-<NUM>)B, PB] covered by the output signal of the Pth order transposer <NUM>-P, i.e. the factor P.

<FIG> schematically illustrates the transition of the signal from the output of the core decoder <NUM> to the output of the transposer <NUM>-<NUM> of transposition order T=<NUM> in the scenario of overlapping frequency ranges. The signal with bandwidth B shown in frequency diagram <NUM> is downsampled by a factor Q in downsampler <NUM> to yield the signal shown in frequency diagram <NUM>. The analysis subbands shown in frequency diagram <NUM> are transposed to subbands with T=<NUM> times higher frequencies. The transposed subbands are illustrated in frequency diagram <NUM>, where the sampling rate is increased from fs/Q to 3fs/Q. As outlined in the text to <FIG>, this can be viewed as a scale change of the frequency axis by a factor <NUM>. It can be seen that the frequency range of the <NUM>rd order transposer <NUM>-<NUM>, i.e. the hatched frequency range [B,<NUM>B], overlaps with the frequency range of the <NUM>nd order transposer <NUM>-<NUM>. In a similar manner to <FIG>, the hatched subbands may be fed into a synthesis filter bank of a reduced size, thereby yielding a signal comprising only frequencies from the hatched subbands. This highpass signal is thus modulated down to the baseband using a downsampling factor <NUM>/<NUM>. The resulting critically sampled output signal of the transposer <NUM>-<NUM> having a sampling frequency <NUM>fs/Q is illustrated in frequency diagram <NUM>.

In a similar manner to <FIG>, it should be noted that the transposition operation shown in frequency diagram <NUM> and the modulation into the baseband shown in frequency diagram <NUM> is performed by mapping the hatched subbands of frequency diagram <NUM> to the synthesis subbands of a synthesis filter bank of reduced size. In the overlapping scenario, the synthesis filter bank size is reduced with respect to the analysis filter bank in order to enable the achievable downsampling factor which is given by the ratio between the full frequency range [<NUM>,PB] which may be covered by the output signal of the Pth order transposer <NUM>-P and the actual frequency range [B, PB] covered by the output signal of the Pth order transposer <NUM>-P, i.e. the factor P/(P-<NUM>).

<FIG> schematically illustrates the transition of the signal from the output of the downsampler <NUM> to the output of the transposer <NUM>-P of transposition order T=P for the case that the transposed frequency range is not overlapping with the relevant frequency range of the lower order transposer T=P-<NUM>, i.e. [(P-<NUM>)B,(P-<NUM>)B]. As outlined in the context with <FIG> the downsampled signal shown in frequency diagram <NUM> is transposed by transposer <NUM>-P. The transposed subbands covering the relevant frequency range [(P-<NUM>)B,PB] are illustrated in frequency diagram <NUM> as the hatched frequency range. The subbands corresponding to the hatched frequency range are fed into the synthesis filter bank of reduced size, thereby generating a signal comprising only frequencies in the range [(P-<NUM>)B,PB]. Consequently, this highpass signal is modulated into the baseband and downsampled using a factor P. As a result, the critically sampled output signal of the transposer <NUM>-P shown in frequency diagram <NUM> is obtained. This output signal of the transposer <NUM>-P comprises frequency components of the frequency range [(P-<NUM>)B,PB]. This has to be considered when mapping the transposer output to QMF subbands for HFR processing.

<FIG> schematically illustrates the transition of the signal from the output of the downsampler <NUM> to the output of the transposer <NUM>-P of transposition order T=P for the case that the transposed frequency range is overlapping with the relevant frequency range of the lower order transposers T=<NUM>,. ,P-<NUM>, i.e. [B,(P-<NUM>)B]. As outlined in the context with <FIG> the downsampled signal shown in frequency diagram <NUM> is transposed in transposer <NUM>-P. The transposed subbands covering the frequency range [B,PB] are illustrated in frequency diagram <NUM> as the hatched frequency range. In a similar manner to <FIG>, it can be seen that the hatched subbands cover frequencies below (P-<NUM>)B. Consequently, the hatched subbands overlap with the frequency ranges of the lower order transposers <NUM>-<NUM>,. ,<NUM>-P-<NUM>. Furthermore, due to the fact that the hatched subbands cover a range larger than [(P-<NUM>)B,PB], only a reduced downsampling factor can be used. As outlined above, this downsampling factor is P/(P-<NUM>) if the frequency range covered by the output signal of the Pth order transposer <NUM>-P is [B,(P-<NUM>)B]. As a result, a downsampled output signal of the transposer <NUM>-P having a sampling frequency (P-<NUM>)fs/Q is obtained.

As already indicated above, it should be noted that the intermediate signals within the transposer <NUM>-P, i.e. notably the signals shown in the frequency diagrams <NUM>, <NUM>, <NUM>, <NUM> are not physical signals present in the HFR system shown in <FIG>. These signals have been shown for illustrative purposes and can be viewed as "virtual" signals within the transposer <NUM>-P, showing the effect of transposition and filtering in the presence of implicit downsampling.

It should be noted that in the example outlined above, the output signal from the core decoder <NUM> may possibly already be critically sampled with the sampling rate fs/Q when entering the HFR module <NUM>. This can be accomplished, e.g., by using a smaller synthesis transform size than the nominal size in the core decoder <NUM>. In this scenario, computational complexity is decreased because of the smaller synthesis transform used in the core decoder <NUM> and because of the obsolete downsampler <NUM>.

Another measure for improving the efficiency of an HFR system, is to combine the individual transposers <NUM>-<NUM>,. , <NUM>-P of <FIG> according to one of the schemes outlined in the context of <FIG> or <FIG>. As an example, instead of using individual transposers <NUM>-<NUM>,. , <NUM>-P for the different transposition orders T = <NUM>,. , P, a multiple transposer system <NUM>, <NUM> or <NUM> may be used. A possible scenario is illustrated in <FIG>, where the transposers for transposition factors T equal or larger than two are grouped together to a multiple transposer <NUM>, which may be implemented according to any of the aspects outlined in relation to <FIG>. In the illustrated example, the output from the multiple transposer <NUM> has a sampling frequency 2fs, i.e. a sampling frequency which is two times higher than the sampling frequency of the input signal to the multiple transposer <NUM>. The output signal of the multiple transposer <NUM> is filtered by a single analysis QMF bank <NUM>-<NUM> having <NUM> channels.

As outlined in the context of <FIG>, the resampling of the core signal, i.e. the resampling of the output signal of the core decoder <NUM>, may be achieved by filtering the signal using a downsampled QMF bank <NUM>-<NUM> having only <NUM> channels. As a consequence, both sets of QMF subband signals have QMF subband signals with a sampling frequency fs/<NUM>. The two sets of QMF subband signals are fed to the HFR processing module <NUM> and finally the adjusted QMF subband signals are synthesized to a time domain signal by the <NUM> synthesis QMF bank <NUM>. It should be noted that in the illustrated scenario the multiple transposer <NUM> produces a transposed time domain signal of twice the sampling rate fs. As outlined in the context of <FIG> and <FIG>, this transposed time domain signal is the sum of several transposed signals of different transposition factors T, where T is an integer greater than <NUM>. The reason for the fact that the multiple transposer <NUM> provides an output signals with a sampling frequency 2fs is that the output signal of the multiple transposer <NUM> covers the high frequency range of the output signal of the HFR module <NUM>, i.e. at most the range [B,fs], wherein B is the bandwidth of the low frequency component and fs is the Nyqvist frequency of the output signal of the HRF module <NUM>.

As outlined in the context of <FIG>, the efficiency of the HFR system <NUM> may be increased further by increasing the level of subsampling of the time domain signals, i.e. by providing critically downsampled signals, preferably at the output of the core decoder and at the output of the transposer. This is illustrated in <FIG>, where the insights outlined in the context of <FIG> and <FIG> may be applied. The output signal of the core decoder <NUM> is downsampled in the downsampling unit <NUM>, yielding a downsampled signal at a sampling frequency fs/Q. This signal is fed to the multiple transposer <NUM> and to the analysis QMF bank <NUM>-<NUM>. The output of the multiple transposer <NUM> has the sampling frequency Sfs/Q, where S = min(P-<NUM>, 2Q-<NUM>), since the output from the multiple transposer <NUM> is a combination of signals with transposition orders from T = <NUM> to P. The transposed signal is fed into an analysis QMF bank <NUM>-<NUM> of size <NUM>S/Q. In a similar manner as outlined above, the two sets of QMF subband signals are processed in the HFR processor <NUM> and eventually converted into a time domain signal using the synthesis QMF bank <NUM>.

In embodiments, the QMF bank analyzing the core coder signal, i.e. the analysis QMF bank <NUM>-<NUM> of <FIG>, may be omitted if the multiple transposer is also configured to pass through an unaltered copy of the core signal, i.e. an unaltered copy of the output signal of the core decoder. In transposer terminology this is equivalent to a transposition using the transposition factor T = <NUM>, i.e. a <NUM>st order transposition. If a <NUM>st order transposition is added to the multiple transposer system <NUM> of <FIG>, a block diagram of the modified HFR module <NUM> may be depicted as shown in <FIG>. As shown in <FIG>, the signal decoded by the core decoder <NUM> is merely used as input to the multiple transposer <NUM>, i.e. the signal decoded by the core decoder <NUM> is not passed to any additional component of the HFR module <NUM>. The multiple transposer <NUM> is configured such that its single output signal has a sampling frequency 2fs. In other words, the multiple transposer <NUM> produces a time domain signal of twice the sampling rate, wherein the time domain signal is the sum of several transposed signals of different transposition factors T, where T takes the values of <NUM> to P. This single output signal from the multiple transposer <NUM> is analyzed by a <NUM> channel QMF bank <NUM>, and the QMF subband signals are subsequently fed into the HFR processing module <NUM> which adjusts the QMF subband signals using the transmitted side information. The adjusted QMF subband signals are finally synthesized by the <NUM> channel synthesis QMF bank <NUM>.

In a similar manner to the downsampling described in the context of <FIG>, the efficiency of the HFR module <NUM> may be increased by means of subsampling of the time domain signals. Such an HFR module <NUM> is shown in <FIG>. A received bit stream is decoded by the core decoder <NUM> which provides a time domain output signal at sampling frequency fs. This time domain output signal is downsampled by a factor Q using the downsampling unit <NUM>. The downsampled signal at sampling frequency fs/Q is passed to the multiple transposer <NUM>. The output from the multiple transposer <NUM> will have the sampling frequency Sfs/Q. This time, however, the parameter S is selected as S = min(P, <NUM>Q) since the transposed signal also comprises the decoded and downsampled output signal of the core decoder <NUM>. The output signal of the multiple transposer <NUM> is segmented into QMF subband signals using an analysis QMF bank <NUM> having <NUM>/Q channels. The QMF subband signals are adjusted using the transmitted side information and subsequently merged by a synthesis <NUM> channel QMF bank <NUM>.

As mentioned above, the multiple transposers <NUM>, <NUM>, <NUM>, and <NUM> illustrated in <FIG> may be based on any of the configurations presented in the context of <FIG>. In addition, the transposer configuration illustrated in <FIG> may be used, albeit its inferior computational efficiency compared to the multiple transposer designs of <FIG>. In a first preferred embodiment, the HFR module configurations illustrated in <FIG> are used in combination with the multiple transposer described in the context of <FIG>. An exemplary mapping of the transposer analysis subbands to the transposer synthesis subbands is illustrated in <FIG>. In a second preferred embodiment, the HFR module configurations illustrated in <FIG> are used in combination with the multiple transposer described in the context of <FIG>. An exemplary mapping of the transposer analysis subbands to the transposer synthesis subbands is in this embodiment illustrated in <FIG>.

With the examples outlined in the context of <FIG>, <FIG>, and <FIG>, a general building block of a maximally decimated, or critically sampled, transposer may be identified. Such a building block <NUM> is illustrated in <FIG>. An input signal of sampling frequency fs is first processed in the factor Q downsampler <NUM>, and filtered through a transposer analysis filter bank <NUM>. The analysis filter bank has a filter bank size, or transform size, of Na, and a hopsize, or input signal stride, of δa samples. The subband signals are subsequently processed by a non-linear processing unit <NUM>, using the transposition factor T. The non-linear processing unit <NUM> may implement any of the non-linear processing outlined in the present document. In an embodiment, the non-linear processing outlined in the context of <FIG>, <FIG>, <FIG> may be performed in the non-linear processing unit <NUM>. Finally, the subband signals are assembled to a time domain signal of sampling frequency Rfs in a transposer synthesis filter bank <NUM>, wherein R is a desired re-sampling factor. The synthesis filter bank has a filter bank size, or transform size, of Ns, and a hopsize, or output signal stride, of δssamples. The expansion factor W comprising the analysis filter bank <NUM>, the non-linear processing unit <NUM> and the synthesis filter bank <NUM> is the ratio of the sampling frequencies of the output signal from the synthesis filter bank and the input signal to the analysis filter bank as <MAT>.

The filter bank, or transform sizes, Na and Ns may be related as <MAT> and the hopsizes, or signal strides, δaand δs may be related as <MAT>.

The maximally decimated, or critically sampled, transposer building block <NUM> may have either the input signal to the analysis filter bank <NUM>, or the output from the synthesis filter bank <NUM>, or both, covering exclusively the spectral bandwidth relevant for the subsequent processing, such as the HFR processing unit <NUM> of <FIG>. The critical sampling of the input signal may be obtained by filtering and possibly modulation followed by decimation of the input signal in the downsampler <NUM>. In an embodiment, the critical sampling of the output signal may be realized by mapping subband signals to a synthesis filter bank <NUM> of a minimal size adequate to cover exclusively the subband channels relevant for the subsequent processing, e.g. as indicated by equation (<NUM>). <FIG> illustrate the condition when the output from the synthesis filter bank covers exclusively the relevant spectral bandwidth and thus is maximally decimated.

A plurality of the building blocks <NUM> may be combined and configured such that a critically sampled transposer system of several transposition orders is obtained. In such a system, one or more of the modules <NUM>-<NUM> of the building block <NUM> may be shared between the building blocks using different transposition orders. Typically, a system using a common analysis filter bank <NUM>, as outlined in the context of <FIG>, may have maximally decimated output signals from the synthesis filter banks <NUM>-<NUM>,. ,<NUM>-P, while the input signal to the common analysis filter bank <NUM> may be maximally decimated with respect to the transposer building block <NUM> requiring the largest input signal bandwidth. A system using a common synthesis filter bank <NUM>, as outlined in the context of <FIG>, may have maximally decimated input signals to the analysis filter banks <NUM>-<NUM>,. ,<NUM>-P, and may also have a maximally decimated output signal from the common synthesis filter bank <NUM>. The system outlined in the context of <FIG>, preferably has both maximally decimated input signals to the analysis filter banks and maximally decimated output signals from the synthesis filter banks. In this case, the structure of the system may be merely a plurality of the transposer building blocks <NUM> in parallel. A system using both a common analysis filter bank <NUM> and a common synthesis filter bank <NUM>, as outlined in the context of <FIG>, typically has a maximally decimated output signal from the common synthesis filter bank <NUM>, while the input signal to the common analysis filter bank <NUM> may be maximally decimated with respect to the signal in which the transposition order requires the largest input signal bandwidth. For this system, the transposition factor T in equation (<NUM>) is replaced by the factor F outlined in the context to <FIG>, <FIG> and <FIG>. It should be noted that the summing units <NUM> of <FIG> and <NUM> of <FIG>, in the above scenarios may be configured to handle and combine the critically sampled subband signals from the transposer building blocks synthesis filter banks. In an embodiment, the summing units may comprise QMF analysis filter banks followed by means to combine the subband signals or time domain resampling and modulation units followed by means to add the signals.

In the present document, a multiple transposition scheme and system has been described which allows the use of a common analysis filter bank and a common synthesis filter bank. In order to enable the use of a common analysis and synthesis filter bank, an advanced nonlinear processing scheme has been described which involves the mapping from multiple analysis subbands to a synthesis subband. As a result of using a common analysis filter bank and a common synthesis filter bank, the multiple transposition scheme may be implemented at reduced computational complexity compared to conventional transposition schemes. In other words, the computational complexity of harmonic HFR methods is greatly reduced by means of enabling the sharing of an analysis and synthesis filter bank pair for several harmonic transposers, or by one or several harmonic transposers in combination with an upsampler.

Furthermore, various configurations of HFR modules comprising multiple transposition have been described. In particular, configurations of HFR modules at reduced complexity have been described which manipulate critically downsampled signals. The outlined methods and systems may be employed in various decoding devices, e.g. in multimedia receivers, video/audio settop boxes, mobile devices, audio players, video players, etc..

Claim 1:
A system configured to generate a high frequency component of an audio signal from a low frequency component of the signal, the system comprising:
- an analysis filter bank configured to provide a set of analysis subband signals from the low frequency component of the signal; wherein the set of analysis subband signals comprises at least two analysis subband signals, wherein the analysis filter bank has a frequency resolution of Δf;
- a nonlinear processing unit configured to determine a set of synthesis subband signals from the set of analysis subband signals using a transposition order P; wherein the nonlinear processing unit is configured to determine an nth synthesis subband signal of the set of synthesis subband signals from a kth analysis subband signal and a (k+<NUM>)th analysis subband signal of the set of analysis subband signals; wherein the nonlinear processing unit is configured to determine a magnitude of the nth synthesis subband signal as the product of an exponentiated magnitude of the kth analysis subband signal and an exponentiated magnitude of the (k+<NUM>)th analysis subband signal, wherein a sum of an exponent of the exponentiated magnitude of the kth analysis subband signal and an exponent of the exponentiated magnitude of the (k+<NUM>)th analysis subband signal equals one; and
- a synthesis filter bank configured to generate the high frequency component of the signal based on the set of synthesis subband signals, wherein the synthesis filter bank has a frequency resolution of FΔf, with F being a resolution factor, with F ≥ <NUM>, wherein the transposition order P is different from the resolution factor F;
wherein
- the analysis filter bank has a number LA of analysis subbands, with LA > <NUM>, where k is an analysis subband index with k = <NUM>, ..., LA - <NUM>; and
- the synthesis filter bank has a number LS of synthesis subbands, with LS > <NUM>, where n is a synthesis subband index with n = <NUM>, ... , LS - <NUM>
- the number LA of analysis subbands is equal to the number LS of synthesis subbands.