Patent Description:
The state of the art codec for parametric coding of stereo signals at low bitrates is the MPEG codec xHE-AAC. It features a fully parametric stereo coding mode based on a mono downmix and stereo parameters inter-channel level difference (ILD) and inter-channel coherence (ICC), which are estimated in subbands. The output is synthesized from the mono downmix by matrixing in each subband the subband downmix signal and a decorrelated version of that subband downmix signal, which is obtained by applying subband filters within the QMF filterbank.

There are some drawbacks related to xHE-AAC for coding speech items. The filters by which the synthetic second signal is generated produce a very reverberant version of the input signal, which requires a ducker. Therefore, the processing heavily smears the spectral shape of the input signal over time. This works well for many signal types but for speech signals, where the spectral envelope changes rapidly, this causes unnatural coloration and audible artifacts, such as double talk or ghost voice. Furthermore, the filters depend on the temporal resolution of the underlying QMF filter bank, which changes with the sampling rate. Therefore, the output signal is not consistent for different sampling rates.

Apart from this, the 3GPP codec AMR-WB+ features a semi-parametric stereo mode supporting bitrates from <NUM> to 48kbit/s. It is based on a mid/side transform of left and right input channel. In low frequency range, the side signal s is predicted by the mid signal m to obtain a balance gain and m and the prediction residual are both encoded and transmitted, alongside with the prediction coefficient, to the decoder. In mid-frequency range, only the downmix signal m is coded and the missing signal s is predicted from m using a low order FIR filter, which is calculated at the encoder. This is combined with a bandwidth extension for both channels. The codec generally yields a more natural sound than xHE-AAC for speech, but faces several problems. The procedure of predicting s by m by a low order FIR filter does not work very well if the input channels are only weakly correlated, as is e.g. the case for echoic speech signals or double talk. Also, the codec is unable to handle out-of-phase signals, which can lead to substantial loss in quality, and one observes that the stereo image of the decoded output is usually very compressed. Furthermore, the method is not folly parametric and hence not efficient in terms of bitrate.

Generally, a fully parametric method may result in audio quality degradations due the fact that any signal portions lost due to parametric encoding are not reconstructed on the decoder-side.

On the hand, waveform-preserving procedures such as mid/side coding or so do not allow substantial bitrates savings as can be obtained from parametric multichannel coders. Examples of methods for encoding/decoding or processing multichannel signals can be found in <NPL>, or in <CIT>, for example.

It is an object of the present invention to provide an improved concept for decoding an encoded multichannel signal.

This object is achieved by an apparatus for decoding an encoded multichannel audio signal of claim <NUM>, a method of decoding an encoded multichannel audio signal of claim <NUM>, or a computer program of claim <NUM>.

The present invention is based on the finding that a mixed approach is useful for decoding an encoded multi-channel signal. This mixed approach relies on using a filling signal generated by a decorrelation filter, and this filling signal is then used by a multi-channel processor such as a parametric or other multi-channel processor to generate the decoded multi-channel signal. Particularly, the decorrelation filter is a broad band filter and the multi-channel processor is configured to apply a narrow band processing to the spectral representation. Thus, the filling signal is preferably generated in the time domain by an allpass filter procedure, for example, and the multichannel processing takes place in the spectral domain using the spectral representation of the decoded base channel and, additionally, using a spectral representation of the filling signal generated from the filling signal calculated in the time domain.

Thus, the advantages of frequency domain multi-channel processing on the one hand and time domain decorrelation on the other hand are combined in a useful way to obtain a decoded multi-channel signal having a high audio quality. Nevertheless, the bitrate for transmitting the encoded multi-channel signal is kept as low as possible due to the fact that the encoded multi-channel signal is typically not a waveform-preserving encoding format but, for example, a parametric multi-channel coding format. Hence, for generating the filling signal, only decoder-available data such as the decoded base channel is used and, in certain embodiments, additional stereo parameters such as a gain parameter or a prediction parameter or, alternatively, ILD, ICC or any other stereo parameters known in the art.

Subsequently, several preferred embodiments are discussed. The most efficient way to code stereo signals is to use parametric methods such as Binaural Cue Coding or Parametric Stereo. They aim at reconstructing the spatial impression from a mono downmix by restoring several spatial cues in subbands and as such are based on psychoacoustics. There is another way of looking at parametric methods: one simply tries to parametrically model one channel by another, trying to exploit inter channel redundancy. This way, one may recover part of the secondary channel from the primary channel but one is usually left with a residual component. Omitting this component usually leads to an unstable stereo image of the decoded output. Therefore, it is necessary to fill in a suitable replacement for such residual components. Since such a replacement is blind, it is safest to take such parts from a second signal that has similar temporal and spectral properties as the downmix signal.

Hence, embodiments of the present invention is particularly useful in the context of parametric audio coder and, particularly, parametric audio decoder where replacements for missing residual parts are extracted from an artificial signal generated by a decorrelation filter on the decoder-side.

Further embodiments relate to procedures for generating the artificial signal. Embodiments relate to methods of generating an artificial second channel from which replacements for missing residual parts are extracted and its use in a fully parametric stereo coder, called enhanced Stereo Filling. The signal is more suitable for coding speech signals than the xHE-AAC signal, since its spectral shape is temporally closer to the input signal. It is generated in time domain by applying a special filter structure, and therefore independent of the filter bank in which the stereo upmix is performed. It can hence be used in different upmix procedures. It could, for instance, be used in xHE-AAC to replace the artificial signals after transforming to QMF domain, which would improve the performance for speech, as well as in the midrange of AMR-WB+ to stand in for the residual in the mid/side prediction, which would improve the performance for weakly correlated input channels and improve the stereo image. This is of special interest for codecs featuring different stereo modes (such as time domain and frequency domain stereo processing).

In preferred embodiments, the decorrelation filter comprises at least one allpass filter cell, the at least one allpass filter cell comprising two Schroeder allpass filter cells nested into a third Schroeder allpass filter, and/or the allpass filter comprises at least one allpass filter cell, the allpass filter cell comprising two cascaded Schroeder allpass filters, wherein an input into the first cascaded Schroeder allpass filter and an output from the cascaded second Schroeder allpass filter are connected, in the direction of the signal flow, before a delay stage of the third Schroeder allpass filter.

In a further embodiment, several such allpass filter cells comprising of three nested Schroeder allpass filters are cascaded in order to obtain a specifically useful allpass filter that has a good impulse response for the purpose of stereo or multi-channel decoding.

It is to be emphasized here that, although several aspects of the present invention are discussed with respect to stereo decoding generating, from a mono base channel, a left upmix channel and a right upmix channel, the present invention is also applicable for multi-channel decoding, where a signal of, for example, four channels is encoded using two base channels, wherein the first two upmix channels are generated from the first base channel and the third and the fourth upmix channel are generated from the second base channel. In other alternatives, the present invention is also useful to generate, from a single base channel, three or more upmix channels always using preferably the same filling signal. In all such procedures, however, the filling signal is generated in a broad band manner, i.e., preferably in the time domain, and the multi-channel processing for generating, from the decoded base channel, the two or more upmix channels is done in the frequency domain.

The decorrelation filter preferably operates fully in the time domain. However, other hybrid approaches are useful as well, where, for example, the decorrelation is performed by decorrelating a low band portion on the one hand and a high band portion on the other hand while, for example, the multi-channel processing is performed in a much higher spectral resolution. Thus, exemplarily, the spectral resolution of the multi-channel processing can, for example, be as high as processing each DFT or FFT line individually, and parametric data is given for several bands, where each band, for example, comprises two, three, or many more DFT/FFT/MDCT lines, and the filtering of the decoded base channel to obtain the filing signal is done broad band like i.e., in the time domain or semi-broad band like, for example, within a low band and a high band or, probably within three different bands. Thus, in any case, the spectral resolution of the stereo processing that is typically performed for individual lines or subband signals is the highest spectral resolution. Typically, the stereo parameters generated in an encoder and transmitted and used by preferred decoder have a medium spectral resolution. Thus, the parameters are given for bands, the bands can have varying bandwidths, but each band at least comprises two or more lines or subband signals generated and used by the multi-channel processors. And, the spectral resolution of the decorrelation filtering is very low and, in the case of time domain filtering extremely low or is medium, in the case of generating different decorrelated signals for different bands, but this medium spectral resolution is still lower than the resolution, in which the parameters for the parametric processing are given.

In a preferred embodiment, the filter characteristic of the decorrelation filter is an allpass filter having a constant magnitude region over the whole interesting spectral range. However, other decorrelation filters that do not have this ideal allpass filter behavior are useful as well as long as, in a preferred embodiment, a region of constant magnitude of the filter characteristic is greater than a spectral granularity of the spectral representation of the decoded base channel and the spectral granularity of the spectral representation of the filling signal.

Thus, it is made sure that the spectral granularity of the filling signal or the decoded base channel, on which the multi-channel processing is performed does not influence the decorrelation filtering, so that a high quality filling signal is generated, preferably adjusted using an energy normalization factor and then used for generating the two or more upmix channels.

Furthermore, it is to be noted that the generation of a decorrelated signal such as described with respect to subsequently discussed <FIG>, <FIG>, or <FIG> can be used in the context of a multichannel decoder, but can also be used in any other application, where a decorrelated signal is useful such as in any audio signal rendering, any reverberating operation etc..

Subsequently, preferred embodiments are discussed with respect to the accompanying drawings in which:.

<FIG> illustrates a preferred embodiment of an apparatus for decoding an encoded multichannel signal. The encoded multi-channel signal comprises an encoded base channel that is input into a base channel decoder <NUM> for decoding the encoded base channel to obtain a decoded base channel.

Furthermore, the decoded base channel is input into a decorrelation filter <NUM> for filtering at least a portion of the decoded base channel to obtain a filling signal.

Both the decoded base channel and the filling signal are input into a multi-channel processor <NUM> for performing a multi-channel processing using a spectral representation of the decoded base channel and, additionally, a spectral representation of the filling signal. The multi-channel processor outputs the decoded multi-channel signal that comprises, for example, a left upmix channel and a right upmix channel in the context of stereo processing or three or more upmix channels in the case of multi-channel processing covering more than two output channels.

The decorrelation filter <NUM> is configured as a broad band filter, and the multi-channel processor <NUM> is configured to apply a narrowband processing to the spectral representation of the decoded base channel and the spectral representation of the filling signal. Importantly, broad band filtering is also done, when the signal to be filtered is downsampled from a higher sampling rate such as downsampled to <NUM> or <NUM> from a higher sampling rate such as <NUM> or lower.

Thus, the multi-channel processor operates in a spectral granularity that is significantly higher than a spectral granularity, with which the filling signal is generated. In other words, a filter characteristic of the decorrelation filter is selected so that the region of a constant magnitude of the filter characteristic is greater than a spectral granularity of the spectral representation of the decoded base channel and a spectral granularity of the spectral representation of the filling signal.

Thus, for example, when the spectral granularity of the multi-channel processor is so that, for each spectral line of a, for example, <NUM> line DFT spectrum the upmix processing is performed, then the decorrelation filter is defined in such a way that the region of constant magnitude of the filter characteristic of the decorrelation filter has a frequency width that is higher than two or more spectral lines of the DFT spectrum. Typically, the decorrelation filter operates in the time domain, and the used spectral band, for example, from <NUM> to <NUM>. Such filters are known to be allpass filters, and it is to be noted here that a perfectly constant magnitude range where the magnitude is perfectly constant can be typically not be obtained by allpass filters, but variations from a constant magnitude by +/- <NUM>% of an average value also are found to be useful for an allpass filter and, therefore, also represent a "constant magnitude of the filter characteristic".

<FIG> illustrates an implementation of the decorrelation filter <NUM> with a time domain filter stage <NUM> and the subsequently connected spectral converted <NUM> generating a spectral representation of the filling signal. The spectral converter <NUM> is typically implemented as an FFT or a DFT processor, although other time-frequency domain conversion algorithms are useful as well.

<FIG> illustrates a preferred implementation of the cooperation between the base channel decoder <NUM> and a base channel spectral converter <NUM>. Typically, the base channel decoder is configured to operate as a time domain base channel decoder generating a time domain base channel signal while the multi-channel processor <NUM> operates in the spectral domain. Thus, the multi-channel processor <NUM> of <FIG> has, as an input stage, the base channel spectral converter <NUM> of <FIG>, and the spectral representation of the base channel spectral converter <NUM> is then forwarded to the multi-channel processor processing elements that are, for example, illustrated in <FIG>, <FIG>, <FIG>, <FIG> or <FIG>. In this context, it is to be outlined that, in general, reference numerals starting from a "<NUM>" represent elements that preferably belong to the base channel decoder <NUM> of <FIG>. Elements having a reference numeral starting with a "<NUM>" preferably belong to the decorrelation filter <NUM> of <FIG>, and elements with a reference numeral starting with "<NUM>" in the figures preferably belong to the multi-channel processor <NUM> of <FIG>. However, it is to be noted here that the separations between the individual elements are only made for describing the present invention, but any actual implementation can have different, typically hardware or alternatively software or mixed hardware/software processing blocks that are separated in a different manner than the logical separation illustrated in <FIG> and other figures.

<FIG> illustrates a preferred implementation of the filter stage <NUM> that is indicated as <NUM>'. Particularly, <FIG> illustrates a basic allpass unit that can be included in the decorrelation filter alone or together with more such cascaded allpass units as, for example, illustrated in <FIG> illustrates the decorrelation filter <NUM> with exemplarily five cascaded basic allpass units <NUM>, <NUM>, <NUM>, <NUM>, <NUM>, while each of basic allpass units can be implemented as outlined in <FIG>. Alternatively, however, the decorrelation filter can include a single basic allpass unit <NUM> of <FIG> and, therefore, represents an alternative implementation of the decorrelation filter stage <NUM>'.

Preferably, each basic allpass unit comprises two Schroeder allpass filters <NUM>, <NUM> nested into a third Schroeder allpass filter <NUM>. In this implementation, the allpass filter cell <NUM> is connected to two cascaded Schroeder allpass filters <NUM>, <NUM>, wherein input into the first cascaded Schroeder allpass filter <NUM> and an output from the cascaded second Schroeder allpass filter <NUM> are connected, in the direction of the signal flow, before a delay stage <NUM> of the third Schroeder allpass filter.

Particularly, the allpass filter illustrated in <FIG> comprises: a first adder <NUM>, a second adder <NUM>, a third adder <NUM>, a fourth adder <NUM>, a fifth adder <NUM> and a sixth adder <NUM>; a first delay stage <NUM>, a second delay stage <NUM> and a third delay stage <NUM>; a first forward feed <NUM> with a first forward gain, a first backward feed <NUM> with a first backward gain, a second forward feed <NUM> with a second forward gain and a second backward feed <NUM> with a second backward gain; and a third forward feed <NUM> with a third forward gain and a third backward feed <NUM> with a third backward gain.

The connections are illustrated in <FIG> are as follows: The input into the first adder <NUM> represents an input into the allpass filter <NUM>, wherein a second input into the first adder <NUM> is connected to an output of the third filter delay stage <NUM> and comprises the third backward feed <NUM> with a third backward gain. The output of the first adder <NUM> is connected to an input into the second adder <NUM> and is connected to an input of the sixth adder <NUM> via the third forward feed <NUM> with the third forward gain. The input into the second adder <NUM> is connected to the first delay stage <NUM> via a first backward feed <NUM> with the first backward gain. The output of the second adder <NUM> is connected to an input of the first delay stage <NUM> and is connected to an input of the third adder <NUM> via the first forward feed <NUM> with the first forward gain. The output of the first delay stage <NUM> is connected to a further input of the third adder <NUM>. The output of the third adder <NUM> is connected to an input of the fourth adder <NUM>. The further input into the fourth adder <NUM> is connected to an output of the second delay stage <NUM> via the second backward feed <NUM> with the second backward gain. The output of the fourth adder <NUM> is connected to an input into the second delay stage <NUM> and is connected to an input into the fifth adder <NUM> via the second forward feed <NUM> with the second forward gain. The output of the second delay stage <NUM> is connected to a further input into the fifth adder <NUM>. The output of the fifth adder <NUM> is connected to an input of the third delay stage <NUM>. The output of the third delay stage <NUM> is connected to an input into the sixth adder <NUM>. The further input into the sixth adder <NUM> is connected to an output of the first adder <NUM> via the third forward feed <NUM> with the third forward gain. The output of the sixth adder <NUM> represents an output of the allpass filter <NUM>.

Preferably, as illustrated in <FIG>, the multi-channel processor <NUM> is configured to determine a first upmix channel and a second upmix channel using different weighted combinations of spectral bands of the decoded base channel and corresponding spectral bands of the filling signal. Particularly, the different weighted combinations depend on a prediction factor and/or a gain factor as derived from encoded parametric information included within the encoded multi-channel signal. Furthermore, the weighted combinations preferably depend on an envelope normalization factor or, preferably an energy normalization factor calculated using a spectral band of the decoded base channel and the corresponding spectral band of the filling signal. Thus, the processor <NUM> of <FIG> receives the spectral representation of the decoded base channel and the spectral representation of the filling signal and outputs, preferably in the time domain, a first upmix channel and a second upmix channel, and the prediction factor, the gain factor, and the energy normalization factor are input in a per-band manner and these factors are then used for all spectral lines within a band, but change for a different band, where this data is retrieved from the encoded signal or locally determined in the decoder.

Particularly, the prediction factor and the gain factor typically represent encoded parameters that are decoded on the decoder side and are then used in the parametric stereo upmixing. Contrary thereto, the energy normalization factor is calculated on the decoder-side typically using a spectral band of the decoded base channel and the spectral band of the filling signal. The same is true for the envelope normalization factor. Preferably, the envelope normalization corresponds to an energy normalization per band.

Although the present invention is discussed with the specific reference encoder illustrated in <FIG> and the specific decoder illustrated in <FIG> or <FIG>, it is, however, to be noted that the generation of a broad band filling signal and the application of the broad band filling signal in multi-channel stereo decoding operating in a narrow band spectral domain can also be applied to any other parametric stereo encoding techniques known in the art. These are parametric stereo encoding known from the HE-AAC standard or from the MPEG surround standard or from Binaural Cue Coding (BCC coding) or any other stereo encoding/decoding tools or any other multi-channel encoding/decoding tools.

<FIG> illustrates a further preferred embodiment of the multi-channel decoder comprising a multi-channel processor stage <NUM> generating a first upmix channel and a second upmix channel and subsequently connected time domain bandwidth extension elements <NUM>, <NUM> that perform a time domain bandwidth extension in a guided or unguided manner to the first upmix channel and the second upmix channel individually. Typically, a windower and energy normalization factor calculator <NUM> is provided to calculate an energy normalization factor to be used by the multi-channel processor <NUM>. In alternative embodiments that are discussed with respect to <FIG> or <FIG> and <FIG> or <FIG>, however, the bandwidth extension is performed with the mono or decoded core signal and, only a single stereo processing element <NUM> of <FIG> or <FIG> is provided for generating, from the high band mono signal, a high band left channel signal and a high band right channel signal that are then added to the low band left channel signal and the low band right channel signal with the use of adders 994a and 994b.

This adding illustrated in <FIG> or <FIG> can, for example, be performed in the time domain. Then, block <NUM> generates a time domain signal. This is the preferred implementation. However, alternatively, the stereo processing <NUM> in <FIG> or <FIG> and the left channel and right channel signals from block <NUM> can be generated in the spectral domain and, the adders 994a and 994b are, for example, implemented by a synthesis filter bank so that the low band data from block <NUM> is input into the low band input of the synthesis filter bank and the high band output of block <NUM> is input into the high band input of the synthesis filter bank and the output of the synthesis filter bank is the corresponding left channel time domain signal or a right channel time domain signal.

Preferably, the windower and factor calculator <NUM> in <FIG> generates and calculates an energy value of the high band signal as, for example, also illustrated at <NUM> in <FIG> or <FIG> and uses this energy estimate for generating high band first and second upmix channels as will be discussed later on with respect to equations <NUM> to <NUM> in a preferred embodiment.

Preferably, the processor <NUM> for calculating the weighted combination receives, as an input, the energy normalization factor per band. In a preferred embodiment, however, a compression of the energy normalization factor is performed and the different weighted combinations are calculated using the compressed energy normalization factor. Thus, with respect to <FIG>, the processor <NUM> receives, instead of the non-compressed energy normalization factor, a compressed energy normalization factor. This procedure is illustrated, with respect to different embodiments, in <FIG>. Block <NUM> receives an energy of the residual or filling signal per time/frequency bin and an energy of the decoded base channel per time and frequency bin, and then calculates an absolute energy normalization factor for a band comprising several such time/frequency bins. Then, in block <NUM>, a compression of the energy normalization factor is performed, and this compression can, for example, be the usage of a logarithm function as, for example, discussed with respect to equation <NUM> later on.

Based on the compressed energy normalization factor generated by block <NUM>, different procedures for generating the compressed energy normalization factor are given. In the first alternative, a function is applied to the compressed factor as illustrated in <NUM>, and this function is preferably a non-linear function. Then, in block <NUM> the evaluated factor is expanded to obtain a specific compressed energy normalization factor. Hence, block <NUM> can, for example, be implemented to the function expression in equation (<NUM>) that will be given later on, and block <NUM> is performed by the "exponent" function within equation (<NUM>). However, a different alternative resulting in a similar compressed energy normalization factor is given in block <NUM> and <NUM>. In block <NUM> an evaluation factor is determined and, in block <NUM>, the evaluation factor is applied to the energy normalization factor obtained from block <NUM>. Thus, the application of the factor to the energy normalization factor as outlined in block <NUM> can, for example, be implemented by subsequently illustrated equation <NUM>.

Thus, as for example, illustrated in equation <NUM> later on, the evaluation factor is determined and this factor is simply a factor that can be multiplied by the energy normalization factor gnorm as determined by block <NUM> without actually performing special function evaluations. Therefore, the calculation of block <NUM> can also dispensed with, i.e., the specific calculation of the compressed energy normalization factor is not necessary, as soon as the original non-compressed energy normalization factor, and the evaluation factor and a further operand within a multiplication such as a spectral value of the filling signal are multiplied together to obtain a normalized filling signal spectral line.

<FIG> illustrates a further implementation, where the encoded multi-channel signal is not simply a mono signal but comprises an encoded mid signal and an encoded side signal, for example. In such a situation, the base channel decoder <NUM> not only decodes the encoded mid signal and the encoded side signal or, generally, the encoded first signal and the encoded second signal, but additionally performs a channel transformation <NUM>, for example, in the form of a mid/side transform and inverse mid/side transformation to calculate a primary channel such as L and a secondary channel such as R, or the transformation is a Karhunen Loeve transformation.

However, the result of the channel transformation and, particularly, the result of the decoding operation is that the primary channel is a broad band channel while the secondary channel is a narrow band channel. Then, the broad band channel is input into the decorrelation filter <NUM> and, a high pass filtering is performed in block <NUM> to generate a decorrelated high pass signal and this decorrelated high pass signal is then added to the narrow band secondary channel in the band combiner <NUM> to obtain the broad band secondary channel so that, in the end, the broad band primary channel and the broad band secondary channel are output.

<FIG> illustrates a further implementation, where a decoded base channel obtained by the base channel decoder <NUM> in a certain sampling rate associated with the encoded base channel is input into a resampler <NUM> in order to obtain a resampled base channel that is then used in the multi-channel processor that operates on the resampled channel.

<FIG> illustrates a preferred implementation of a reference stereo encoding. In block <NUM>, an inter-channel phase difference IPD is calculated for the first channel such as L and the second channel such as R. this IPD value is then, typically quantized and output for each band in each time frame as encoder output data <NUM>. Furthermore, the IPD values are used for calculating parametric data for the stereo signal such as a prediction parameter gt,b for each band b in each time frame t and a gain parameter rt,b for each band b in each time frame t.

Furthermore, both first and second channels are also used in a mid/side processor <NUM> to calculate, for each band, a mid signal and a side signal.

Depending on the implementation, only the mid signal M can be forwarded to an encoder <NUM>, and the side signal is not forwarded to the encoder <NUM> so that the output data <NUM> only comprises the encoded base channel, the parametric data generated by block <NUM> and the IPD information generated by block <NUM>.

Subsequently, a preferred embodiment is discussed with respect to a reference encoder, but it is to be noted that any other stereo encoders as discussed before can be used as well.

A DFT based stereo encoder is specified for reference. As usual, time frequency vectors Lt and Rt of the left and right channel are generated by simultaneously applying an analysis window followed by a Discrete Fourier Transform (DFT). The DFT bins are then grouped into subbands (Lt,k)k ∈ Ib resp. (Rt,k)k ∈ Ib, where Ib denotes the set of subband indices.

Calculation of IPDs and Downmixing. For the downmix, a bandwise inter-channel- phase-difference (IPD) is calculated as <MAT>.

Where z* denotes the complex conjugate of z. This is used to generate a band-wise mid and side signal <MAT> and <MAT> for k ∈ Ib, where β is an absolute phase rotation parameter e.g. given by <MAT>.

Calculation of parameters. In addition to the band-wise IPDs, two further stereo parameters are extracted. The optimal coefficient for predicting St,b by Mt,b, i.e. the number gt,b such that the energy of the remainder <MAT> is minimal, and a relative gain factor rt,b which, if applied to the mid signal Mt, equalizes the energy of pt and Mt in each band, i.e., <MAT>.

The optimal prediction coefficient can be calculated from the energies in the subbands <MAT> and the absolute value of the inner product of Lt and Rt <MAT> as <MAT>.

From this it follows that gt,b lies in [-<NUM>, <NUM>]. The residual gain can be calculated similarly from the energies and the inner product as <MAT> which implies <MAT>.

<FIG> illustrates a preferred implementation of the decoder-side. In block <NUM>, representing the base channel decoder of <FIG>, the encoded base channel M is decoded.

Then, in block 940a, the primary upmix channel such as L is calculated. Furthermore, in block 940b, the secondary upmix channel is calculated which is, for example, channel R.

Both blocks 940a and 940b are connected to the filling signal generator <NUM> and receive the parametric data generated by block <NUM> in <FIG> or <NUM> of <FIG>.

Preferably, the parametric data is given in bands having the second spectral resolution and the blocks 940a, 940b operate in high spectral resolution granularity and generate spectral lines with a first spectral resolution that is higher than the second spectral resolution.

The output of blocks 940a, 940b are, for example, input into frequency-time converters <NUM>, <NUM>. These converters can be a DFT or any other transform, and typically also comprise a subsequent synthesis window processing and a further overlap-add operation.

Additionally, the filling signal generator receives the energy normalization factor and, preferably, the compressed energy normalization factor, and this factor is used for generating a correctly leveled/weighted filling signal spectral line for blocks 940a and 940b.

Subsequently, a preferred implementation of blocks 940a, 940b is given. Both blocks comprise the calculation 941a of phase rotation factor, the calculation of a first weight for the spectral line of the decoded base channel as indicated by 942a and 942b. Furthermore, both blocks comprise the calculation 943a and 943b for the calculation of the second weight for the spectral line of the filling signal.

Furthermore, the filling signal generator <NUM> receives the energy normalization factor generated by block <NUM>. This block <NUM> receives the filling signal per band and the base channel signal per band and, then, calculates the same energy normalization factor used for all lines in a band.

Finally, this data is forwarded to the processor <NUM> for calculating the spectral lines for the first and the second upmix channels. To this end, the processor <NUM> receives the data from blocks 941a, 941b, 942a, 942b, 943a, 943b and the spectral line for the decoded base channel and the spectral line for the filling signal. The output of block <NUM> is then a corresponding spectral line for the first and the second upmix channel.

Subsequently, preferred implementations of a decoder are given.

A DFT based decoder for reference is specified which corresponds to the encoder described above. The time-frequency transform from both the encoder is applied to the decoded downmix yielding time-frequency vectors M̃t,b. Using the dequantized values <MAT>, g̃t,b, and r̃t,b, left and right channel are calculated as <MAT> and <MAT> for k ∈ Ib where p̃t,k is a substitute for the missing residual pt,k from the encoder, and gnorm is the energy normalizing factor <MAT> which turns the relative residual prediction gain rt,b into an absolute gain. A simple choice for p̃t,k would be <MAT> where db > denotes a band-wise frame-delay but this has certain drawbacks, namely.

It is therefore better to use time-frequency bins of the artificial signal which is described below.

The phase rotation factor β is again calculated as <MAT>.

For replacing missing residual parts in the stereo upmix, a second signal is generated from the time-domain input signal m̃, outputting a second signal m̃F. The design constrain for this filter is to have a short, dense impulse response. This is achieved by applying several stages of basic allpass filters obtained by nesting two Schroeder allpass filter into a third Schroeder filter, i.e. <MAT> where <MAT> and <MAT>.

These elementary allpass filters <MAT> have been proposed by Schroeder in the context of artificial reverb generation, where they are applied with both large gains and large delays. Since it is not desirable in this context to have a reverberant output signal, gains and delays are chosen to be rather small. Similarly to the reverb case, a dense and random-like impulse response is best obtained by choosing delays di that are pairwise coprime for all allpass filters.

The filter runs at a fixed sampling rate, regardless of the bandwidth or sampling rate of the signal that is delivered by the core coder. When used with the EVS coder, this is necessary since the bandwidth may be changed by a bandwidth detector during operation and the fixed sampling rate guarantees a consistent output. The preferred sampling rate for the allpass filter is <NUM>, the native super wide band sampling rate, since the absence of residual parts above <NUM> are usually not audible anymore. When used with the EVS coder, the signal is directly constructed from the core, which incorporates several resampling routines as displayed in <FIG>.

A filter that has been found to work well at <NUM> sampling rate is <MAT> where Bi are basic allpass filters with gains and delays displayed in Table <NUM>. The impulse response of this filter is depicted in <FIG>. For complexity reasons, one can also apply such a filter at lower sampling rates and/or reduce the number of basic allpass filter units.

The allpass filter unit also provides the functionality to overwrite parts of the input signal by zeros, which is encoder-controlled. This can for instance be used to delete attacks from the filter input.

To obtain a smoother output it has been found beneficial to apply a compressor to the energy- adjusting gain gnorm which compresses the values towards one. This also compensates a bit for the fact that part of the ambience is typically lost after coding the downmix at lower bitrates.

Such a compressor can be constructed by taking <MAT> where, <MAT> and the function c satisfies <MAT>.

The value of c around t then specifies how strongly this region is compressed, where the value <NUM> corresponds to no compression and the value <NUM> corresponds to total compression. Furthermore, the compression scheme is symmetric if c is even, i.e., c(t) = c(-t). One example is <MAT> which gives rise to <MAT>.

In this case, (<NUM>) can be simplified to <MAT> and one can save the special function evaluations.

When used with the EVS codec, a low delay audio codec for communication scenarios, it is desirable to perform the stereo upmix of the bandwidth extension in time domain, to safe delay induced by the time domain bandwidth extension (TBE). The stereo bandwidth upmix aims at restoring correct panning in the bandwidth extension range, but does not add a substitute for the missing residual. It is therefore desirable to add the substitute in frequency domain stereo processing, as is depicted in <FIG>.

The notation m̃ for the input signal at the decoder, m̃F for the filtered input signal, M̃t,k for the time-frequency bins of m̃ and p̃t,k for the time frequency bins of m̃F are used.

One then faces the problem that M̃t,k is not known in the bandwidth extension range, hence the energy normalizing factor <MAT> cannot be computed directly if some of the indices k∈Ib lie in the bandwidth extension range. This problem is solved as follows: let IHB and ILB denote the high band resp. low band indices of the frequency bins. Then an estimate EM̃,HB of Σk∈IHB|M̃t,k|<NUM> is obtained by calculating the energy of the windowed high band signal in time domain. Now if Ib,LB and Ib,HB denote the low band and high band indices in Ib, the indices of band b, then one has <MAT>.

Now the summands in the second sum on the right hand side are unknown, but since m̃F is obtained from m̃ by an allpass filter, one can assume that the energy of p̃t,k and m̃t,k is similarly distributed and therefore one will have <MAT>.

Therefore, the second sum on the right hand side of (<NUM>) can be estimated as <MAT>.

The artificial signal is also useful for stereo coders, which code a primary and a secondary channel. In this case, the primary channel serves as input for the allpass filter unit. The filtered output may then be used to substitute residual parts in the stereo processing, possibly after applying a shaping filter to it. In the simplest setting primary and secondary channel could be a transformation of the input channels like a mid/side or KL-transform, and the secondary channel could be limited to a smaller bandwidth. The missing part of the secondary channel could then be replaced by the filtered primary channel after applying a high pass filter.

A particularly interesting case for the artificial signal is, when the decoder features different stereo processing methods as depicted in <FIG>. The methods may be applied simultaneously (e.g. separated by bandwidth) or exclusively (e.g. frequency domain vs. time domain processing) and connected to a switching decision. Using the same artificial signal in all stereo processing methods smooths discontinuities both in the switching case and the simultaneous case.

The new method has many benefits and advantages over State of the Art Methods as for instance applied in xHE-AAC.

Time domain processing allows for a much higher time resolution as subband processing, which is applied in Parametric Stereo, which makes it possible to design a filter whose impulse response is both dense and fast decaying. This leads to the input signals spectral envelope getting less smeared out over time, or the output signal being less colored and therefore sounding more natural.

Better suitability for speech, where the optimal peak region of the filter's impulse response should lie between <NUM> and <NUM>.

The filter unit features a resampling functionality for input signals with different sampling rates. This allows for operating the filter at a fixed sampling rate, which is beneficial since it guarantees a similar output at different sampling rates; or smooths discontinuities when switching between signals of different sampling rate. For complexity reasons, the internal sampling rate should be chosen such that the filtered signal covers only the perceptually relevant frequency range.

Since the signal is generated at the input of the decoder and not connected to a filter bank, it may be used in different stereo processing units. This helps to smooth discontinuities when switching between different units, or when operating different units on different parts of the signal.

It also saves complexity, since no re-initialization is needed when switching between units.

The gain compression scheme helps to compensate for loss of ambience due to core coding.

The method relating to bandwidth extension of ACELP frames mitigates the lack of missing residual components in a panning based time domain bandwidth extension upmix, which increases stability when switching between processing the high band in DFT domain and in time domain.

The input may be replaced by zeros on a very fine time scale, which is beneficial for handling attacks.

Subsequently, additional details with respect to <FIG> or <FIG>, <FIG> or <FIG> and <FIG> are discussed.

<FIG> or <FIG> illustrates the base channel decoder <NUM> as comprising a first decoding branch having a low band decoder <NUM> and a bandwidth extension decoder <NUM> to generate a first portion of the decoded base channel. Furthermore, the base channel decoder <NUM> comprises a second decoding branch <NUM> having a full band decoder to generate a second portion of the decoded base channel.

The switching between both elements is done by a controller <NUM> illustrated as a switch controlled by a control parameter included in the encoded multi-channel signal for feeding a portion of the encoded base channel either into the first decoding branch comprising block <NUM>, <NUM> or into the second decoding branch <NUM>. The low band decoder <NUM> is implemented, for example, as an algebraic code excited linear prediction coder ACELP and the second full band decoder is implemented as a transform coded excitation (TCX) / high quality (HQ) core decoder.

The decoded downmix from blocks <NUM> or the decoded core signal from block <NUM> and, additionally, the bandwidth extension signal from block <NUM> are taken and forwarded to the procedure in <FIG> or <FIG>. Additionally, the subsequently connected decorrelation filter comprises resamplers <NUM>, <NUM>, <NUM> and, if necessary and where appropriate, delay compensation elements <NUM>, <NUM>. An adder combines the time domain bandwidth extension signal from block <NUM> and the core signal from block <NUM> and forwards same to a switch <NUM> controlled by encoded multi-channel data in the form of a switch controller in order to switch between either the first coding branch or the second coding branch depending on which signal is available.

Furthermore, a switching decision <NUM> is configured that is, for example, implemented as a transient detector. However, the transient detector does not necessarily have to be an actual detector for detecting a transient by a signal analysis, but the transient detector can also be configured to determine a side information or a specific control parameter in the encoded multi-channel signal indicating a transient in the base channel.

The switching decision <NUM> sets a switch in order to either feed the signal output from switch <NUM> into the allpass filter unit <NUM> or a zero input which results in actually deactivating the filling signal addition in the multi-channel processor for certain very specifically selectable time regions, since the EVS allpass signal generator (APSG) indicated at <NUM> in <FIG> or <FIG> operates completely in the time domain. Thus, the zero input can be selected on a sample-wise basis without having any reference to any window lengths reducing the spectral resolution as is required for spectral domain processing.

The device illustrated in <FIG> is different from the device illustrated in <FIG> in that the resamplers and delay stages are omitted in <FIG>, i.e., elements <NUM>, <NUM>, <NUM>, <NUM>, <NUM> are not required in the <FIG> device. Hence, in the <FIG> embodiment, the allpass filter units operate at <NUM> rather than at <NUM> as in <FIG>.

<FIG> or <FIG> illustrates the integration of the allpass signal generator <NUM> into the DFT stereo processing including a time domain bandwidth extension upmix. Block <NUM> outputs the bandwidth extension signal generated by block <NUM> to a high band upmixer <NUM> (TBE upmix - (Time domain) bandwidth extension upmix) for generating a high band left signal and a high band right signal from the mono band width extension signal generated by block <NUM>. Furthermore, a resampler <NUM> is provided connected before a DFT for the filling signal indicated at <NUM>. Additionally, a DFT <NUM> for the decoded base channel which is either a (fullband) decoded downmix or the (lowband) decoded core signal is provided.

Depending on the implementation, when the decoded downmix signal from the fullband decoder <NUM> is available, then block <NUM> is deactivated, and the stereo processing block <NUM> already outputs the fullband upmix signals such as a fullband left and right channel.

However, when the decoded core signal is input into DFT block <NUM>, then the block <NUM> is activated and a left channel signal and a right channel signal are added by adders 994a and 994b. However, the addition of the filling signal is nevertheless performed in the spectral domain indicated by block <NUM> in accordance with the procedures as, for example, discussed within a preferred embodiment based on the equations <NUM> to <NUM>. Thus, in such a situation, the signal output by DFT block <NUM> corresponding to the low band mid signal does not have any high band data. However, the signal output by block <NUM>, i.e., the filling signal has low band data and high band data.

In the stereo processing block, the low band data output by block <NUM> is generated by the decoded base channel and the filling signal but the high band data output by block <NUM> only consists of the filling signal and does not have any high band information from the decoded base channel, since the decoded base channel was band limited. The high band information from the decoded base channel is generated by bandwidth extension block <NUM>, is upmixed into a left high band channel and right high band channel by block <NUM> and is then added by the adders 994a, 994b.

The device illustrated in <FIG> is different from the device illustrated in <FIG> in that the resampler is omitted in <FIG>, i.e., element <NUM> is not required in the <FIG> device.

<FIG> illustrates preferred implementation of a system having multiple stereo processing units 904a to 904b, 904c as discussed before with respect to the switching between stereo modes. Each stereo processing blocks receives side information and, additionally, a certain primary signal but exactly the same filling signal irrespective of whether a certain time portion of the input signal is processed using the stereo processing algorithm 904a, a stereo processing algorithm 904b or another stereo processing algorithm 904c.

The inventive encoded audio signal can be stored on a digital storage medium or can be transmitted on a transmission medium such as a wireless transmission medium or a wired transmission medium such as the Internet.

The implementation can be performed using a non-transitory storage medium or a digital storage medium, for example a floppy disk, a DVD, a Blu-Ray, a CD, a ROM, a PROM, an EPROM, an EEPROM or a FLASH memory, having electronically readable control signals stored thereon, which cooperate (or are capable of cooperating) with a programmable computer system such that the respective method is performed.

The above described embodiments are merely illustrative for the principles of the present invention. It is understood that modifications and variations of the arrangements and the details described herein will be apparent to others skilled in the art. It is the intent, therefore, to be limited only by the scope of the impending patent claims and not by the specific details presented by way of description and explanation of the embodiments herein.

It is further to be noted that methods disclosed in the specification or in the claims may be implemented by a device having means for performing each of the respective steps of these methods.

Claim 1:
Apparatus for decoding an encoded multichannel audio signal, comprising:
a base channel decoder (<NUM>) for decoding an encoded base audio channel to obtain a decoded base channel;
a decorrelation filter (<NUM>) for filtering at least a portion of the decoded base channel to obtain a filling signal; and
a multichannel processor (<NUM>) for performing a multichannel processing using a spectral representation of the decoded base channel and a spectral representation of the filling signal,
wherein the decorrelation filter (<NUM>) is a broad band filter and the multichannel processor (<NUM>) is configured to apply a narrow band processing to the spectral representation of the decoded base channel and the spectral representation of the filling signal, and
wherein the multichannel processor (<NUM>) is configured to determine (<NUM>) a first upmix channel and a second upmix channel using different weighted combinations of spectral bands of the decoded base channel and corresponding spectral bands of the filling signal, the different weighted combinations depending on a prediction factor and/or a gain factor and/or an envelope or energy normalization factor calculated using a spectral band of the decoded base channel and a corresponding spectral band of the filling signal.