Patent Description:
As wireless communication advances, generational standards advance based on the requirements of new use cases. In a fifth generation (<NUM>) of wireless communication, there may be new use cases for New Radio (NR) such as ultra-low latency and massive machine type communication. In LTE turbo codes were used for channel coding, but may not provide the performance that new use cases demand. New systems and methods designed for NR may be required to accommodate new use cases.

The invention is defined by a method for channel coding and a corresponding apparatus according to independent claims <NUM> and <NUM>.

According to the invention as claimed, coded bits from a circular buffer are divided into a first and second part and mapped according to different modulation orders whereby the lengths of the parts depend on the modulation orders and are further derived from a length, in symbols and based on a first modulation scheme having the first modulation order, that results from puncturing the coded bits using a rate-matching function. Non-claimed examples described herein may be useful for understanding the invention.

The full duplex radio may include an interference management unit <NUM> to reduce and or substantially eliminate self-interference via either hardware (e.g., a choke) or signal processing via a processor (e.g., a separate processor (not shown) or via processor <NUM>).

In view of <FIG>, and the corresponding description of <FIG>, one or more, or all, of the functions described herein with regard to one or more of: WTRU 102a-d, Base Station 114a-b, eNode-B 160a-c, MME <NUM>, SGW <NUM>, PGW <NUM>, gNB 180a-c, AMF 182a-ab, UPF 184a-b, SMF 183a-b, DN 185a-b, and/or any other device(s) described herein, may be performed by one or more emulation devices (not shown).

Communication in a RAN may be carried out according to a number of protocols. These protocols may be categorized into a number of layers, such as packet data convergence protocol (PDCP), Radio-link Control (RLC), Medium-access control (MAC), and a Physical layer (PHY). These layers may exist and have reciprocal functions in both a transmitting entity and a receiving entity. A transmitting entity may be a base station, such as a gNB, and a receiving entity, such as a WTRU, or these roles may be reversed. In the PHY layer channel coding/decoding, modulation/demodulation, multi-antenna mapping, and other related functions may be handled.

For channel coding, various approaches may be used such as Turbo Code, Low-Density Parity Code (LDPC), and/or Polar Codes. For NR use scenarios Polar Codes may be used because they present favorable results in encoding and decoding in addition to code construction when compared with other approaches.

There may be several methods of constructing Polar Codes to optimize communication performance. Code construction for Polar Codes may require calculating reliabilities of input bits and selecting frozen and unfrozen bits by sorting their reliabilities. A given code construction procedure may assume some specific conditions of channel characteristics and the final code construction may be changed if the channel characteristics, including the specific conditions, change.

In one scenario, rate-matching may be based on a puncturing scheme such as Quasi-Uniform Puncturing (QUP) where output bits may be punctured by a pattern of bit reversal that starts from the beginning. QUP shows good performance but code reconstruction is needed for every unique puncturing number. This means that the positions of frozen bits may be different for each additional puncturing bit. Code reconstruction of polar codes may require a large computational load if frequent rate matching needs to be applied for a coded block.

In one embodiment, there may be a puncturing scheme where output bits are punctured by a pattern of bit reversal, similar to QUP, but different in that the puncturing starts at the end of the coded bits instead. The decoder may assume that the corresponding input bits (gradually decreased index from the end) are set to <NUM> for calculating a log likelihood (LL). In this puncturing scheme no code reconstruction may be required, but the positions of frozen bits may be changed for each puncturing case.

Polar Code specific HARQ schemes may be employed to take advantage of the benefits of a Polar Code approach. A rate-matching scheme may be combined with any type of channel coding and modulation scheme, but performance improvements may be seen when using a Polar Code specific scheme, such as those discussed herein.

<FIG> illustrates a non-exhaustive configuration of components used to implement one or more rate-matching embodiments for a transmitter. <FIG> illustrates an example procedure for rate-matching according to one or more embodiments for a transmitter. <FIG> and <FIG> will be discussed in conjunction with one another. Initially at <NUM>, K information bits are encoded <NUM> by the channel encoder <NUM> resulting in N output bits, also known as a coded stream (e.g., codeword). At <NUM>, the N output bits are processed at an interleaver <NUM>. After interleaving <NUM>, there may be a circular buffer <NUM> before the block division that may exist for HARQ operation where some coded bits are selected <NUM> from the circular buffer <NUM> for each transmission. The selected coded stream from the circular buffer <NUM> is then divided <NUM> into at least two parts, each part also known as a coded block, by a block divider <NUM>. The constant modulation for each block may result in a simple implementation for managing operations for a filter and/or power amp in the transmitter. The structure of the interleaver and the block divider may minimize required buffers to decode the received symbols at a receiver side during a HARQ processes.

After the division, each part (i.e., coded block) is mapped/modulated <NUM> by modulator 205a, 205b,. 205n (collectively known as modulator <NUM>). In one example, a first part may be mapped by M<NUM> =<NUM>m<NUM> modulation, a second part may be mapped by M<NUM> =<NUM>m<NUM> modulation, and this may be repeated for n blocks using Mn =<NUM>mn, where mn is the nth codeword and Mn is the n-ary (e.g., primary, secondary, etc.) number modulation. The modulated bits, also known as symbols, may be transmitted <NUM> by a transmitter <NUM>.

<FIG> illustrates an example where the N bit coded stream is divided into two parts of a first part <NUM> with M<NUM> modulation and a second part <NUM> with M<NUM> modulation. However, any of the embodiments discussed herein may not be restricted to two parts and may have any number of parts as explained with reference to <FIG> and <FIG>.

In one scenario, a rate-matching scheme may assume a single modulation scheme (M<NUM>-ary modulation) during one transmission time interval (TTI) where <MAT> is representative of the number of modulation symbols for the whole coded bit stream, (i. e, codeword). If P bits are punctured from the output bits, the number of modulation symbols is reduced to <MAT> where [x] is the minimum integer larger than x.

In an embodiment related to the example shown in <FIG>, a rate-matching scheme may have <MAT> consecutive bits mapped as M<NUM>-ary symbols and <MAT> consecutive bits are mapped as M<NUM>-ary symbols. According to the invention, no explicit puncturing of the bit stream is employed, but an implicit puncturing scheme is performed where the modulation order of each divided codeword part is selected in accordance with the number of bits in these parts. By applying two different modulation orders (i.e., m<NUM> and m<NUM>), it can achieve the equivalent rate-matching function as puncturing P bits for single modulation order (i.e., m<NUM>).

In one example it may be assumed that N=<NUM> and P=<NUM> for a coding scheme. The scheme may initially select the modulation order of the parts, such as m<NUM>=<NUM> and m<NUM>=<NUM>. Then, using the segmentation described herein, the number of bits in each part is determined, that is N<NUM> = <MAT> and N<NUM> = N - N<NUM>=<NUM>. That is, the first <NUM> bits are modulated as Binary Phase Shift Keying (BPSK) symbols and the remaining <NUM> bits are modulated as <NUM> Quadrature Phase Shift Keying (QPSK) symbols. The total number of modulation symbols is <NUM> symbols and it has the same number of symbols when <NUM> bits are punctured and BPSK is adopted as the modulation scheme for all punctured bits. In another example, different modulation techniques may be used such as any type of phase-shift keying (PSK), frequency-shift keying (FSK), amplitude-shift keying (ASK), and/or quadrature amplitude modulation (QAM).

For values of m<NUM> and m<NUM>, the condition of m<NUM> - m<NUM>=<NUM> might be preferred to improve granularity but another configuration where m<NUM> - m<NUM>><NUM> may also be applied without a loss of generality.

As described herein, a reconstruction process of encoding at the codeword parts may not be required, hence the position of frozen bits may be kept the same. This outcome may provide a significant reduction in encoding computation in case of rate matching for polar codes.

<FIG> illustrates a non-exhaustive configuration of components used to implement one or more rate-matching embodiments for a receiver. <FIG> illustrates an example procedure for rate-matching for a receiver according to one or more embodiments. <FIG> and <FIG> will be discussed in conjunction with one another. Initially at <NUM>, symbols may be received at a receiver <NUM> from a channel. The received symbols may then be divided <NUM> into two parts at a block divider <NUM>. A log likelihood ratio (LLR) values or LL values may be computed <NUM> by a demodulator (or demapper) for each part 303a, 303b,. 303n, where n is the number of parts. Depending on the modulation order and the constellation adopted for each block, the LLR or LL calculation method may be different. In one example, depending on the selection of the modulation orders for each codeword part, for example, m<NUM> and m<NUM> in case of two parts, the transmitter may have sent the modulation indices to the receiver along with the other decoding related information to be used at the receiver. After the LLR or LL calculation <NUM>, the parts are combined <NUM> with other bits in the circular buffer <NUM> and then de-interleaved <NUM> by a de-interleaver <NUM> based on the interleaving pattern of the transmitter side. After de-interleaving <NUM>, the de-interleaved parts are input to a channel decoder <NUM> to be decoded <NUM> and the original information bits are output from the channel decoder <NUM> for processing <NUM>.

<FIG> shows a Block Error Rate (BLER) test of the rate-matching techniques and embodiments disclosed herein compared to an alternative puncturing scheme with polar code encoding. The horizontal axis shows a normalized signal to noise ration Es/N<NUM> <NUM>. The vertical axis shows the BLER <NUM> ratio that is defined by the number of erroneous blocks received to the total number of blocks sent (an erroneous block being a transport block that has different decoded bit(s) from the information bits sent by the transmitter). The rate-matching techniques and embodiments disclosed herein are indicated by the designation irregular modulation scheme. These results comprise examples of a <NUM>/<NUM> coding rate with an irregular modulation scheme at <NUM>, a <NUM>/<NUM> coding rate with the irregular modulation scheme at <NUM>, and a <NUM>/<NUM> coding rate with an irregular modulation scheme at <NUM>. The results from the techniques and embodiments disclosed herein were compared to alternative methodologies of polar code with N=<NUM> and K=<NUM> and rate-matched to example situations of a <NUM>/<NUM> coding rate <NUM>, a <NUM>/<NUM> coding rate with a puncturing scheme at <NUM>, a <NUM>/<NUM> coding rate with a puncturing scheme at <NUM>, and a <NUM>/<NUM> coding rate with a puncturing scheme at <NUM>. As can be seen, the disclosed techniques and embodiments have coding gains of <NUM> dB, <NUM> dB and <NUM> dB over the alternative puncturing scheme at a BLER of <NUM>-<NUM> while having the same spectral efficiency.

In one embodiment, a HARQ procedure may involve a retransmission technique that is based on the rate matching block as discussed herein. The bit position may be defined as bit index <NUM> ~ N-<NUM> for bits after channel coding and interleaving. Some of the parameters used for describing this HARQ scheme are: bi may represent a starting bit index of an i-th retransmission; Li may represent a length of an i-th retransmission; and ci may represent an offset value of an i-th retransmission. The starting bit index of the i-th retransmission (i.e., the (i + <NUM>)-th transmission), bi, may be defined as in Equation <NUM>. For i><NUM>, <MAT>.

Where a modular operator amodb is the remainder of a/b. Where i=<NUM> corresponds to the first transmission and i><NUM> corresponds to the retransmission when there are errors in decoding the first transmission. Thus, i=<NUM> corresponds to the first retransmission, also known as the second transmission, after the first transmission (i=<NUM>). Accordingly, where i=<NUM>, then bi=<NUM>.

Additionally, b<NUM> may be set to N<NUM> + c<NUM>, without a loss of generality, and the second transmission in HARQ may start from the same starting bit index of M<NUM>-ary modulation.

The values of Li (i><NUM>) may be determined by a base station, or other such node, depending on an established link adaptation scheme and this information may be sent to the receiver via corresponding control channel information. The base station may select a modulation and coding scheme (MCS) level for downlink and/or uplink transmission according to measurement information or adaptation strategy, and Li may be calculated based on the base station's scheduling algorithm. The modulation order for the transmitted bits may be constant. For some cases, Li may be equal for all i><NUM> and Li = L. Further, ci may be set to zero for all values of i. Some performance gain may be expected by setting ci to other values rather than zero.

<FIG> illustrates an example where the N bit coded stream is divided into two parts of a first part <NUM> with M<NUM> modulation and a second part <NUM> with M<NUM> modulation. However, any of the embodiments discussed herein may not be restricted to two parts and may have any number of parts. Along the horizontal axis may be considered the bit index, which starts at <NUM> and goes to N-<NUM>. The example of <FIG> shows an instance where ci = <NUM> for all i. It then follows that in this example, for i=<NUM> then b<NUM> = <NUM> at <NUM>; for i=<NUM> then b<NUM> = (N<NUM> + L<NUM>)modN at <NUM>; for i=<NUM> then b<NUM> = N<NUM> at <NUM>; and for i = <NUM> then b<NUM> = (N<NUM> + L<NUM> + L<NUM>)modN at <NUM>.

Before block division in the transmitter as shown in <FIG>, the interleaved bits may be stored in a circular buffer and the bits for each transmission of HARQ operation may be selected in a bit selection block. In the first transmission, all N(= L<NUM>) bits may be selected and Li bits may be selected for i-th retransmission.

In the receiver, the received symbols in the retransmissions may not be divided in the block divider in <FIG> differently from the first transmission. The proper positions for LLR or LL values in a circular buffer may be selected in a bit selection block and they may be stored or combined with the conventional values in the positions of the circular buffer for HARQ operation.

For a modulation adopted for the first transmission and retransmissions thereafter, the reliability of each bit position in a mapping may vary, which may be compensated for by combining a retransmitted block with a different order of mapping input bits. The offset ci may change the order of mapping input bits.

<FIG> shows an example of an LTE QAM constellation and effective constellations when offset according to the embodiments disclosed herein. When the QAM constellation shown in <NUM> is applied in the first transmission, some coding gain may be achieved in the first retransmission (the second transmission) by setting c<NUM> to <NUM>. Four bits indicate a QAM symbol and the reliability of each bit position is different in <NUM>. Where "abcd" indicates a QAM symbol, and "ab" has larger reliability than "cd", then there may be some coding gain by combining less reliable bit(s) in the first transmission with more reliable bit(s) in the first retransmission (the second transmission). In <NUM>, the effective constellation in the second transmission is shown when c<NUM>=<NUM>. By shifting two symbols, "ab" and "cd" in the first transmission then they may be combined with "cd" and "ab" in the second transmission respectively.

For the retransmissions of i><NUM>, ci may be set to maximize a coding gain. In one example QAM, when ci = <NUM> then i=odd, and when ci = <NUM> then i=even.

Instead of offsets, a different modulation mapping may be used to improve gains of each retransmission. For example, a bit reversed mapping as shown in <NUM> may be used in the first retransmission (the second transmission). In such an example, the bit reversed mapping may be used when i=odd and the same mapping as the first transmission may be used when i=even.

For a HARQ scheme according to one of the embodiments disclosed herein, the effective code rate Ri of for each transmission may be derived as shown in Equation <NUM>.

The effective spectral efficiency Si of the disclosed HARQ scheme for each transmission may be derived as shown in Equation <NUM>.

In Equation <NUM>, <MAT> is the modulation order adopted for i-th retransmission. In one option, the modulation order of <MAT>, may be selected in conjunction with the puncturing number, P. Hence, the modulation order of each retransmission may be selected such that the overall puncturing number, P, is satisfied after all retransmissions are completed. In this case, the transmitter may identify <MAT>, where i = {<NUM>,. }, such that an overall puncturing ratio, P, is satisfied for the encoded codeword. The modulation series corresponding to the retransmission parts may be sent to the receiver via a control channel along with the other retransmission related information.

<FIG> and <FIG> are an example of a HARQ scheme according to one or more techniques or embodiments as disclosed herein. Similar numbering and meaning may be taken from the elements and related description of <FIG>. There may be a first transmission with a first part <NUM> and a second part <NUM>.

In the retransmissions, a single constant modulation may be used as Equation <NUM> and multiple modulations may also be applied. The retransmitted bit block with length Lj may be divided into multiple blocks, each with a different modulation. The division border in the retransmitted bit block may follow the same border as in the first transmission. For example, M<NUM>-ary modulation may be used for the second part and M<NUM>-ary modulation may be used for the first part in the retransmission <NUM>, <NUM>. By using different modulations from the first transmission <NUM>, the decoder may have more combining gain.

Initially at <NUM>, K information bits are input to the channel encoder and N output bits, also known as coded stream (e.g., codeword), are generated at <NUM>. Next, the generated N output bits are interleaved <NUM> and stored in a circular buffer. The coded stream is then divided <NUM> into at least two parts N<NUM> and N<NUM>, each part also known as a coded block, by a block divider. After the division, each part (i.e., coded block) is mapped/modulated <NUM> by modulators of different orders. In one example, a first part may be mapped by M<NUM> =<NUM>m<NUM> modulation, a second block may be mapped by M<NUM> =<NUM>m<NUM> modulation, and this may be repeated for n blocks using Mn =<NUM>mn, where mn is the nth codeword and Mn is the n-ary (e.g., primary, secondary, etc.) number modulation. The modulation generates symbols that are then transmitted <NUM> in a first transmission <NUM> starting at bit b<NUM> <NUM> with length L<NUM>. After the first transmission <NUM>, the retransmissions may start when the receiver finds error(s) in the decoded bits and send a negative acknowledgement to the transmitter for each transmission. The retransmissions may not start if the receiver finds no error in the decoded bits and may send a positive acknowledgement to the transmitter. In a situation where retransmission is needed, at <NUM> a HARQ based retransmission (second transmission <NUM>) may start at bit b<NUM> <NUM> with length L<NUM>. Note that because the second transmission <NUM> starts at bit b<NUM> <NUM>, which is the beginning of the second part <NUM>, the buffer becomes circular in nature as shown with arrow <NUM> and continues into the first part <NUM>. At <NUM> a HARQ based retransmission (third transmission <NUM>) may start at bit b<NUM> <NUM> with length L<NUM>. At <NUM> a HARQ based retransmission (fourth transmission <NUM>) may start at bit b<NUM> <NUM> with length L<NUM>. Note that because the fourth transmission <NUM> starts at bit b<NUM> <NUM>, which is past the beginning of the second part <NUM>, the buffer becomes circular in nature as shown with arrow <NUM> and continues into the first part <NUM>. This example may be similar to a "circular buffer" in one or more LTE specifications. Each transmission may be a different revision number as discussed herein.

In one instance the starting bit index in a HARQ first retransmission may start in the starting bit index of the second part. Alternatively or in combination, the starting bit index in all retransmissions may have offsets to improve performance. The number of bits in each part and modulation order may be determined based on the equation described with reference to the disclosed rate matching method, which is equivalent to puncturing P bits. The channel coded bits that are interleaved and divided into multiple parts may use different modulations for each part.

In one example, the relation between rv and bi may be defined as in Table <NUM> below. In the table, rv may be defined as an integer with a length of two bits that may correspond to the bit position previously defined as bi. The rv may be assigned by base stations, or other such nodes, and changed for each resource allocation. The rv may be sent to a WTRU by a control channel, such as PDCCH, or may be defined by a predetermined rule like <NUM>→<NUM>→<NUM>→<NUM>.

<FIG> shows an example of a BLER performance according to embodiments for a HARQ scheme when polar codes are assumed as the channel coding scheme as disclosed herein. The horizontal axis shows a normalized signal to noise ration Es/N<NUM> <NUM>. The vertical axis shows the BLER <NUM> ratio that is defined by the number of erroneous blocks received to the total number of blocks sent (an erroneous block being a transport block that has different decoded bit(s) from the information bits sent by the transmitter. The disclosed HARQ scheme provides coding gains of <NUM> dB, <NUM> dB and <NUM> dB at a BLER of <NUM>-<NUM> over the 1st <NUM> transmission from <NUM>nd <NUM>, <NUM>rd <NUM>, and <NUM>th <NUM> transmissions (1st, 2nd and 3rd retransmissions). In each retransmission, the incremental redundancy corresponding to the first transmission is retransmitted for HARQ operation.

Claim 1:
A method for channel coding, the method comprising:
selecting (<NUM>) a plurality of coded bits from a circular buffer;
dividing (<NUM>) the plurality of coded bits from the circular buffer into a first part (<NUM>) and a second part (<NUM>);
mapping (<NUM>) the first part according to a first modulation scheme (M1) and the second part according to a second modulation scheme (M2),
wherein the second modulation scheme includes a second modulation order greater than a first modulation order of the first modulation scheme, wherein the length of the first part (N1) and the length of the second part (N2) depend on the first and second modulation orders, and
wherein the length of the first part and the length of the second part are further derived from a length in symbols based on the first modulation scheme that results from puncturing the coded bits using a rate-matching function; and
transmitting (<NUM>) a first transmission based on the mapping.