Patent Description:
A radio frequency transmitter is a common communication element, and may be configured to transmit a radio frequency (RF) output signal. The RF output signal is mostly generated by a radio frequency front-end of the radio frequency transmitter.

Currently, more radio frequency front-ends are integrated with a current source set. The radio frequency transmitter may control the current source set, generate a set output signal, and obtain an RF output signal based on the set output signal.

However, because a circuit structure of the current source set is complex, in a case in which the radio frequency transmitter outputs an output signal of a millimeter wave band, a parasitic effect in a current source array is obvious. Consequently, efficiency of the current source array is low, and improvement of performance of the radio frequency transmitter is limited. Therefore, efficiency of the radio frequency transmitter currently needs to be further improved.

<CIT> discloses transmitter comprising: a modulation circuit configured to modulate a baseband signal into a multi-bit digital signal including a component in a radio frequency band; a switch-mode amplifier disposed corresponding to each bit of the multi-bit digital signal output from the modulation circuit and configured to amplify the multi-bit digital signal on a bit-by-bit basis; a signal-synthesizing circuit configured to signal-synthesize the multi-bit digital signal output from the respective switch-mode amplifiers as a transmission signal; and an antenna configured to transmit the transmission signal, wherein the signal-synthesizing circuit includes: a frequency-variable band-limiting unit that band-limits an output signal from each of the switch-mode amplifiers; a voltage/current source conversion unit that converts an output signal from each of the band-limiting unit from voltage to current, the voltage/current source conversion unit including at least a variable capacitance; and a synthesizing point configured to connect output nodes of the respective voltage/current source conversion unit and synthesize output signals output from the respective voltage/current source conversion unit, and the transmitter further comprises an impedance correction unit that corrects an impedance, the impedance correction unit being disposed on a signal path between the synthesizing point of the signal-synthesizing circuit and the antenna serving as a load.

In view of this, an embodiment of this application provides a current source array and a radio frequency transmitter as defined in the claims. A compensation circuit is disposed in the radio frequency transmitter, to improve efficiency of the radio frequency transmitter.

Because a structure of the current source set is complex, in a case in which the radio frequency transmitter works in a millimeter wave band, an obvious parasitic effect will appear in the current source set. The parasitic effect may also be understood as that a parasitic capacitor and a parasitic inductor exist between the N current source subsets of the current source set, and therefore load impedance of different current source subsets is different. In other words, an impedance mismatch occurs in the current source set. Therefore, a power loss of the current source set is increased, and efficiency of the radio frequency transmitter is reduced. In this embodiment of this application, the compensation circuit is disposed in the radio frequency front-end, and the compensation circuit compensates for a difference of load impedance between the N current source subsets, so that the impedance mismatch in the current source set may be improved. This helps improve the efficiency of the radio frequency transmitter.

The specific operation method in the method embodiment may also be applied to an apparatus embodiment or a system embodiment. It should be noted that in the descriptions of this application, "at least one" means one or more, and "a plurality of" means two or more. In view of this, "a plurality of" may also be understood as "at least two" in embodiments of this application. The term "and/or" describes an association relationship for describing associated objects and represents that three relationships may exist. In addition, the character "/" generally represents an "or" relationship between the associated objects. In addition, it should be understood that, in the description of this application, terms "first" and "second" are only used to distinguish the purpose of the description, but cannot be understood as indication or implication of relative importance, and cannot be understood as an indication or implication of a sequence.

The following clearly describes the technical solutions in embodiments of this application with reference to the accompanying drawings in embodiments of this application.

As a modern wireless communication rate is increasing, a radio frequency transmitter with a high output power and high efficiency has become an urgent need. For example, in fields, such as millimeter wave band communication in the 5th generation (<NUM>) communication and the terahertz (THz) imaging, performance requirements of a radio frequency transmitter are urgently increasing, such as a high output power, high efficiency, and a high integration level.

Currently, more radio frequency transmitters are integrated with a current source set, to improve an integration level and transmit efficiency of the radio frequency transmitters. <FIG> is a schematic diagram of a structure of a radio frequency transmitter according to an embodiment of this application. The radio frequency transmitter <NUM> may be a digital transmitter. The radio frequency transmitter <NUM> includes a control circuit <NUM> and a radio frequency front-end <NUM>. The radio frequency front-end <NUM> includes a current source set <NUM> and a matching network <NUM> and a compensation circuit <NUM>.

The current source set <NUM> includes N current source subsets (<NUM>, <NUM>,. , and 1021N), where N is an integer greater than <NUM>, and the current source subset includes at least one current source unit. As shown in <FIG>, output ends of the N current source subsets are connected in parallel by using an output signal cable <NUM>. A first end p1 of the output signal cable <NUM> is connected to a matching network <NUM>, and the output signal cable <NUM> may output a set output signal of the current source set <NUM> to the matching network <NUM> by using the first end p1. A set output signal of the current source set <NUM> includes N sub-signals respectively output by the N current source subsets. It may also be understood that the N sub-signals respectively output by the N current source subsets are superimposed with each other to form a set output signal O of the current source set <NUM>.

The control circuit <NUM> may output a plurality of control signals, where the plurality of control signals are in a one-to-one correspondence with a plurality of current source units in the current source set <NUM>. For example, if the current source subset includes M current source units, the current source set <NUM> includes M * N current source units. The control circuit <NUM> may output M * N control signals, where an <m, n>th control signal I<m, n> corresponds to an mth current source unit A<m, n> in an nth current source subset of the current source set <NUM>, where m ranges from <NUM> to M, and n ranges from <NUM> to N.

In a possible implementation, the control signals may include a digital sub-signal and a drive signal. In other words, the control signal I<m, n> includes a digital sub-signal D<m, n> and a drive signal S<m, n>. Specifically, the control circuit <NUM> may output M * N digital sub-signals and M * N drive signals. The <m, n>th digital sub-signal D<m, n> corresponds to the mth current source unit A<m, n> in the nth current source subset of the current source set <NUM>, where m ranges from <NUM> to M, and n ranges from <NUM> to N. An <m, n>th drive signal S<m, n> also corresponds to the current source unit A<m, n>. The current source unit A<m, n> may output a unit output signal O<m, n> under control of the digital sub-signal D<m, n> and the drive signal S<m, n>.

It should be noted that quantities of current source units in different current source subsets may also be different. In this case, this embodiment of this application is still applicable.

Generally, the radio frequency front-end <NUM> may include one or more current source sets. Based on a quantity of the current source sets, the radio frequency front-end <NUM> may include at least a single-ended type and a differential type. Specifically, the single-ended radio frequency front-end <NUM> includes only one current source set. The differential radio frequency front-end <NUM> may include two current source sets, and set output signals of the two current source sets are reverse signals of each other. Then, two types of the radio frequency front-ends <NUM> are further described by using the following examples.

The radio frequency front-end <NUM> in <FIG> belongs to the single-ended type, and includes only one current source set <NUM>. Generally, the current source units in the current source set <NUM> have a same circuit structure. For example, a circuit structure of the current source unit <m, n> may be shown in <FIG>. The current source unit includes a first drive tube M1 and a second drive tube M2, and the first drive tube M1 and the second drive tube M2 form a cascade circuit.

Specifically, a source of the first drive tube M1 is connected to a drain of the second drive tube M2, and a source of the second drive tube M2 is grounded. A gate of the first drive tube M1 is configured to receive the digital sub-signal D<m, n> corresponding to the current source unit A<m, n>, and a drain of the first drive tube M1 is configured to output the unit output signal O<m, n> of the current source unit <m, n>. A gate of the second drive tube M2 is configured to receive the drive signal S<m, n> corresponding to the current source unit A<m, n>.

In the current source set <NUM>, unit output signals of the current source units are superimposed on each other in the output signal cable <NUM>, to form the set output signal O of the current source set <NUM>. In a possible implementation, as shown in <FIG>, the radio frequency front-end <NUM> further includes a matching network <NUM>. The matching network <NUM> may perform impedance matching on the set output signal O of the current source set <NUM>.

For example, in the single-ended radio frequency front-end <NUM>, a circuit structure of the matching network <NUM> may be shown in <FIG>. Specifically, the matching network <NUM> includes a capacitor Cp, an inductor L<NUM>, an inductor L<NUM>, and a capacitor Co. One end of the capacitor Cp is connected to one end of the inductor L<NUM>, and the end of the capacitor Cp connected to the inductor L<NUM> may receive the set output signal O of the current source set <NUM>. The other end of the inductor L<NUM> is configured to receive a power supply voltage Vi, and the other end of the capacitor Cp is grounded. The inductor L<NUM> is magnetically coupled to the inductor L<NUM>. One end of the inductor L<NUM> is connected to one end of the capacitor Co, and the other end of the inductor L<NUM> and the other end of the capacitor Co are grounded. The end of the capacitor Co connected to the inductor L<NUM> may output an RF output signal RFout. The end of the inductor L<NUM> connected to the capacitor Cp and the end of the inductor L<NUM> connected to the capacitor Co are mutually homologous ends.

In the matching network <NUM> shown in <FIG>, capacitance values of the capacitor Cp and the capacitor Co, and inductance values of the inductor L<NUM> and the inductor L<NUM> are all configured based on load impedance RL of the radio frequency front-end <NUM> and optimal load impedance Zopt of the current source set <NUM>. The optimal load impedance Zopt is load impedance obtained through calculation in which efficiency, an output power, and the like of the current source set <NUM> are optimal. For the optimal load impedance Zopt, refer to the current technology. In other words, the matching network <NUM> may match the load impedance RL of the radio frequency front-end <NUM> as the optimal load impedance Zopt of the current source set <NUM>. This helps reduce a power loss generated when the RF output signal RFout passes through a load circuit.

For example, <FIG> shows an application of a differential radio frequency front-end <NUM> in a radio frequency transmitter. As shown in <FIG>, the radio frequency front-end <NUM> includes a current source set <NUM> and a current source set <NUM>. The current source set <NUM> includes N current source subsets (<NUM>, <NUM>,. , and 1025N). Generally, for specific implementations of the current source set <NUM> and the current source set <NUM> in the differential-type radio frequency front-end <NUM>, refer to the implementation of the current source set <NUM> in the single-ended type.

It should be noted that, generally, the current source set <NUM> and the current source set <NUM> have a same quantity of current source units. A difference lies in that the control circuit <NUM> provides a plurality of control signals for current source units in the current source set <NUM> and current source units in the current source set <NUM> respectively, so that a set output signal O+ of the current source set <NUM> and a set output signal O- of the current source set <NUM> are reverse signals of each other. The current source set <NUM> may also be referred to as a positive-phase current source set, a current source unit in the current source set <NUM> may also be referred to as a positive-phase current source unit, and an output signal of the positive-phase current source unit may also be referred to as a positive-phase unit output signal. The current source set <NUM> may also be referred to as a negative-phase current source set, a current source unit in the current source set <NUM> may also be referred to as a negative-phase current source unit, and an output signal of the negative-phase current source unit may also be referred to as a negative-phase unit output signal.

Specifically, the control circuit <NUM> may output a plurality of digital sub-signals, positive-phase drive signals, and negative-phase drive signals. The plurality of digital sub-signals output by the control circuit <NUM> are in a one-to-one correspondence with a plurality of positive-phase current source units in the current source set <NUM>, and the plurality of digital sub-signals are further in a one-to-one correspondence with a plurality of negative-phase current source units in the current source set <NUM>. It may also be understood that any digital sub-signal output by the control circuit <NUM> corresponds to one positive-phase current source unit and one negative-phase current source unit. The digital sub-signal may be used to control a corresponding positive-phase current source unit or a corresponding negative-phase current source unit. The plurality of positive-phase drive signals output by the control circuit <NUM> are in a one-to-one correspondence with the plurality of positive-phase current source units in the current source set <NUM>, and the plurality of negative-phase drive signals output by the control circuit <NUM> are in a one-to-one correspondence with the plurality of negative-phase current sources in the current source set <NUM>.

For example, as shown in <FIG>, a positive-phase current source unit A+<m, n> corresponds to a digital sub-signal D<m, n>, and corresponds to a positive-phase drive signal an S+<m, n>. In other words, a control signal I+<m, n> corresponding to the positive-phase current source unit A+<m, n> includes the digital sub-signal D<m, n> and the positive-phase drive signal S+<m, n>. The positive-phase current source unit A+<m, n> may output a unit output signal O+<m, n> under control of the digital sub-signal D<m, n> and the positive-phase drive signal S+<m, n>. Unit output signals of the current sources in the current source set <NUM> form a set output signal O+ of the current source set <NUM>.

As shown in <FIG>, a negative-phase current source unit A-<m, n> corresponds to a digital sub-signal D<m, n>, and corresponds to a negative-phase drive signal S-<m, n>. In other words, a control signal I-<m, n> corresponding to the negative-phase current source unit A-<m, n> includes the digital sub-signal D<m, n> and the negative-phase drive signal S-<m, n>. The negative-phase current source unit A-<m, n> may output a unit output signal O-<m, n> under control of the digital sub-signal D<m, n> and the negative-phase drive signal S-<m, n>. Unit output signals of the current sources in the current source set <NUM> form a set output signal O- of the current source set <NUM>.

In a possible implementation, as shown in <FIG>, the matching network <NUM> is separately connected to a first end p1 of an output signal cable <NUM> in the current source set <NUM> and a first end q1 of an output signal cable <NUM> in the current source set <NUM>. The matching network <NUM> may perform impedance matching on the set output signal O+ of the current source set <NUM> and the set output signal O- of the current source set <NUM>.

For example, in the differential radio frequency front-end <NUM>, a circuit structure of the matching network <NUM> may be shown in <FIG>. Specifically, the matching network <NUM> includes a capacitor Cp, an inductor L<NUM>, an inductor L<NUM>, an inductor L<NUM>, an inductor L<NUM>, and a capacitor Co. One end of the capacitor Cp is connected to one end of the inductor L<NUM>, and the end of the capacitor Cp connected to the inductor L<NUM> may receive the set output signal O+ of the current source set <NUM>. The other end of the inductor L<NUM> is connected to one end of the inductor L<NUM>. The other end of the inductor L<NUM> may further receive a supply voltage Vi. The other end of the inductor L<NUM> is connected to the other end of the capacitor Cp. The other end of the capacitor Cp may receive the set output signal O- of the current source set <NUM>. The inductor L<NUM> is magnetically coupled to the inductor L<NUM>, and the inductor L<NUM> is magnetically coupled to the inductor L<NUM>. One end of the inductor L<NUM> is connected to one end of the capacitor Co, and the other end of the inductor L<NUM> is connected to one end of the inductor L<NUM>. The other end of the inductor L<NUM> and the other end of the capacitor Co are grounded. The end of the capacitor Co connected to the inductor L<NUM> may output an RF output signal RFout.

In the matching network <NUM> shown in <FIG>, capacitance values of the capacitor Cp and the capacitor Co, and inductance values of the inductor L<NUM>, the inductor L<NUM>, the inductor L<NUM>, and the inductor L<NUM> are configured based on load impedance RL of the radio frequency front-end <NUM> and optimal load impedance Zopt of the current source set <NUM>.

Then, the control circuit <NUM> is further described by using the single-ended radio frequency front-end <NUM> as an example. It should be noted that, unless otherwise specified, the following implementation of the control circuit <NUM> is also applicable to the differential type radio frequency front-end <NUM>.

The radio frequency transmitter provided in this embodiment of this application may be a digital transmitter. In other words, the control circuit <NUM> may receive a digital signal and generate a plurality of control signals based on the received digital signal. Compared with conventional analog transmitters, digital transmitters have characteristics such as a high integration level, high efficiency and a high power.

As described above, the control signals include a digital sub-signal and a drive signal. For example, as shown in <FIG>, the control circuit <NUM> in the digital transmitter mainly includes an encoder <NUM>, a radio frequency signal source <NUM>, and a drive circuit <NUM>. The encoder <NUM> may separately provide a plurality of digital sub-signals for the drive circuit <NUM> and the current source set <NUM>. Specifically, the encoder <NUM> may directly send the plurality of digital sub-signals to the current source set <NUM>, or the drive circuit <NUM> may forward received plurality of digital sub-signals to the current source set <NUM>. This is not limited in this embodiment of this application.

The radio frequency signal source <NUM> may provide a radio frequency input signal to the drive circuit <NUM>. The drive circuit <NUM> may generate a plurality of drive signals based on the radio frequency input signal and the plurality of digital sub-signals.

Generally, a digital polarized transmitter and a digital orthogonal transmitter are two common digital transmitters. The following separately uses the digital polarized transmitter and the digital orthogonal transmitter as examples for description.

In the digital orthogonal transmitter, the encoder <NUM> may receive an orthogonal baseband signal. For example, the orthogonal baseband signal includes a baseband signal I and a baseband signal Q that are orthogonal to each other. The baseband signal I may be represented as I<NUM>,. , and IB, and the baseband signal Q may be represented as Q<NUM>,. B represents a quantity of bits of the baseband signal I and the baseband signal Q. The encoder <NUM> may encode the orthogonal baseband signal, and convert the orthogonal baseband signal into the foregoing plurality of digital sub-signals based on a quantity of current source units in the current source set <NUM>.

The radio frequency signal source <NUM> may generate orthogonal radio frequency signals CKI and CKQ. For example, as shown in <FIG>, the radio frequency signal source <NUM> includes a local-frequency signal source, an orthogonal generator, and a symbol mapping circuit. The local-frequency signal source may generate a local-frequency signal LO. The orthogonal generator may generate, based on the local-frequency signal LO, local-frequency signals LOI and LOQ that are orthogonal to each other. The symbol mapping circuit may convert the local-frequency signals LOI and LOQ into the orthogonal radio frequency signal CKI and the orthogonal radio frequency signal CKQ based on symbol signals (CI and CQ) of the orthogonal baseband signals.

The drive circuit <NUM> may convert the plurality of digital sub-signals, the orthogonal radio frequency signal CKI, and the orthogonal radio frequency signal CKQ into the foregoing plurality of drive signals. The orthogonal radio frequency signal CKI is used to generate a first drive signal SI, and the orthogonal radio frequency signal CKQ is used to generate a second drive signal SQ. Specifically, in the digital orthogonal transmitter, the plurality of drive signals generated by the drive circuit <NUM> include a plurality of first drive signals S<NUM> and a plurality of second drive signals SQ.

In the current source set <NUM>, some current source units are in a one-to-one correspondence with the plurality of first drive signals SI, and this part of current source units may also be referred to as first current source units. In the current source set <NUM>, another part of current source units are in a one-to-one correspondence with the second drive signal SQ, and this part of current source units may also be referred to as second current source units. A plurality of first current source units are in a one-to-one correspondence with a plurality of second current source units in the current source set <NUM>.

For example, as shown in <FIG>, a first current source unit AI corresponds to a second current source unit AQ, and a first drive signal S<NUM> that controls the first current source unit AI corresponds to a second drive signal SQ that controls the second current source unit AQ. A unit output signal of the first current source unit AI is orthogonal to a unit output signal of the second current source unit AQ, and the unit output signals of the first current source unit AI and the second current source unit AQ may be used to generate a sub-signal o of an output signal O of the current source set <NUM>. It may also be understood that, in the current source set <NUM>, unit output signals of the plurality of first current source units and output signals of the plurality of second current source units are superimposed at the output signal cable <NUM>, to generate the output signal O of the current source set <NUM>.

It should be noted that, if the radio frequency front-end <NUM> belongs to the differential type, the orthogonal radio frequency signal CKI generated by the symbol mapping circuit includes a positive-phase orthogonal radio frequency signal CKI+ and a negative-phase orthogonal radio frequency signal CKI-. The orthogonal radio frequency signal CKQ includes a positive-phase orthogonal radio frequency signal CKQ+ and a negative-phase orthogonal radio frequency signal CKQ-. The positive-phase orthogonal radio frequency signal CKI+ is used to generate a first drive signal SI+ of the first current source units in the current source set <NUM>. The negative-phase orthogonal radio frequency signal CKI- is used to generate a first drive signal SI- of the first current source units in the current source set <NUM>. The positive-phase orthogonal radio frequency signal CKQ+ is used to generate a second drive signal SQ+ of the second current source units in the current source set <NUM>. The negative-phase orthogonal radio frequency signal CKQ- is used to generate a second drive signal SQ- of the second current source units in the current source set <NUM>.

For example, <FIG> shows a correspondence between a current source unit and a drive signal in the current source set <NUM> and the current source set <NUM> in a case in which the digital orthogonal transmitter includes the differential radio frequency front-end <NUM>. Specifically, a current source unit AI+ is any first positive-phase current source unit in the current source set <NUM>, and a current source unit AQ+ is a second positive-phase current source unit that is in the current source set <NUM> and that corresponds to the current source unit AI+. A current source unit AI- is a first negative-phase current source unit that is in the current source set <NUM> and that corresponds to the current source unit AI+. A current source unit AQ- is a second negative-phase current source unit that is in the current source set <NUM> and that corresponds to the current source unit AI-. In addition, a correspondence also exists between the current source unit AI- and the current source unit AQ+.

In <FIG>, a digital signal D<NUM> and the first drive signal SI+ are used to control the current source unit AI+. A digital signal DQ and a second drive signal SQ+ are used to control the current source unit AQ+. The unit output signals of the current source unit AI+ and the current source unit AQ+ form a sub-signal o+ of a set output signal O+ of the current source set <NUM>.

The digital signal DI and the first drive signal SI- are used to control the current source unit AI-. The digital signal DQ and a second drive signal SQ- are used to control the current source unit AQ-. The unit output signals of the current source unit AI- and the current source unit AQ- form a sub-signal o- of a set output signal O- of the current source set <NUM>. The sub-signal o+ and the sub-signal o- are reverse signals of each other.

Still refer to <FIG>. In the digital polarized transmitter, the encoder <NUM> may receive a baseband amplitude signal a. For example, the baseband amplitude signal may be represented as a<NUM>,. The encoder <NUM> may encode the baseband amplitude signal a, and convert the baseband amplitude signal a into the foregoing plurality of digital sub-signals based on a quantity of current source units in the current source set <NUM>.

The radio frequency signal source <NUM> may generate a phase modulation signal PM. For example, as shown in <FIG>, the radio frequency signal source <NUM> includes a local-frequency signal source, an orthogonal generator, and a phase modulator. The local-frequency signal source may generate a local-frequency signal LO. The orthogonal generator may perform phase modulation on the local-frequency signal to obtain local-frequency signals LOI and LOQ that are orthogonal to each other. The phase modulator may convert the local-frequency signals LOI and LOQ into the phase modulation signal PM based on a baseband phase signal φ corresponding to the baseband amplitude signal a. Both of the baseband amplitude signal a and the baseband phase signal φ are obtained based on a baseband signal input to the radio frequency transmitter <NUM>.

The drive circuit <NUM> may convert the plurality of digital sub-signals provided by the encoder <NUM> and the phase modulation signal PM provided by the radio frequency signal source <NUM> into the plurality of drive signals.

The foregoing describes a basic architecture of the radio frequency transmitter <NUM>. However, whether the digital orthogonal transmitter, the digital polarization transmitter, or another radio frequency transmitter implemented based on a current source set operates in a millimeter wave band, a problem of low efficiency may occur.

Specifically, in a case in which the radio frequency transmitter <NUM> operates in the millimeter wave band, a complex interconnection cable in the current source set <NUM> causes a severe parasitic effect. This reduces efficiency of the radio frequency transmitter <NUM>. As shown in <FIG>, a current source subset (<NUM>, <NUM>,. , and 1021N) is connected in parallel to the output signal cable <NUM>. Due to the parasitic effect, a parasitic capacitor and a parasitic inductor appear between adjacent current source subsets. For example, a parasitic inductor Lk1 is connected in series between the current source subset <NUM> and the current source subset <NUM>, and a parasitic capacitor Ck1 is connected in parallel between the current source subset <NUM> and the current source subset <NUM>. Similarly, a parasitic inductor Lk2 is connected in series between the current source subset <NUM> and the current source subset <NUM> (not shown in the figure), and a parasitic capacitor Ck2 is connected in parallel between the current source subset <NUM> and the current source subset <NUM>.

Due to existence of the parasitic inductor and the parasitic capacitor, load impedance of different current source subsets in the current source set <NUM> is different. In other words, an impedance mismatch occurs in the current source set <NUM>. The impedance mismatch increases a power loss of the current source set <NUM>. This reduces the efficiency of the radio frequency transmitter <NUM>.

In view of this, as shown in <FIG>, the radio frequency transmitter <NUM> provided in this embodiment of this application further includes a compensation circuit <NUM>, and the compensation circuit <NUM> is connected to a second end p2 of the output signal cable. The compensation circuit <NUM> may compensate for a difference of load impedance between current source subsets in the current source set <NUM>. Therefore, disposing the compensation circuit <NUM> in the radio frequency transmitter <NUM> helps reduce the difference of load impedance between current source subsets. This helps improve the efficiency of the radio frequency transmitter <NUM>.

Then, the compensation circuit <NUM> is further described by using a single-ended type radio frequency front-end and a differential type radio frequency front-end as examples.

As shown in <FIG>, in the single-ended radio frequency front-end <NUM>, the compensation circuit <NUM> includes a compensation inductor Ls1 and a compensation capacitor Cs1. The compensation circuit <NUM> is grounded through the compensation inductor Ls1 and the compensation capacitor Cs1. Specifically, one end of the compensation inductor Ls1 is connected to a second end p2 of the output signal cable <NUM>. The other end of the compensation inductor Ls1 is connected to one end of the compensation capacitor Cs1. The other end of the compensation capacitor Cs1 is grounded. Specifically, grounding in this application refers to ground potential of an alternating-current signal. For example, for a single-ended circuit, ground potential may be ground potential of a direct-current signal or power potential of a direct-current signal, and for a differential circuit, ground potential may be intermediate potential of a differential signal.

The compensation inductor Ls1 and the compensation capacitor Cs1 may form an LC resonant circuit, so that two transmission zeros fz1 and fz2 are added to a load impedance curve of each current source subset. Between the two transmission zeros fz1 and fz2, load impedance of each current source subset increases as an operating frequency increases. After reaching a maximum value, the load impedance of each current source subset decreases as the operating frequency increases.

An inductance value of the compensation inductor Ls1 and a capacitance value of the compensation capacitor Cs1 are properly configured, so that the load impedance of each current source subset may form a maximum value close to optimal load impedance Zopt between the two transmission zero points fz1 and fz2, an operating frequency corresponding to the maximum value is within an operating band of the radio frequency transmitter <NUM>, therefore, the difference of load impedance of each current source subset is reduced.

In addition, the inductance value of the compensation inductor Ls1 and the capacitance value of the compensation capacitor Cs1 are properly configured, so that the load impedance of each current source subset may be adjusted to a value close to the optimal load impedance Zopt in the operating frequency band. It can be learned that, by using the compensation circuit <NUM> provided in this embodiment of this application, load impedance of the current source subsets can be close to same load impedance. This helps reduce a difference of load impedance between the current source subsets, and improve efficiency of the radio frequency transmitter <NUM>. In addition, because the load impedance of each current source subset after compensation is close to same load impedance, namely, the optimal load impedance Zopt, the compensation circuit <NUM> provided in this embodiment of this application further helps increase an output power of the radio frequency transmitter <NUM>.

In this embodiment of this application, the compensation inductor Ls1 may be an inductor with an adjustable inductance value or may be an inductor with a non-adjustable inductance value. The compensation capacitor Cs1 may be a capacitor with an adjustable capacitance value or may be a capacitor with a non-adjustable capacitance value. In a case in which at least one of the compensation inductor Ls1 and the compensation capacitor Cs1 is an adjustable element (an adjustable inductor or an adjustable capacitor), values of the transmission zero points fz1 and fz2 may be flexibly adjusted by adjusting a value of the adjustable element. This helps flexibly adjust an operating bandwidth of the radio frequency transmitter <NUM>. For example, in a case in which fz1 is less than fz2, increasing a value of fz1 helps increase the operating bandwidth of the radio frequency transmitter <NUM>. This improves performance of the radio frequency transmitter <NUM> in a higher frequency range.

The differential radio frequency front-end <NUM> includes two current source sets: the current source set <NUM> (a positive-phase current source set) and the current source set <NUM> (a negative-phase current source set).

In a possible implementation, the radio frequency front-end may include two compensation circuits. One compensation circuit is connected to the second end p2 of the output signal cable <NUM>, and is configured to compensate for a difference of load impedance between the N current source subsets in the current source set <NUM>. The other compensation circuit is connected to a second end q2 of the output signal cable <NUM>, and is configured to compensate for a difference of load impedance between the N current source subsets in the current source set <NUM>. For specific structures of the two compensation circuits, refer to the compensation circuit <NUM> shown in <FIG>.

In another possible implementation, as shown in <FIG>, one end of the compensation circuit <NUM> is connected to the second end p2 of the output signal cable <NUM>, and the other end of the compensation circuit <NUM> is connected to the second end q2 of the output signal cable <NUM>. The compensation circuit <NUM> may not only compensate for a difference of load impedance between the N current source subsets in the current source set <NUM>, but also compensate for a difference of load impedance between the N current source subsets in the current source set <NUM>.

For example, as shown in <FIG>, the compensation circuit <NUM> includes a compensation inductor Ls2, a compensation capacitor Cs2, and a compensation inductor Ls3. The compensation inductor Ls2, the compensation capacitor Cs2, and the compensation inductor Ls3 are sequentially connected in series. Specifically, one end of the compensation inductor Ls2 is connected to the second end p2 of the output signal cable <NUM>, and the other end of the compensation inductor Ls2 is connected to one end of the compensation capacitor Cs2. The other end of the compensation capacitor Cs2 is connected to one end of the compensation inductor Ls3, and the other end of the compensation inductor Ls3 is connected to the second end q2 of the output signal cable <NUM>.

It may be understood that polarities of two ends of the compensation capacitor Cs2 are opposite. Therefore, an electric potential <NUM> point exists between the two ends of the compensation capacitor Cs2. It may also be understood that a virtual ground exists between the two ends of the compensation capacitor Cs2. Therefore, the compensation circuit <NUM> may have a compensation effect similar to that of the compensation circuit <NUM> in <FIG> for both of the current source set <NUM> and the current source set <NUM>.

In this embodiment of this application, the compensation inductor Ls2 may be an inductor with an adjustable inductance value or may be an inductor with a non-adjustable inductance value. The compensation capacitor Cs2 may be a capacitor with an adjustable capacitance value or may be a capacitor with a non-adjustable capacitance value. The compensation inductor Ls3 may be an inductor with an adjustable inductance value or may be an inductor with a non-adjustable inductance value. In a case in which at least one of the compensation inductor Ls2, the compensation capacitor Cs2, and the compensation inductor Ls3 is an adjustable element (the adjustable inductor or the adjustable capacitor), values of the transmission zero points fz1 and fz2 may be flexibly adjusted by adjusting a value of the adjustable element. This helps flexibly adjust an operating bandwidth of the radio frequency transmitter <NUM>.

In conclusion, in this embodiment of this application, the compensation circuit <NUM> is disposed in the radio frequency transmitter <NUM>, to compensate for a difference of load impedance between the N current source subsets in the current source set. This improves efficiency of the radio frequency transmitter <NUM>.

Based on a same technical concept, an embodiment of this application further provides a current source array. The current source array may be configured to implement any radio frequency transmitter provided in the foregoing embodiments. For example, the current source array may be applied to a digital polarized transmitter or a digital orthogonal transmitter. The current source array may be used to implement a single-ended radio frequency front-end or a differential radio frequency front-end. The following separately describes the foregoing scenarios.

As shown in <FIG>, the current source array includes F rows of current source units, an output signal cable <NUM>, a compensation circuit <NUM>, and E branch signal cables <NUM>. In a possible implementation, a substrate <NUM> may be further included. The substrate <NUM> may be configured to carry the F rows of current source units, the output signal cable <NUM>, the compensation circuit <NUM>, and the E branch signal cables <NUM>.

A blank grid in the current source array represents a current source unit. In a case in which the radio frequency front-end of the digital polarized transmitter belongs to the single-ended type, the compensation circuit <NUM> includes a compensation inductor LS1 and a compensation capacitor CS1. One end of the compensation inductor LS1 is connected to a first end of the output signal cable <NUM>, and the other end of the compensation inductor LS1 is connected to one end of the compensation capacitor CS1. The other end of the compensation capacitor CS1 is grounded.

Refer to <FIG> and <FIG>. The F rows of current source units in <FIG> may be equivalent to the current source units in the current source set <NUM> in <FIG>, and the output signal cable <NUM> in <FIG> may be equivalent to the output signal cable <NUM> in <FIG>.

As shown in <FIG>, the E branch signal cables <NUM> are disposed between the F rows of current source units, and the E branch signal cables are all parallel to a row arrangement direction of the current source units. In addition, one or more rows of current source units are spaced between any adjacent branch signal cables <NUM>. The branch signal cable <NUM> may transmit a unit output signal of a current source unit adjacent to the branch signal cable <NUM> to the output signal cable <NUM>. A first end p1 of the output signal cable <NUM> may output a set output signal O, and a second end p2 of the output signal cable <NUM> is connected to the compensation inductor LS1.

As shown in <FIG>, the current source array further includes an output signal cable <NUM>. In the F rows of current source units, K branch signal cables <NUM> are connected to a first output signal cable, and E - K branch signal cables <NUM> are connected to the output signal cable <NUM>, where K is an integer greater than or equal to <NUM>. Generally, in the structure shown in <FIG>, K = F / <NUM>, and F is an even number.

A current source unit connected to the output signal cable <NUM> by using the branch signal cable <NUM> may be understood as a positive-phase current source unit A+. A current source unit connected to the output signal cable <NUM> by using the branch signal cable <NUM> may be understood as a negative-phase current source unit A-.

Generally, a row of current source units formed by the positive-phase current source unit A+ and the negative-phase current source unit A- may be alternately arranged. In a case in which the radio frequency front-end of the digital polarized transmitter belongs to the differential type, the compensation circuit <NUM> may include a compensation inductor Ls2, a compensation capacitor CS2, and a compensation inductor LS3. The compensation inductor Ls2, the compensation capacitor CS2, and the compensation inductor LS3 are sequentially connected in series.

As shown in <FIG>, in this embodiment of this application, the compensation inductor LS2 may be disposed adjacent to the output signal cable <NUM>, the compensation inductor LS3 may be disposed adjacent to the output signal cable <NUM>, and the compensation capacitor CS2 may be disposed between the compensation inductor LS2 and the compensation inductor LS3. A disposing manner shown in <FIG> is used to help reduce a cabling length.

Refer to <FIG> and <FIG>. In the current source array in <FIG>, a plurality of rows of current source units connected to the output signal cable <NUM> may be equivalent to the current source set <NUM> in <FIG>, where the current source units are positive-phase current source units A+. A plurality of rows of current source units connected to the output signal cable <NUM> may be equivalent to the current source set <NUM> in <FIG>, where the current source units are negative-phase current source units A-.

The output signal cable <NUM> is equivalent to the output signal cable <NUM> in <FIG>, and may receive a unit output signal of the positive-phase current source unit A+, to output a set output signal O+ of the current source set <NUM>. The output signal cable <NUM> is equivalent to the output signal cable <NUM> in <FIG>, and may receive a unit output signal of the negative-phase current source unit A-, to output a set output signal O- of the current source set <NUM>.

As shown in <FIG>, the output signal cable <NUM> and the output signal cable <NUM> are disposed perpendicular to a row arrangement direction of the F rows of current source units, and the output signal cable <NUM> and the output signal cable <NUM> are disposed adjacent to each other. This disposing manner is used to help reduce a cabling length of the branch signal cables <NUM>.

As shown in <FIG>, the first end p1 of the output signal cable <NUM> and a first end q1 of the output signal cable <NUM> are disposed on a side close to a first row of current source unit in the F rows of current source units. The compensation circuit <NUM> is disposed on a side close to an Fth row of current source unit in the F rows of current source units. In other words, F rows of current source units are spaced between the first end p1 and the second end p2 of the output signal cable <NUM>. The same applies to the output signal cable <NUM>. This disposing manner is used to help reduce cabling lengths of the output signal cable <NUM> and the output signal cable <NUM>.

As shown in <FIG>, the current source array includes a plurality of first current source unit AI and a plurality of second current source unit AQ. For any row of current source units, the first current source unit AI and the second current source unit AQ are alternately disposed. The E branch signal cables <NUM> in the current source array are connected to the output signal cable <NUM>. The current source array shown in <FIG> is similar to that in <FIG>. Details are not described again.

The current source array shown in <FIG> has a structure similar to that of the current source array shown in <FIG>, and a difference lies in that any row of positive-phase current source units A+ further includes a first current source unit AI+ and a second current source unit AQ+ that are alternately disposed. Any row of negative-phase current source units A- further includes a first current source unit AI- and a second current source unit AQ- that are alternately disposed.

In a possible implementation, the current source array may further include another H rows of current source units and G branch signal cables <NUM>, where H and G are integers greater than <NUM>, to increase a quantity of current source units. For example, the radio frequency front-end is a differential digital orthogonal transmitter, and a current source array applicable to the digital orthogonal transmitter may be shown in <FIG>.

The G branch signal cables <NUM> are disposed between the H rows of current source units, and the G branch signal cables <NUM> are parallel to a row arrangement direction of the H rows of current source units. In the G branch signal cables <NUM>, one or more rows of current source units are spaced between any two adjacent branch signal cables <NUM>. In the G branch signal cables <NUM>, L branch signal cables <NUM> are connected to the output signal cable <NUM>, and G - L second branch signal cables are connected to the output signal cable <NUM>, where L is an integer greater than or equal to <NUM>. In this case, the compensation circuit <NUM> may further compensate for a difference of load impedance between a plurality of rows of current source units adjacent to the branch signal cable <NUM>.

As shown in <FIG>, the output signal cable <NUM> and the output signal cable <NUM> may be disposed between the F rows of current source units and the H rows of current source units, to reduce a cabling length. It may also be understood that the F rows of current source units are disposed on a side that is of the output signal cable <NUM> and that is far from the output signal cable <NUM>, and the H rows of current source units are disposed on a side that is of the output signal cable <NUM> and that is far from the output signal cable <NUM>.

Although some embodiments of this application have been described, persons skilled in the art can make changes and modifications to these embodiments once they learn the basic inventive concept. Therefore, the following claims are intended to be construed as to cover the preferred embodiments and all changes and modifications falling within the scope of this application.

Claim 1:
A current source array (<NUM>), comprising F rows of current source units, a first output signal cable (<NUM>), a second output signal cable (<NUM>), E first branch signal cables (<NUM>), and a compensation circuit (<NUM>), wherein both F and E are integers greater than <NUM>;
the E first branch signal cables (<NUM>) are disposed between the F rows of current source units, and the E first branch signal cables (<NUM>) are parallel to a row arrangement direction of the F rows of current source units;
in the E first branch signal cables (<NUM>), one or more rows of current source units are spaced between any two adjacent first branch signal cables (<NUM>);
in the E first branch signal cables (<NUM>), K first branch signal cables (<NUM>) are connected to the first output signal cable (<NUM>), and E - K first branch signal cables (<NUM>) are connected to the second output signal cable (<NUM>), wherein K is an integer greater than or equal to <NUM>;
a first end (p1) of the first output signal cable (<NUM>) is configured to output a first output signal of the current source array, and a first end (q1) of the second output signal (<NUM>) cable is configured to output a second output signal of the current source array;
both of a second end (p2) of the first output signal cable (<NUM>) and a second end (q2) of the second output signal cable (<NUM>) are connected to the compensation circuit (<NUM>); and
the compensation circuit (<NUM>) is configured to compensate for a difference of load impedance between a plurality of rows of current source units adjacent to the first branch signal cable (<NUM>).