Patent Description:
An early example of integrated radar and communication subsystem can be traced back to the NASA Space Shuttle Orbiter, where a Ku-band radio was either operated as a radar during rendezvous maneuvers or as a <NUM>-way ground-to-shuttle communications system. The first OFDM radar (referred to as Multi-Carrier Phase-Coded (MCPC) radar) was not originally motivated by the possibility of simultaneous communication and radar capabilities, it was however later recognized as a viable option for combination with OFDM communications. The use of <NUM>. 11ad WiFi for radar appears to have been proposed in <NUM>. Current embodiments of this idea focus on monostatic radars, with or without antenna arrays for angular resolution.

Ongoing interest appears to be driven by the trend towards intelligent transportation and autonomous vehicles, their need for situational awareness, and the advantages that can be gained from vehicle-to-vehicle (V2V) communication. Joining communication and radar functionality potentially removes the need for separate radios for both. For the vehicular use case with attention to long range, it is claimed that a single target can be located at up to <NUM> range while Gb/s data rates are achieved simultaneously. Angular resolution from the low number of antenna elements in commercial <NUM>. 11ad modules is not sufficient, however.

<CIT> discloses an apparatus for measuring the position of a vehicle or a surface thereof on a roadway. The device of the <CIT> is a device that is fixed in one place, not a mobile device, and it does not disclose a configuration in which the transmit antenna and the receive antenna are determined based on the orientation of the mobile device. <CIT> fails to disclose the feature directed to "configure at least one RF antenna, among the plurality of RF antennas (<NUM>), as a TX antenna and at least one of the remaining RF antennas as at least one RX antenna based on an orientation of the mobile device (<NUM>).

<CIT> relates to tracking objects via radio reflections. <CIT> merely discloses elliptic positioning. <CIT> does not suggest or teach that determining the transmit antenna and the receive antenna based on the orientation of the mobile device. Thus, also <CIT> fails to disclose that the system is a mobile device. Thus, <CIT> fails to disclose the feature directed to "configure at least one RF antenna, among the plurality of RF antennas (<NUM>), as a TX antenna and at least one of the remaining RF antennas as at least one RX antenna based on an orientation of the mobile device (<NUM>).

<CIT> provides a network waveform system that can include multiple radars disposed at different geographical positions within an environment.

<CIT> a system and method for determining the position of an object in relation to a positioning system using a sparse antenna array.

<CIT> discloses methods and systems for calibration of a multifunctional automotive radar system.

The present disclosure provides <NUM>-dimensional, short-distance ranging to mobile devices, without the requirement for dedicated hardware components beyond what is already available in <NUM> mmWave or <NUM>. 11ad enabled devices.

The ranging capability may be used for gesture recognition in close distance to the device, for depth sensing at longer distances, liveliness detection, detection of bio-signals that have a range-component, such as breathing and heartbeat, or to increase the sensing quality when combining camera images with depth information.

In a first embodiment, a mobile device is provided as defined by the appended set of claims.

In a second embodiment, a method for operating a mobile device comprising a plurality of RF antennas configured to transmit (TX) or receive (RX) at least one RF signal is provided as defined by the appended set of claims.

For a more complete understanding of the disclosure and its advantages, reference is now made to the following description, taken in conjunction with the accompanying drawings, in which:.

<FIG>, discussed herein, and the various embodiments used to describe the principles of the disclosure in this patent document are by way of illustration only and should not be construed in any way to limit the scope of the disclosure. Those skilled in the art will understand that the principles of the disclosure may be implemented in any suitably arranged device or system.

<FIG> illustrates an example wireless network <NUM> according to one embodiment of the disclosure. The embodiment of the wireless network <NUM> shown in <FIG> is for illustration only.

The wireless network <NUM> includes an eNodeB (eNB) <NUM>, an eNB <NUM>, and an eNB <NUM>. The eNB <NUM> communicates with the eNB <NUM> and the eNB <NUM>. The eNB <NUM> also communicates with at least one Internet Protocol (IP) network <NUM>, such as the Internet, a proprietary IP network, or other data network.

Depending on the network type, the term "base station" or "BS" can refer to any component (or collection of components) configured to provide wireless access to a network, such as transmit point (TP), transmit-receive point (TRP), an enhanced base station (eNodeB or eNB or gNB) , a macrocell, a femtocell, a WiFi access point (AP) , or other wirelessly enabled devices. Base stations may provide wireless access in accordance with one or more wireless communication protocols, e.g., <NUM> 3GPP New Radio Interface/Access (NR), long term evolution (LTE) , LTE advanced (LTE-A) , High Speed Packet Access (HSPA), Wi-Fi <NUM>. <NUM>1a/b/g/n/ac, etc. For the sake of convenience, the terms "BS" and "TRP" are used interchangeably in this patent document to refer to network infrastructure components that provide wireless access to remote terminals. Also, depending on the network type, the term "user equipment" or "UE" can refer to any component such as "mobile station," "subscriber station," "remote terminal," "wireless terminal," "receive point," or "user device. " For the sake of convenience, the terms "user equipment" and "UE" are used in this patent document to refer to remote wireless equipment that wirelessly accesses a BS, whether the UE is a mobile device (such as a mobile telephone or smartphone) or is normally considered a stationary device (such as a desktop computer or vending machine).

The eNB <NUM> provides wireless broadband access to the network <NUM> for a first plurality of user equipments (UEs) within a coverage area <NUM> of the eNB <NUM>. The first plurality of UEs includes a UE <NUM>, which may be located in a small business (SB); a UE <NUM>, which may be located in an enterprise (E); a UE <NUM>, which may be located in a WiFi hotspot (HS); a UE <NUM>, which may be located in a first residence (R); a UE <NUM>, which may be located in a second residence (R); and a UE <NUM>, which may be a mobile device (M) like a cell phone, a wireless laptop, a wireless PDA, or the like. The eNB <NUM> provides wireless broadband access to the network <NUM> for a second plurality of UEs within a coverage area <NUM> of the eNB <NUM>. The second plurality of UEs includes the UE <NUM> and the UE <NUM>. In some embodiments, one or more of the eNBs <NUM>-<NUM> may communicate with each other and with the UEs <NUM>-<NUM> using <NUM>, long-term evolution (LTE), LTE-A, WiMAX, or other advanced wireless communication techniques.

It should be clearly understood that the coverage areas associated with eNBs, such as the coverage areas <NUM> and <NUM>, may have other shapes, including irregular shapes, depending upon the configuration of the eNBs and variations in the radio environment associated with natural and man-made obstructions.

As described in more detail below, one or more of BS <NUM>, BS <NUM> and BS <NUM> include 2D antenna arrays as described in embodiments of the disclosure. In some embodiments, one or more of BS <NUM>, BS <NUM> and BS <NUM> support the codebook design and structure for systems having 2D antenna arrays.

Although <FIG> illustrates one example of a wireless network <NUM>, various changes may be made to <FIG>. For example, the wireless network <NUM> could include any number of eNBs and any number of UEs in any suitable arrangement. Also, the eNB <NUM> could communicate directly with any number of UEs and provide those UEs with wireless broadband access to the network <NUM>. Similarly, each eNB <NUM>-<NUM> could communicate directly with the network <NUM> and provide UEs with direct wireless broadband access to the network <NUM>. Further, the eNB <NUM>, <NUM>, and/or <NUM> could provide access to other or additional external networks, such as external telephone networks or other types of data networks.

<FIG> and <FIG> illustrate example wireless transmit and receive paths according to one embodiment of the disclosure. In the following description, a transmit path <NUM> may be described as being implemented in an eNB (such as eNB <NUM>), while a receive path <NUM> may be described as being implemented in a UE (such as UE <NUM>). However, it will be understood that the receive path <NUM> could be implemented in an eNB and that the transmit path <NUM> could be implemented in a UE. In some embodiments, the receive path <NUM> is configured to support the codebook design and structure for systems having 2D antenna arrays as described in embodiments of the disclosure.

The transmit path <NUM> includes a channel coding and modulation block <NUM>, a serial-to-parallel (S-to-P) block <NUM>, a size N Inverse Fast Fourier Transform (IFFT) block <NUM>, a parallel-to-serial (P-to-S) block <NUM>, an add cyclic prefix block <NUM>, and an up-converter (UC) <NUM>. The receive path <NUM> includes a down-converter (DC) <NUM>, a remove cyclic prefix block <NUM>, a serial-to-parallel (S-to-P) block <NUM>, a size N Fast Fourier Transform (FFT) block <NUM>, a parallel-to-serial (P-to-S) block <NUM>, and a channel decoding and demodulation block <NUM>.

In the transmit path <NUM>, the channel coding and modulation block <NUM> receives a set of information bits, applies coding (such as a low-density parity check (LDPC) coding), and modulates the input bits (such as with Quadrature Phase Shift Keying (QPSK) or Quadrature Amplitude Modulation (QAM)) to generate a sequence of frequency-domain modulation symbols. The serial-to-parallel block <NUM> converts (such as de-multiplexes) the serial modulated symbols to parallel data in order to generate N parallel symbol streams, where N is the IFFT/FFT size used in the eNB <NUM> and the UE <NUM>. The size N IFFT block <NUM> performs an IFFT operation on the N parallel symbol streams to generate time-domain output signals. The parallel-to-serial block <NUM> converts (such as multiplexes) the parallel time-domain output symbols from the size N IFFT block <NUM> in order to generate a serial time-domain signal. The add cyclic prefix block <NUM> inserts a cyclic prefix to the time-domain signal. The up-converter <NUM> modulates (such as up-converts) the output of the add cyclic prefix block <NUM> to an RF frequency for transmission via a wireless channel. The signal may also be filtered at baseband before conversion to the RF frequency.

A transmitted at least one RF signal from the eNB <NUM> arrives at the UE <NUM> after passing through the wireless channel, and reverse operations to those at the eNB <NUM> are performed at the UE <NUM>. The down-converter <NUM> down-converts the received signal to a baseband frequency, and the remove cyclic prefix block <NUM> removes the cyclic prefix to generate a serial time-domain baseband signal. The serial-to-parallel block <NUM> converts the time-domain baseband signal to parallel time domain signals. The size N FFT block <NUM> performs an FFT algorithm to generate N parallel frequency-domain signals. The parallel-to-serial block <NUM> converts the parallel frequency-domain signals to a sequence of modulated data symbols. The channel decoding and demodulation block <NUM> demodulates and decodes the modulated symbols to recover the original input data stream.

Each of the eNBs <NUM>-<NUM> may implement a transmit path <NUM> that is analogous to transmitting in the downlink to UEs <NUM>-<NUM> and may implement a receive path <NUM> that is analogous to receiving in the uplink from UEs <NUM>-<NUM>. Similarly, each of UEs <NUM>-<NUM> may implement a transmit path <NUM> for transmitting in the uplink to eNBs <NUM>-<NUM> and may implement a receive path <NUM> for receiving in the downlink from eNBs <NUM>-<NUM>.

Each of the components in <FIG> and <FIG> can be implemented using only hardware or using a combination of hardware and software/firmware. As a particular example, at least some of the components in <FIG> and <FIG> may be implemented in software, while other components may be implemented by configurable hardware or a mixture of software and configurable hardware. For instance, the FFT block <NUM> and the IFFT block <NUM> may be implemented as configurable software algorithms, where the value of size N may be modified according to the implementation.

Furthermore, although described as using FFT and IFFT, this is by way of illustration only and should not be construed to limit the scope of the disclosure. Other types of transforms, such as Discrete Fourier Transform (DFT) and Inverse Discrete Fourier Transform (IDFT) functions, could be used. It will be appreciated that the value of the variable N may be any integer number (such as <NUM>, <NUM>, <NUM>, <NUM>, or the like) for DFT and IDFT functions, while the value of the variable N may be any integer number that is a power of two (such as <NUM>, <NUM>, <NUM>, <NUM>, <NUM>, or the like) for FFT and IFFT functions.

Although <FIG> and <FIG> illustrate examples of wireless transmit and receive paths, various changes may be made to <FIG> and <FIG>. For example, various components in <FIG> and <FIG> could be combined, further subdivided, or omitted and additional components could be added according to particular needs. Also, <FIG> and <FIG> are meant to illustrate examples of the types of transmit and receive paths that could be used in a wireless network. Any other suitable architectures could be used to support wireless communications in a wireless network.

<FIG> illustrates an example UE <NUM> capable of millimeter(mm)-wave communications according to one embodiment of the disclosure. However, UEs come in a wide variety of configurations, and <FIG> does not limit the scope of the disclosure to any particular implementation of a UE.

The UE <NUM> includes an antenna <NUM>, a radio frequency (RF) transceiver <NUM>, transmit (TX) processing circuitry <NUM>, a microphone <NUM>, and receive (RX) processing circuitry <NUM>. The UE <NUM> also includes a speaker <NUM>, a main processor <NUM>, an input/output (I/O) interface (IF) <NUM>, input device(s) <NUM>, a display <NUM>, and a memory <NUM>. The memory <NUM> includes a basic operating system (OS) program <NUM> and one or more applications <NUM>.

The RF transceiver <NUM> receives, from the antenna <NUM>, an incoming at least one RF signal transmitted by an eNB of the network <NUM>. The RF transceiver <NUM> down-converts the incoming at least one RF signal to generate an intermediate frequency (IF) or baseband signal. The RX processing circuitry <NUM> transmits the processed baseband signal to the speaker <NUM> (such as for voice data) or to the main processor <NUM> for further processing (such as for web browsing data).

The TX processing circuitry <NUM> receives analog or digital voice data from the microphone <NUM> or other outgoing baseband data (such as web data, e-mail, or interactive video game data) from the main processor <NUM>. The RF transceiver <NUM> receives the outgoing processed baseband or IF signal from the TX processing circuitry <NUM> and up-converts the baseband or IF signal to at least one RF signal that is transmitted via the antenna <NUM>.

The main processor <NUM> can include one or more processors or other processing devices and execute the basic OS program <NUM> stored in the memory <NUM> in order to control the overall operation of the UE <NUM>. The main processor <NUM> controls the reception of forward channel signals and the transmission of reverse channel signals by the RF transceiver <NUM>, the RX processing circuitry <NUM>, and the TX processing circuitry <NUM> in accordance with well-known principles. In some embodiments, the main processor <NUM> includes at least one microprocessor or microcontroller.

The main processor <NUM> is also capable of executing other processes and programs resident in the memory <NUM>, such as operations for channel quality measurement and reporting for systems having 2D antenna arrays as described in embodiments of the disclosure as described in embodiments of the disclosure. The main processor <NUM> can move data into or out of the memory <NUM> as required by an executing process. In some embodiments, the main processor <NUM> is configured to execute the applications <NUM> based on the OS program <NUM> or in response to signals received from eNBs or an operator. The main processor <NUM> is also coupled to the I/O interface <NUM>, which provides the UE <NUM> with the ability to connect to other devices such as laptop computers and handheld computers. The I/O interface <NUM> is the communication path between these accessories and the main controller <NUM>.

The main processor <NUM> is also coupled to the keypad <NUM> and the display unit <NUM>. The operator of the UE <NUM> can use the keypad <NUM> to enter data into the UE <NUM>. The display <NUM> may be a liquid crystal display or other display capable of rendering text and/or at least limited graphics, such as from web sites.

The memory <NUM> is coupled to the main processor <NUM>. Part of the memory <NUM> could include a random-access memory (RAM), and another part of the memory <NUM> could include a Flash memory or other read-only memory (ROM). As a particular example, the main processor <NUM> could be divided into multiple processors, such as one or more central processing units (CPUs) and one or more graphics processing units (GPUs).

<FIG> illustrates an exemplary mm-wave communication system according to embodiments of the disclosure. The embodiment shown in <FIG> is for illustration only.

For mm-Wave bands, the number of antenna elements can be large for a given form factor due to the small wavelengths of the signal. The physical extent of antenna elements generally scales with the wavelength associated with the intended communication frequency band and typically falls in the range between ½ to <NUM> wavelength in either dimension. As an example, planar antenna elements for use in the common <NUM> or <NUM> frequency bands will typically occupy an area of (<NUM>/<NUM> wavelength)-squared, or approximately <NUM> x <NUM><NUM> (<NUM>) or <NUM> x <NUM><NUM> (<NUM>). Small arrays of such antennas are compatible with the physical constraints of handheld mobile devices and are used to the enable Gbps-level high-throughput communications at the mm-wave frequencies.

The number of digital chains is limited due to constraints on hardware size, power consumption and implementation cost, which differ between mobile device and base station. In one embodiment, it is advantageous and customary that one digital chain be mapped to a large number of antenna elements which can be controlled by a bank of analog phase shifters. One digital chain then connects to one sub-array which produces a narrow analog beam through analog beamforming. As a general rule, the angular beam-width of a square N-element array, fed by coherent electrical signals, is on the order of <MAT> [rad]. The center-peak of this analog beam can be pointed to or swept across a wide range of angles by selectively delaying element signals relative to each other, typically by adjusting a bank of phase shifters. For clarity, <FIG> only shows the signal path in transmit direction. It is readily understood by those skilled in the art that the signal paths will also include hardware in receive direction (from antennas to digital output via low-noise amplifiers, phase shifters, mixers, analog-to-digital converters, and FFT blocks).

Mobile devices, particularly hand-held smart phones pose additional challenges in that (<NUM>) the orientation of the device relative to the base station and (<NUM>) the signal path loss between any one antenna module and the base station are not known. A common approach to mitigate these issues is to provide several antenna arrays with different physical placements and orientations on the mobile device, such that e.g. blockage of one or few of the modules by e.g. the users hand or body can be mitigated by enabling one or several of the other modules.

<FIG> illustrates an exemplary multi-module analog beamforming for mobile device mm-wave communications according to one embodiment of the disclosure. The embodiment shown in <FIG> is for illustration only.

Architecturally, a straightforward approach to low-complexity implementations of mm-wave communications systems with dynamically configurable front-end modules is shown in <FIG>. At any given time, one <NUM> of <NUM> modules can be in transmit mode and at least one of the remaining <NUM> modules <NUM>, <NUM>, <NUM> can be in receive mode. Other, more complex methods are possible that allow simultaneous transmission or reception from more than one module, but not the same due to lack of antenna isolation, mostly. In a maximum configuration, the system may be designed to allow transmission and reception through part or all of modules simultaneously. Considering power consumption and hardware complexity scale accordingly, the system might be designed to have the minimum switched configuration.

In one embodiment, there are very few elements per each module, such as <NUM> antenna elements. Realizable angular (half power-) beam widths from <NUM> elements are on the order of <NUM> deg. to <NUM> deg. While this is advantageous for the good spatial coverage in the mobile use case, it also means that any signal arriving e.g. at a receiver module cannot be assigned an accurate Direction-of-Arrival (DoA) value.

In radar systems, considerable effort has been applied to the topic of the ideal radar waveform. While simple pulsed radar was used for early implementations and radar functionality is intuitive in this case, increased range resolution requires shorter pulses, and shorter pulses contain less energy, which reduces detection range. As a first improvement, the frequency modulation continuous wave (FM-CW) radar waveform was introduced. Here, a linear frequency ramp is transmitted that sweeps across a bandwidth B centered about a carrier frequency. A transmitter, conventionally using a dedicated antenna emits a chirp signal towards the target(s), which reflect some of the signal and a delayed copy appears at the (one or multiple) receiver antenna(s). This radar signal contains significantly more energy as power is transmitted for the duration of the chirp. Higher performing waveforms in terms of resolution, range-doppler ambiguity, and interference robustness employ orthogonal codes in conjunction with phase modulation and correlation methods for radar pulse compression. Pulse compression is done in the digital domain and uses the distinct autocorrelation properties of the code sequences.

While the required special autocorrelation-property code sequences may be present in various communication physical layer standards, their use is governed by the need of the particular communications protocol and may not fit the radar purpose. For example, the codes may not be sent frequently enough, or not through a sufficient number of subcarriers (bandwidth) to yield the required ranging resolution and update rate.

Alternatively, the delay-time profile of the target reflection(s) is derived from a frequency domain measurement, similar to what is customary in a variant of Time-Domain-Reflectometry (TDR). In this type of TDR, the amplitude and phase response of reflections from a device-under-test (e.g. a damaged cable) are measured with a vector network analyzer (VNA) over a particular frequency range and the time domain reflection profile is computed via an inverse FFT.

Consider an ongoing OFDM transmission across a large number N of sub-carriers across a total bandwidth (B=NΔf). Preferably, the sub-carriers span the entire channel bandwidth (for best range resolution) without un-used gaps during a radar "frame" period. We assume that any relevant target-reflected signals arrive at the receiver with a delay spread no larger than the OFDM cyclic prefix duration. As a result, for every sub-carrier n (n=<NUM>. N-<NUM>) and symbol period k (t=k*TS), the amplitude & phase of the received signals <MAT> can be normalized to the amplitude & phase of the associated transmitted signals <MAT>. This effectively samples the frequency-domain channel response Hn = H(fn) at the occupied sub-carrier frequencies once per symbol period: <MAT> at the sub-carrier frequencies fn.

The receive samples need to be taken before the channel equalization step in the OFDM receiver. The sampled time domain response is then obtained as the inverse FFT: <MAT> at the time-instances tk =(k/N)TS.

With TS = <NUM>/Δf and B = NΔf, time samples of the channel response are spaced by Δt = (tk+<NUM> - tk) = <NUM>/B which leads to a range resolution ΔR = cΔt/<NUM> = c/<NUM>B, again. Here, c is the speed of light and B is a bandwidth.

The clear advantage of this approach to radar using OFDM modulated communication is that it makes no specific assumption about autocorrelation or any other properties of the signal, except that the subcarriers should be filled with valid (i.e. sufficient amplitude) signals. Since sub-carriers are dynamically assigned and may not all be populated simultaneously, the channel response can be averaged over symbol periods with non-zero sub-carrier powers, as long as sufficient measurement time (radar frame duration) is available. Hence, this radar approach can be taken in the context of <NUM> NR communications without interfering with communications in the cell.

In contrast to mono-static radars with co-located, synchronized transmitters and receivers, bi- or multi-static radars have two or more locations with receivers and transmitters.

<FIG> illustrates an exemplary bi-static radar mechanism <NUM> according to one embodiment of the disclosure. Without TX/RX synchronization, the receiver will derive its time reference from the direct-path (meaning a shortest delay) signal and refer all other, more delayed signals to targets. Assuming the distance between a TX-RX pair is known, in the absence of angle information, the time-of-arrival difference between the direct and reflected paths defines the target location(s) on the surface of an ellipsoid, i.e. the surface that is obtained by rotating an ellipse around its major axis, with the TX and RX at the foci. In that sense, a mono-static radar is a special case of a bi-static radar with the constant delay surface degenerated to a sphere.

<FIG> illustrates an exemplary mono-static radar mechanism <NUM> according to one embodiment of the disclosure. For large target distances, the equi-range ellipsoid degenerates into a sphere. If the TX-RX base distance is small enough, direct electrical synchronization similar to the monostatic case is practical and reliance of a direct path (which may be strongly attenuated or in an angular region of low antenna gain) is not needed. In this case, the arrangement is generally referred to a coherent multi-static radar.

It is evident that the mono-static round trip distance variable 2R will be replaced by the total propagation length or bistatic range (R<NUM> + R<NUM>). Resolution limitations apply to this total path length. TX and RX antenna apertures may be different, in general. In the radar equation, the <NUM>/R<NUM> distance dependence will be replaced by <MAT>, in analogy.

Also note that the reflected signal is no longer from a direct reflection and hence the observed radar cross section will differ from the mono-static case. For small to moderate bistatic angles (between TX and RX propagation directions), the RCS is approximately that of the bisector between TX and RX directions and lower than for the monostatic case. For large bistatic angles, RCS equals that of an equivalent shadow-area aperture with the associated roll-off angular pattern and a value approximately equal to the monostatic RCS. For angles approaching <NUM> deg. , the direct TX-RX path will likely dominate, possibly leading to a jammed RX without discernable signal. Hence, we expect that the bi-static radar produces substantially new information when the base-distance is comparable to the individual path lengths.

If TX and/or RX have angular elevation resolution in addition to time delay, the target location is narrowed down from the ellipsoid surface to the path obtained from the intersect of the ellipsoid with the plane spanned by the constant-elevation arc. If there is angular azimuth resolution also, then the target can be located as a point on the ellipse for full 3D resolution.

As discussed above, fine angular resolution requires large antenna aperture areas with correspondingly large number of antenna elements, both of which are not compatible with integration into handheld consumer devices. Methods to resolve the target location without reliance on Direction-of-Arrival (DoA) information are available.

In the Time-of-Arrival (ToA) method, there are several receivers and at least one transmitter at known locations. As will become apparent, this method is also referred to as "elliptic positioning". Receivers and transmitter are synchronized to a common time reference. For large physical distances, synchronization can be accomplished via the direct-path TX-RX signals, or, when the stations are close, time-synchronization can be done via electrical connection. To determine the target location unambiguously in an N-dimensional space, at least N independent range measurements are necessary.

In the main case of interest here, each RF module at location can be configured into either TX or RX mode. When n modules are available, a total of <MAT> unique pairs can be created.

With three available modules (#<NUM>, #<NUM>, #<NUM>), C<NUM>,<NUM> = <NUM> unique ranging measurements can be taken: (#<NUM>-#<NUM>), (#<NUM>-#<NUM>), (#<NUM>-#<NUM>). No redundancy is available in for a target outside the <NUM>-sensor plane.

With four available modules, C<NUM>,<NUM> = <NUM>. Three extra measurements are available. These can either be used to improve accuracy, or specific module-pairs can be chosen to maximize accuracy in the first place.

In the case where a particular location is either permanently configured as a TX-only or RX-only module, at least a total of N TX/RX combinations are necessary (e.g. (n RX)*(m TX) ≥ N). Here again, additional RXs and/or TXs can improve the target location accuracy result by averaging.

<FIG> illustrates an exemplary diagram showing how to determine possible target locations in mm-wave communication systems, using <NUM> Dimensional (3D) ellipsoids according to one embodiment of the disclosure. The embodiment shown in <FIG> is for illustration only.

The transmitter <NUM> sends out a signal that is reflected by the target and subsequently acquired by one of the receivers <NUM>. The signal path from the transmitter <NUM> via the target reflection to a receiver traces out an ellipsoidal surface for the possible target locations, with the TX and RX positions as the foci. Multiplication of the measured time delay with propagation velocity provides the bistatic range for the respective TX/RX pair. The intersection of the respective ellipsoids from several transmit-receive pairs yields the object location estimate.

Possible target locations are placed anywhere on the surface of a 3D ellipsoid <NUM>, generated by the rotation of an ellipse around its major axis. All have the same propagation path length between TX <NUM> and RX <NUM> at the foci of the ellipsoid <NUM>. For larger bistatic range values, the ellipsoid <NUM> expands as illustrated by ellipsoids <NUM> and <NUM>.

With many targets present, the elimination of false targets becomes challenging. Each target generates one ellipse for each TX - RX pair. If N=(n*m) TX/RX pairs are used (= # measurements taken) and K targets are present then we have (NK * (N-<NUM>)K)/<NUM> intersections, of which only K correspond to actual targets. For instance, in a modest case of N=<NUM>, K=<NUM>, we will have <NUM> total positions out of which only <NUM> correspond to the real targets. Therefore, this method is useful for a low number of targets, such as a single user or a user's hand in proximity to the device.

<FIG> illustrates an exemplary time-of-arrival bistatic radar ranging geometry for three configurable TX/RX modules according to one embodiment of the disclosure. The embodiment shown in <FIG> is for illustration only.

Minimally, three configurable TX/RX modules are needed to find the target's 3D position <NUM> at the intersection of the ellipses. Consider the geometry in <FIG>. The first two measurements are taken with module <NUM> as TX (TX1), modules <NUM> & <NUM> as RX (RX2, RX3). In the <NUM>rd measurement, module <NUM> can be disabled, module <NUM> is TX and module <NUM> is RX.

The measurements yield three bistatic ranges: r<NUM> = (rt<NUM> + rt<NUM>), r<NUM> = (rt<NUM> + rt<NUM>), r<NUM> = (rt<NUM> + rt<NUM>), which yields the individual module-to-target distances: rt<NUM> = (r<NUM> + r<NUM> - r<NUM>)/<NUM>, rt<NUM> = (r<NUM> - r<NUM> + r<NUM>)/<NUM>, rt<NUM> = (-r<NUM> + r<NUM> + r<NUM>)/<NUM>, and together with the module locations [xi, yi, zi], (i = <NUM>. <NUM>), the target location is found by solving: <MAT> for [xt, yt, zt].

<FIG> illustrates another exemplary time-of-arrival bistatic radar ranging geometry <NUM> for three configurable TX/RX modules according to one embodiment of the disclosure. The embodiment shown in <FIG> is for illustration only.

With more than three modules, N = Cn,<NUM> > <NUM> unique range measurements are possible. Consider the system geometry shown in <FIG>. The time-of-arrival bistatic radar ranging geometry <NUM> include one TX at the origin <NUM>.

The i-th receiver is located at known position xi = [xi, yi, zi]T, (i = <NUM>. N) and the target is located at xt = [xt, yt, zt]T. The bistatic range Rbi for the i-th receiver (the measured variable) is the sum of transmitter-target and target-receiver distances (Rt + Rti) :
<MAT>.

The unknowns in this equation are xt and Rt. For all ToAs (Rbi's) at the receivers, in matrix form:
<MAT> <MAT> <MAT>.

The approximate (in least-squares sense) solution for the target location xt is now obtained as:
<MAT> where: <MAT> is the pseudoinverse of Ab and:
<MAT>.

With that, the target position xt is known in least-squares approximation. It is intuitive, that the error variance decreases with higher accuracy range measurements Rbi (better signal/noise ratio), more favorable RX vs. TX placements (which lead to ellipsoids intersecting at larger angles), and favorable target distances (which are on the order of up to a few TX/RX spacings).

In case wave propagation for particular TX/RX pair(s) is obstructed, other pairs may be selected from the total available set of Cn,<NUM> combinations. <NUM>-D target location can be determined as long as at least three measurements are possible as shown below.

In the Time-Difference-of-Arrival (TDoA) method, multiple synchronized receiver stations at different and known spatial coordinates collect the signal emitted or reflected off a target. Here, the mutual differences between arrival times of the signals at pairs of receivers are measured.

The process of finding the target position is similar to the previous case, with the difference that time delays are measured relative to a reference receiver (we choose RX <NUM> as the reference, arbitrarily).

The i-th receiver is located at known position xi = [xi, yi, zi]T, (i = <NUM>. N) and the target is located at xt = [xt, yt, zt]T. The measured range-difference di,<NUM> between the i-th receiver and <NUM>st (reference-) receiver is di,<NUM> = Rti - Rt<NUM>. After rearranging the expression for <MAT> we get an expression similar to the ToA case:
<MAT>.

The unknowns in this equation are xt and Rt<NUM>. For all N-<NUM> TDoAs (di,<NUM>'s), in matrix form:
<MAT> <MAT> <MAT>.

The approximate (in least-squares sense) solution for the target location xt is now obtained as:
<MAT> where : <MAT> is the pseudoinverse of Ad and:
<MAT>.

Again, the target position xt is known in a least-squares approximation. Note that the TDoA method requires one more receiver than the ToA method (N+<NUM> receivers yield N TDoAs).

It is to be expected that the error variance in the TDoA method is larger than in the direct ToA-measurement method since the time-difference measurement may contain more measurement noise than the direct time measurement data. This is also shown in simulation in [<NUM>] and we remark that direct time measurement is preferable.

Mobile communications devices (e.g., smart phones) are starting to include mmWave-communications capability, either in the <NUM> mm-wave bands at <NUM>, <NUM>, or in the <NUM> ISM band. There are a few major aspects that govern the product design for this capability as follows.

Hence, the disclosed embodiments are to make use of the presence of a number of antennas and associated RF modules to improve on the ranging performance relative to what can be obtained with one module only, as is customarily done in the prior art.

<FIG> illustrates an exemplary block diagram of multi-RF ranging apparatus according to embodiments of the disclosure. The embodiment shown in <FIG> is for illustration only.

The ranging methods provided in the disclosure may be incorporated in mobile communications devices as shown in the block diagram of <FIG>. Variations and extensions may be readily apparent to those skilled in the art.

A mobile device with mm-wave communications ability contains a number of RF modules <NUM>-<NUM>, <NUM>-<NUM>, ···<NUM>-K. The modules may or may not be identical. Without loss of generality, we assume here that the modules communicate in the same RF frequency band or have the capability to operate in the respective other modules' frequency band(s). Each of the modules contains at least one antenna. Module level antenna arrays with more than one element are common and a preferred solution for communications. Antenna arrays allow beam forming and -steering, which is, however, not a requirement for the disclosed ranging system.

At least <NUM> antenna modules (K=<NUM>) modules may be required to determine the location of close-by reflecting targets in <NUM>-dimensional coordinates. The mm-wave enabled mobile devices may typically be equipped with four RF modules, which allows for one to be blocked (e.g. covered by the user's hand) while still retaining the ranging capability. The modules share a common time reference as indicated by the sync signal in the block diagram. For time-of-arrival (ToA) processing, the radar returns are measured relative to a common Sync signal which is generated in the radio baseband signal processing block as shown. For time-difference-of-arrival (TDoA) processing, RF modules may be required to be synchronized in a pair-wise fashion, which can be achieved without an explicit synchronization signal from the baseband unit. In any case, due to the physical close-ness within the mobile device, it will be advantageous to provide a common sync-to-baseband signal.

RF modules are assumed to contain at least all required mm-wave circuitry and can be configured alternately in exclusive transmit or exclusive receive modes. Simultaneous transmit and receive capability is not required for the disclosed ranging methods but can be used advantageously if available as a result of the communications capabilities. Challenges due to limited transmit/receive isolation within one module will need to be overcome.

The interface between the RF modules and the radio baseband processing device may be analog or digital, depending on the location of the receiver analog-to-digital converters (ADC's) and transmitter digital-to-analog converters (DAC's). In either case, the receive signal at this interface contains sufficient (amplitude, phase, timing, and the like) information to extract the necessary radar parameters (delay, amplitude, phase, frequency, Doppler shift, and the like) on a per-module basis. Moreover, the interface carries transmit signal information on a per-module basis, per the particular receive/transmit mode configuration of the entire set of modules.

The baseband signal processor block <NUM> customarily provides standards-based communications modem functions such as media access control (MAC), physical layer (PHY) interface functions, TX waveform generation based on required uplink data and RX demodulation to provide downlink data. In a communications-only device, the up- & downlink data paths terminate in an applications processor that may execute or manage all other functions of the mobile device, other than the modem functions.

The ranging system includes a radar processor block <NUM>. No constrains exist regarding the physical location of this block - it may be a stand-alone device or may be integrated either in the radio baseband processor or applications processor block (however, most likely it will be in the radio baseband), or consist of software functions only, re-utilizing existing hardware. The functions of the radar processor block <NUM> include configuring the RF modules in the sequence required for the ranging function, taking into account previously gathered information regarding blocked and available modules, generating appropriate "radar time intervals" as dedicated radar-only intervals, interleaved with communications operation, or re-using native communications transmit time slots, generating appropriate "radar waveforms" for transmission during radar intervals (these may be dedicated waveforms, or reuse native communications waveforms), extracting raw radar parameters from the receive signals, executing radar processing, and delivering time-stamped, reflection-strength and likelihood-tagged lists of identified targets at a desired frame-rate to the device's application processor.

In effect, this adds an additional sensor to the mobile device, whereby part or all of the sensor hardware is re-used from the available communications hardware.

The applications processor <NUM> combines the various communications, sensor, and user input data sources received from a mobile device <NUM> to form a useful device function. Radar-related applications such as the aforementioned presence & blocking detection, gesture recognition, bio-signal processing, and radar + other-sensor signal fusion will be executed on this processor as software functions. Advantages to the usability, desirability and marketability of the device may be derived from the inclusion of the radar ranging feature.

The disclosure utilizes transmitter-receiver pairs for ranging, whereby the respective transmitter and receiver are on physically separate RF modules, using separate antennas or antenna arrays.

The RF modules are mounted at known positions and thereby the positions of the antennas (or antenna arrays) phase centers can be known. Using disjoint transmitter (TX) and receiver (RX) locations for ranging is known as multi-static radar.

Starting with one reflecting or scattering target in the proximity, one TX-RX pair (e.g. TX1-RX2) can resolve the signal's time-of-flight along the TX - a target object(s) - RX path. This constrains the target location onto the surface of an ellipsoid where the phase centers of the TX, RX antenna (arrays) are at the foci of the ellipsoid. In other words, a first-TX/RX-pair measurement reduces the target location from a 3D unknown to a 2D unknown.

Also, slightly more information can be available when the antenna radiation patterns are also known. In that case, the target location is constrained to areas of the ellipsoid surface that are illuminated by the transmit antenna or that is in the field of view of the receive antenna.

In cases where only one TX-RX pair is available, this antenna-pattern-constrained area on the ellipsoid may be the only ranging information available. Since it is not possible to resolve angular information beyond that, a maximum-likelihood estimate of the target position may be chosen to be the center of the illuminated area on the ellipsoid. The directions of peak antenna pattern intensity are typically well known by lab characterization during the mobile device development and stored in the device's non-volatile memory (referred to a code book). Hence this information is readily available.

Also, the strength of the reflected signal for a single measurement is, in general, not a good indication of the target location as it is affected by distance, radar cross section, and angular position relative to the antennas' radiation patterns.

In most cases, however, the antenna beams may be steerable, e.g. via phase steering of the individual elements in the TX, RX, or both arrays. A refinement of the position information may be obtained by observing changes in the received signal strength while changing beam steering direction settings (sweeping through the code book). Since the sweep may be accomplished within a short time interval relative to changes in the target position, the code book setting that results in the largest reflected signal strength gives an improved indication of the angular target location, while the increased number of radial distance measurements can improve the target distance estimate.

<FIG> illustrates an exemplary diagram showing how to determine a possible target location, using two independent measurements according to embodiments of the disclosure. The embodiment shown in <FIG> is for illustration only.

A first independent measurement using a pair of TX1 and RX2 constrains the target onto the surface of a first ellipsoid <NUM>, and a second independent measurement using a pair of TX1 and RX3 constrains the target onto the surface of a second ellipsoid <NUM>. Here, the TX and RX antennas are installed in different locations within the housing of a UE to provide adequate spatial resolutions. Taken together with the first and the second ellipsoids <NUM> and <NUM>, the target object is determined be located on the intersection path <NUM> of the two ellipsoids <NUM> and <NUM>, i.e. on a one-dimensional path. Also, in this case where only two TX-RX pair are available, the antenna-pattern-constrained area on the <NUM>-D path may be the most ranging information available. Again, a maximum-likelihood estimate of the target position may be chosen to be the combined center of the illuminated area on the <NUM>-D path. A further refinement of the position information may be obtained also in this case by observing changes in the received signal strength while changing beam steering direction settings.

<FIG> illustrates another exemplary diagram showing how to determine a possible target location, using three independent measurements according to embodiments of the disclosure. The embodiment shown in <FIG> is for illustration only.

With the first independent measurement using a pair of TX1 and RX2 and a second independent measurement using a pair of TX1 and RX3, a third independent measurement yields a unique point for the target location. The target object is located on an intersection point <NUM> of the three ellipsoids <NUM>, <NUM> and <NUM>.

It is apparent from the schematic drawings that best accuracy will be achieved when the ellipsoids intersect at large angles in the vicinity of the target location. As the distance between the device and target increases, the ellipsoids will degenerate into spheres, intersect angles approach zero and intersect paths / points become very inaccurate. In that case, the method has to revert back to conventional radar operation, extracting radial distance from the radar response and angular (azimuth, elevation) information from the available beamforming capability of the modules. It may be possible to generate a set of (distance, azimuth, elevation) triplets from the individual radar modules to compute an improved target location estimate, particularly taking advantage of the differing antenna patters of the modules in different locations on the device.

<FIG> illstrates an exemplary <NUM>-D ranging operation with simultaneous measurements at <NUM> RX antennas in the mm-wave communication apparatus according to embodiments of the disclosure. The embodiment shown in <FIG> is for illustration only.

In the embodiments above, module #<NUM> was used as RX during the first set of measurements as illustrated in <FIG> and as TX during the latter set of measurements in association with <FIG>. This will generally be the preferred mode for generating a sufficiently large set of measurements. For example, four independent measurements can be done in this manner: TX1 - RX2, TX1 - RX3, TX4 - RX2, and TX4 - RX3, which is a sufficient amount to use the least-squares method for calculation. TX and RXs can be swapped on a per-measurement basis if advantageous, based on the environment around the device.

<FIG> illstrates an exemplary <NUM>-D ranging operation through simultaneous measurements with <NUM> RX antennas in the mm-wave communication apparatus according to embodiments of the disclosure. The embodiment shown in <FIG> is for illustration only.

The least-squares method also produces a solution in the presence of noise or otherwise perturbed measurement data at the expense of additional measurements. Note that rarely there will be only one target but rather a cluster of closely spaced, not (easily) resolvable targets, which contributes to measurement "noisiness".

<FIG> illustrate three measurements with three RF modules when the <NUM>-th RF module is covered by an obstruction according to one embodiment of the disclosure. The embodiments shown in <FIG> are for illustration only.

As a result of fundamental properties of electromagnetic signal reflections at dielectric boundaries such as the air-skin interface, the strength of the reflected signal will depend on the incident angle as well as the polarization of the signal. Generally, polarizing the signal such that the electric field vector is perpendicular to the plane of propagation will yield a stronger reflection. In case the signal is polarized such that the electric field vector is parallel to the plane of propagation, there exists a particular incident angle that exhibits no reflection at all (known as the Brewster angle).

Since the position and orientation of the target object (and therefore the plane of propagation) at the time of measurement are not known, it may be advantageous to perform measurements at various combinations of TX and RX polarizations to select the strongest return, average the returns based for improved accuracy, and the like.

Each TX antenna and each RX antenna are configured to form one of four pairs comprising a TX antenna transmitting the first polarized signal and an RX antenna receiving the first polarized signal, a TX antenna transmitting the first polarized signal and an RX antenna receiving the second polarized signal, a TX antenna transmitting the second polarized signal and an RX antenna receiving the first polarized signal, or a TX antenna transmitting the second polarized signal and an RX antenna receiving the second polarized signal.

For each TX-RX pair, four possibilities exist (TX for H and RX for H, TX for H and RX for V, TX for V and RX for H, or TX for V and RX for V), where H indicates a horizontal polarization, and V indicate a vertical polarization.

<FIG> illustrates an exemplary proximity detection for safety or user-convenience according to embodiment of the disclosure. The embodiment shown in <FIG> is for illustration only.

In one embodiment, the multi-communications module radar functionality is used to simply detect the presence of close-by reflecting at least one object. There are several motivations for this proximity detection.

First motivation is safety. Most of the energy in mm-wave beams when targeted at biological tissue is absorbed in the skin layer, causing local heating. Measurements have shown an approximate <NUM> degree C relative skin-surface temperature increase per <NUM> mW/cm<NUM> power flux @ <NUM>. This is also supported by simulations with established skin-layer models. Typical clothing does not provide a significant amount of shielding. Awareness in the device regarding the physical close-ness of a lively object (for example the user) allows the device to adapt its mm-wave transmit power in accordance with regulatory limits, also for the ongoing communications use case. This also prevents the device to employ the user's body as a reflector when determining a mm-wave signal path between itself and a communications base station (eNB).

Second motivation is the convenience functions of the mobile device, such that e.g. certain functions are activated when a user gets close, the device can enter a low power sleep mode when no user is present, or the device enters an alarm state when no user is present in the vicinity, e.g., mobile phone with separation anxiety, or the device may pause the playback of a movie when the user distance increases beyond a configurable distance, and the like.

In the ranging applications, if the radar detects a large range of targets over a wide angular spread, and the target distance is not stable over time (as would be the case in a static/nonlively environment), then an object is detected as live and proximate.

<FIG> illustrate an exemplary bio-signal measurement and resultant graphs for heart and respiration rates according to embodiments of the disclosure. The embodiments shown in <FIG> are for illustration only.

Human respiration rate is in the range <NUM> to <NUM> with a chest deflection of <NUM> to <NUM>, typically. Human heart rate is in the range <NUM> to <NUM> with <NUM> to <NUM> chest deflection. Since the frequency ranges are uniquely different, measurement of heart and respiration rates are possible in humans via radar ranging measurements. Range resolution well below one wavelength may be required even at <NUM>, which is achieved by tracking the phase of the target responses. A chest deflection of <NUM> causes a <NUM>-degree phase change for a <NUM> reflected signal and is well detectable.

The radar methods described in the disclosure can equally be used for this type of detection. The major challenge with this application is the separation of whole-body movements from the heart beat and respiration signal, the latter being much smaller in displacement. With the measurement device being handheld, an additional challenge lies in compensating for its movements. This can, in principle, be done via the device's built-in acceleration sensors.

<FIG> illustrates exemplary ranging update at frames in radar slow-time generating raw-data for gesture recognition according to embodiments of the disclosure. The embodiments shown in <FIG> is for illustration only.

One of main purposes of this embodiment is to simplify and enhance the user interaction with the mobile device. As such, determining the location of a close-by object such as a hand, is the technological basis upon which higher level applications are built.

An underlying assumption for the ranging operation is that it is "fast" with respect to any relative location changes between the device and the target object(s). Since the radar signal's round-trip distance is very small compared to typical wireless communication distances, we expect large receiver signal-to-noise ratios during radar operation, requiring little averaging and as a result, allowing for short measurement periods. Since re-configuration of the modules between TX and RX modes can be done within micro-seconds, a full set of ranging measurements across all the required TX/RX pairs requires on the order of <NUM> to no more than <NUM> milli-second (ms). Hence, we consider the geometric arrangement and environment as static (frozen) for this time. This is often referred to as taking measurements in fast time.

Object tracking requires measurements in slow time, i.e., repeated location measurements within periodic or non-periodic time periods that sample the target object's trajectory. Slow time measurements may be done at a rate of <NUM>. <NUM> per second, yielding sufficient accuracy and low resource usage / power consumption in the radar subsystem (~ <NUM>% duty cycle for <NUM> RF on-time at <NUM> fps update rate). Schematically, this operation can be seen in the sequence below. Over the course of <NUM> ranging operations at frames n, n+<NUM>, and n+<NUM>, a target trajectory segment is established. Note that smoothing may be required before further processing to suppress the effects of noise.

In a subsequent step, the time-stamped ranging measurements may be used as inputs to a classification algorithm to associate known trajectories (gestures) with the observed coordinate list. For a single dominant target, the ranging output may be a formatted list as shown below: <MAT>.

Here, Timestamp(n) marks the time at which the n-th measurement was taken, typically in units of "system ticks", x(n), y(n), z(n) are the observed target coordinates, RCS(n) is the apparent radar cross section, and Prob(n) is the likelihood of a true detection.

In case several significant targets are detected, each time stamp may have several entries for location, RCS and detection probability.

Prob(n) is derived from the power in the radar reflection relative to a noise-floor estimate during raw radar signal processing. The closer the signal to the noise floor, the (exponentially) higher the likelihood for a false detection. Hence, when Prob(n) falls below a threshold (e.g., <NUM>), the data point may be discarded prior to classification.

<FIG> illustrates an exemplary flowchart for ranging operations in the mm-wave communication apparatus according to embodiments of the disclosure. The embodiment of the method <NUM> shown in <FIG> is for illustration only. One or more of the components illustrated in <FIG> can be implemented in specialized processing circuitry configured to perform the noted functions or one or more of the components can be implemented by one or more processors executing instructions to perform the noted functions.

The mm-wave communication device includes a plurality of radio frequency (RF) antennas configured to transmit (TX) or receive (RX) mm-wave RF signal, and a processor coupled to the plurality of RF antennas. The device also includes a housing with a display, wherein the plurality of RF antennas is installed on different sides of the housing. The device further includes a sensor to determine an orientation of the apparatus. The device can include at least four RF antennas.

In operation <NUM>, the device configures at least one RF antenna, among the plurality of RF antennas, as a TX antenna and at least one of the remaining RF antennas as RX antenna(s).

In one embodiment, the processor determines an orientation of the device, and select one of the plurality of RF antennas as the TX antenna based on the orientation of the device. For example, the device is oriented in a portrait mode, the RF antenna on the top-short side can be configured as a TX antenna, and RF antennas on the other sides, such as the top-short side, and the top and bottom long sides, can be configured as at least one RX antenna. Alternatively, the device is oriented in a landscape mode, the RF antenna on the top-long side can be configured as a TX antenna, and on the other sides, such as the bottom-long side, and the top and bottom short sides, can be configured as at least one RX antenna.

In operation <NUM>, the TX antenna transmits the at least one RF signal, and at least one RX antenna receive portions of the at least one RF signal, wherein the portions are reflected from at least one object. The at least one RF signal can be an orthogonal frequency division multiplexing (OFDM) communication signal.

In operation <NUM>, the processor determines whether the number of TX-RX pairs is sufficient for calculating a position of a target object. For example, three TX-RX pairs will be sufficient for determining a <NUM>-D target location, when using the least-squares method. If the number of TX-RX pairs is not sufficient for calculating a position of a target object, the processor proceeds to operation <NUM>; otherwise, proceeds to operation <NUM>.

In operation <NUM>, the processor selects a different RF antenna as a new TX antenna, among the at least one RX antenna and then proceeds to operation <NUM> to reconfigure the plurality of RF antennas.

In operation <NUM>, the device calculates each of flight times of the at least one RF signal with respect to each of the at least one RX antenna. The processor generates a plurality of ellipsoids, each ellipsoid with a first focus at the TX antenna and a second focus at each of the at least one RX antenna.

Claim 1:
A mobile device (<NUM>) comprising:
a plurality of radio frequency, RF, antennas (<NUM>) configured to transmit, TX, or receive, RX, at least one RF signal; and
a processor (<NUM>) configured to:
configure at least one RF antenna, among the plurality of RF antennas (<NUM>), as a TX antenna and at least one of the remaining RF antennas as at least one RX antenna based on an orientation of the mobile device (<NUM>),
cause the TX antenna to transmit the at least one RF signal,
cause the at least one RX antenna to receive portions of the at least one RF signal, the portions being reflected from at least one object.
calculate each of flight times of the at least one RF signal with respect to each of the at least one RX antenna, and
identify a location of the at least one object based on intersections of each of a plurality of ellipsoids with a first focus at the TX antenna and a second focus at each of the at least one RX antenna,
wherein the plurality of ellipsoids is determined based on each of the flight times of the at least one RF signal and each of the plurality of RF antennas is reconfigurable as the TX antenna or the at least one RX antenna.