PATENT DOCUMENT

Abstract:
Various embodiments of an analog-to-digital (A/D) device are described herein, with the A/D device using at least a comparison unit, a comparative operation control circuit, a delay circuit, and a successive operation control circuit arranged so as, it may, among other things, mitigate conversion errors that may be due to differences in properties of circuit elements therein. And the A/D device may be implemented in, among other things, a signal processing device.

Full Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is based upon and claims the benefit of priority from the prior Japanese Patent Application NO. 2009-074286 filed on Mar. 25, 2009, and the prior Japanese Patent Application NO. 2009-149496 filed on Jun. 24, 2009, the entire contents of which are incorporated herein by reference. 
     FIELD 
     The embodiments discussed herein are related to a comparison circuit, an analog-to-digital conversion device using the comparison circuit thereof, and a signal processing device using the analog-to-digital conversion device thereof. 
     BACKGROUND 
     Voltage-comparative-type comparators frequently used for a control circuit and the like generally include two MOS transistors of which the gates receive one of differential input signals, two current routes of which the currents are controlled by the MOS transistors thereof according to the voltage of the differential input signal, and a latch unit configured to amplify and hold potential difference between the current routes. 
     Accordingly, in the event of executing comparison between the voltages of the differential input signals with the above comparator according to the difference of properties of the above MOS transistors, or the amplification and holding property of the latch unit, error occurs. As a result thereof, conversion error occurs in an analog-to-digital converter configured of this voltage-comparative-type comparator. 
     Related art is discussed in P. M. Figueiredo, P. Cardoso, A. Lopes, C. Fachada, N. Hamanishi, K. Tanabe, and J. Vital, “A 90 nm CMOS 1.2V 6 b 1 GS/s Two-Step Subranging ADC,”  IEEE International Solid - State Circuits Conference , Session 31/31.2, February 2006, and J. Craninckx and G. Van der Plas, “A 65 fJ/Conversion-Step 0-to-50 MS/s 0-to-0.7 mW 9 b Charge-Sharing SAR ADC in 90 nm Digital CMOS,”  IEEE International Solid - State Circuits Conference , Session 13/13.5, vol. XL, pp. 246-247, 600, February 2007. 
     SUMMARY 
     According to one aspect of the embodiments includes: an input circuit made up of a first transistor configured to receive a first signal at a gate electrode, and a second transistor configured to receive a second signal at a gate electrode; a first current route of which the current is controlled by the first transistor according to the voltage of the first signal; a second current route of which the current is controlled by the second transistor according to the voltage of the second signal; a latch circuit configured to amplify potential difference between a first node within the first current route and a second node within the second current route; a comparative operation control circuit including a first switch configured to execute supply of high potential or supply of ground potential to the drain of the first transistor, or supply of high potential or blocking of supply of ground potential to the drain, a second switch configured to execute supply of high potential or supply of ground potential to the drain of the second transistor, or supply of high potential or blocking of supply of ground potential to the drain, and a third switch configured to execute supply or blocking of supply of ground potential to the first current route and the second current route; and a comparative operation setting circuit configured to independently control the supply or blocking of supply of the first switch, the second switch, and the third switch. 
     The object and advantages of the embodiments will be realized and attained by means of the elements and combinations particularly pointed out in the claims. 
     It is to be understood that both the foregoing general description and the following detailed description and are exemplary and explanatory and are not restrictive of the embodiments, as claimed. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG. 1  is a circuit diagram illustrating a comparison circuit according to a first embodiment. 
         FIGS. 2A ,  2 B,  2 C, and  2 D are circuit diagrams illustrating a delay circuit and a logic circuit. 
         FIG. 3  illustrates a flowchart relating to the operation of the logic circuit for controlling the delay circuit. 
         FIG. 4  illustrates a timing chart for describing the operation of the lock circuit. 
         FIGS. 5A and 5B  are diagrams representing change in the signal potentials of signals when delaying the leading-edge point-of-time of a signal. 
         FIG. 6  is a circuit diagram illustrating a comparison circuit according to a second embodiment. 
         FIG. 7  is a circuit diagram illustrating a comparison circuit according to a third embodiment. 
         FIG. 8  is a circuit diagram illustrating a comparison circuit according to a fourth embodiment. 
         FIG. 9  is a circuit diagram illustrating a comparison circuit according to a fifth embodiment. 
         FIG. 10  illustrates an ADC (Analog Digital Converter) according to a sixth embodiment. 
         FIGS. 11A and 11B  illustrate examples of inverters of a delay circuit. 
         FIGS. 12A and 12B  are diagrams for describing operation according to the sixth embodiment regarding a latch unit, an input unit, a comparative operation control circuit, and the delay circuit. 
         FIG. 13  is a diagram representing relationship of the logic of a signal, trailing time difference of signals, and potential difference between signals making up an input signal. 
         FIG. 14  illustrates operation waves when detecting potential difference of complementary signals made up of signals according to the ADC circuit according to the sixth embodiment. 
         FIG. 15  is a flowchart for describing the control of a calculation executed by a successive comparative operation control circuit, and a detection method of difference input potentials executed by the control thereof. 
         FIGS. 16A and 16B  are tables for describing a method for deriving relationship between a binary numeral represented by a signal, and a binary numeral represented by a digital signal to be output by analog-to-digital conversion by the ADC in the case that there is no linearity with correlation as to difference between the potentials of signals. 
         FIG. 17  illustrates an ADC according to a seventh embodiment. 
         FIG. 18  illustrates an ADC circuit according to an eighth embodiment. 
         FIG. 19  illustrates an ADC circuit according to a ninth embodiment. 
         FIG. 20  is a diagram illustrating an ADC according to a tenth embodiment. 
         FIG. 21  is a diagram illustrating a reception device using the ADCs illustrated by the sixth through tenth embodiments. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
     The present invention encompasses modifications, obtained by adding design modifications to the embodiments described below, which one skilled in the art can conceive, and modifications obtained by executing recombination of components in the embodiments. Also, the present invention also encompasses modifications wherein the components are replaced with other components which yield the same operations, effects, or the like, and the present invention is not restricted to the following embodiments. 
     First Embodiment 
       FIG. 1  is a circuit diagram illustrating a comparison circuit  10  according to the first embodiment. The comparison circuit  10  includes a latch unit  20 , an input unit  30 , a comparative operation control circuit  40 , and a comparative operation setting circuit  50 . 
     The comparative operation setting circuit  40  includes P-type MOS transistors  42  and  43 , and an N-type MOS transistor  41 . 
     The N-type MOS transistor  41  has a drain to be connected to the sources of N-type MOS transistors  31  and  32  of the input unit  30 , a source to be connected to ground VSS  70 , and a gate for receiving a signal L. The N-type MOS transistor  41  supplies the ground potential from the ground VSS  70  to the input unit  30  when the logic of the signal L is “H”, and blocks supply of ground potential form the ground VSS  70  to the input unit  30  when the logic of the signal L is “L”. The N-type MOS transistor  41  serves as a switch for connecting or blocking the input unit  30  and the ground VSS  70 . 
     The P-type MOS transistor  42  has a source to be connected to a high-potential VDD power source  60 , a drain to be connected to the source of an N-type MOS transistor  23  of the latch unit, and a gate for receiving a signal LM. 
     The P-type MOS transistor  42  blocks supply of high potential VDD from the high-potential VDD power source  60  to the latch unit  20  and the input unit  30  when the logic of the signal LM is “H”, and supplies the high-potential VDD from the high-potential VDD power source  60  to the latch unit  20  and the input unit  30  when the logic of the signal LM is “L”. The P-type MOS transistor  42  serves as a switch for connecting or blocking the latch unit  20  and the input unit  30 , and the high-potential VDD power source  60 . 
     The P-type MOS transistor  43  has a source to be connected to the high-potential VDD power source  60 , a drain to be connected to the source of an N-type MOS transistor  24  of the latch unit, and a gate for receiving a signal LP. 
     The P-type MOS transistor  43  blocks supply of high potential VDD from the high-potential VDD power source  60  to the latch unit  20  and the input unit  30  when the logic of the signal LP is “H”, and supplies the high-potential VDD from the high-potential VDD power source  60  to the latch unit  20  and the input unit  30  when the logic of the signal LM is “L”. The P-type MOS transistor  43  serves as a switch for connecting or blocking the latch unit  20  and the input unit  30 , and the high-potential VDD power source  60 . 
     The input unit  30  includes N-type MOS transistors  31  and  32 . The N-type MOS transistor  31  has a drain to be connected to the source of the N-type MOS transistor  23  of the latch unit, a source to be connected to the drain of the N-type MOS transistor  41  of the comparative operation control circuit  40 , and a gate for receiving an input signal VIP. The on resistance value of the N-type MOS transistor  31  varies according to the potential of the input signal VIP. The N-type MOS transistor  32  has a drain to be connected to the source of an N-type MOS transistor  24  of the latch unit, a source to be connected to the drain of the N-type MOS transistor  41  of the comparative operation control circuit  40 , and a gate for receiving a signal VIM. The on resistance value of the N-type MOS transistor  32  varies according to the potential of the signal VIM. 
     As a result thereof, signals obtained by inverting the logics of the input signals VIP and VIM occur on the drains of the N-type MOS transistors  31  and  32 , respectively. 
     The latch unit  20  includes P-type MOS transistors  21  and  22 , and the N-type MOS transistors  23  and  24 . 
     The P-type MOS transistor  21  has a drain to be connected to the drain of the N-type MOS transistor  23 , a gate to be connected to the drain of the P-type MOS transistor  22 , and a source to be connected to the high-potential VDD power source  60 . 
     The P-type MOS transistor  22  has a drain to be connected to the drain of the N-type MOS transistor  24 , a gate to be connected to the drain of the P-type MOS transistor  23 , and a source to be connected to the high-potential VDD power source  60 . 
     The N-type MOS transistor  23  has a drain to be connected to the drain of the P-type MOS transistor  21 , a gate to be connected to the drain of the N-type MOS transistor  24 , and a source to be connected to the drain of the N-type MOS transistor  31  of the input unit  30 . As a result thereof, a signal obtained by inverting the logic of the input signal VIP appears on the drain of the N-type MOS transistor  23 . 
     The N-type MOS transistor  24  has a drain to be connected to the drain of the P-type MOS transistor  22 , a gate to be connected to the drain of the N-type MOS transistor  23 , and a source to be connected to the drain of the N-type MOS transistor  32  of the input unit  30 . As a result thereof, a signal obtained by inverting the logic of the input signal VIM appears on the drain of the N-type MOS transistor  24 . 
     Output signals OM and OP are output from the latch unit  20 . The output signal OM is connected to a node A between the drain of the P-type MOS transistor  21  and the drain of the N-type MOS transistor  23 . The output signal OP is connected to a node B between the drain of the P-type MOS transistor  22  and the drain of the N-type MOS transistor  24 . 
     The gate of the P-type MOS transistor  21  and the drain of the P-type MOS transistor  22  of the latch unit  20  are connected to the node B, and the gate of the P-type MOS transistor  22  and the drain of the P-type MOS transistor  21  are connected to the node A. That is to say, the P-type MOS transistor  21  and the P-type MOS transistor  22  are connected crosswise to the nodes A and B, and accordingly, the P-type MOS transistor  21  and the P-type MOS transistor  22  amplify potential difference between the node A and node B. 
     The gate of the N-type MOS transistor  23  and the drain of the N-type MOS transistor  24  of the latch unit  20  are connected to the node B, and the gate of the N-type MOS transistor  24  and the drain of the N-type MOS transistor  23  are connected to the node A. That is to say, the N-type MOS transistor  23  and the N-type MOS transistor  24  are connected crosswise to the nodes A and B, and accordingly, the N-type MOS transistor  23  and the N-type MOS transistor  24  amplify potential difference between the node A and node B. 
     Thus, the P-type MOS transistor  21  and N-type MOS transistor  23  of the latch unit  20 , and N-type MOS transistor  31  are connected serially between the high-potential VDD power source  60  and the drain of the N-type MOS transistor  41 , and make up a first current route including the node A. The P-type MOS transistor  22  and N-type MOS transistor  24  of the latch unit  20 , and N-type MOS transistor  32  are connected serially between the high-potential VDD power source  60  and the drain of the N-type MOS transistor  41 , and make up a second current route including the node B. 
     Therefore, when the signals L, LM, and LP are “H”, supply of the high-potential VDD to the latch unit  20  and the input unit  30  is blocked by the P-type MOS transistors  42  and  43 , and ground potential is supplied to the input unit  30  by the N-type MOD transistor  41 . In the above case, the potentials of the nodes A and B start to decrease from the high-potential VDD. The on resistance of the N-type MOS transistor  31  varies according to the potential of the input signal VIP, and the on resistance of the N-type MOS transistor  32  varies according to the potential of the input signal VIM, and accordingly, the current amount flowing to each current route varies. Thus, the rates of decrease at the nodes A and B vary. Of the potentials of the nodes A and B, one first reaching the threshold of the latch unit  20  becomes “L”. If we say that the potential of the node A first reaches the threshold of the latch unit  20 , the P-type MOS transistor  22  is turned on, and the potential of the node B increases, and the logic thereof becomes “H”. Conversely, the P-type MOS transistor  21  is turned off, and the potential of the node A decreases, and the logic thereof becomes “L”. 
     Note that, when the signals L, LM, and LP are “L”, the high-potential VDD is supplied to the latch unit  20  and the input unit  30  by the P-type MOS transistors  42  and  43 , and supply of the ground potential to the input unit  20  is blocked by the N-type MOS transistor  41 . 
     As a result thereof, the potential difference between the nodes A and B is 0, or almost eliminated. 
     The comparative operation setting circuit  50  includes a delay circuit  51 , and a logic circuit  56  for controlling the delay circuit  51 . 
       FIGS. 2A ,  2 B,  2 C, and  2 D are circuit diagrams illustrating the delay circuit  51  and the logic circuit  56 . 
       FIG. 2A  is a circuit diagram illustrating the delay circuit  51 . The delay circuit  51  includes a delay circuit  52 , a delay circuit  53 , and AND circuits  54  and  55 . 
     The delay circuit  52  receives the signal L, and outputs a signal LPA illustrating change in potential wherein the delay corresponding to a binary numeral DCP represented by digital signals DCP&lt; 0 &gt; through DCP&lt; 2 &gt; is added to change in the potential of the signal L. Note that the binary numeral DCP represented by the digital signals DCP&lt; 0 &gt; through DCP&lt; 2 &gt; advances, such as shown later, from (111) to (000), and when changing from the digital signal DCM&lt; 0 &gt; to the digital signal DCM&lt; 2 &gt;, maintains (000). 
     The delay circuit  53  receives the signal L, and outputs a signal LMA illustrating change in potential wherein the delay corresponding to the binary numeral DCM represented by the digital signals DCM&lt; 0 &gt; through DCM&lt; 2 &gt; is added to change in the potential of the signal L. Note that the binary numeral DCM represented by the digital signals DCM&lt; 0 &gt; through DCM&lt; 2 &gt; advances, such as shown later, from (000) to (111), and when changing from the digital signal DCM&lt; 0 &gt; to the digital signal DCM&lt; 2 &gt;, maintains (111). 
     The AND circuit  54  outputs a signal LP having the logical AND between the logic of a signal LPA and the logic of the signal L. The AND circuit  55  outputs a signal LM having the logical AND between the logic of a signal LMA and the logic of the signal L. Accordingly, the signal LP is changed from the logic “L” to logic “H” generally at the same time as the signal LPA being changed from the logic “L” to logic “H”. On the other hand, the signal LP is changed from the logic “H” to logic “L” generally at the same time as the signal L being changed from the logic “H” to logic “L”. The signal LM is changed from the logic “L” to logic “H” generally at the same time as the signal LMA being changed from the logic “L” to logic “H”. On the other hand, the signal LM is changed from the logic “H” to logic “L” generally at the same time as the signal L being changed from the logic “H” to logic “L”. 
     Thus, upon a switch made up of the N-type MOS transistor  41  being turned on by the signal L, a switch made up of the P-type MOS transistor  42  is turned on with delay according to the binary numeral DCM represented by the digital signals DCM&lt; 0 &gt; through DCM&lt; 2 &gt;, a switch made up of the P-type MOS transistor  43  is turned on with delay according to the binary numeral DCP represented by the digital signals DCP&lt; 0 &gt; through DCP&lt; 2 &gt;. 
       FIG. 2B  is a circuit diagram illustrating a first example  58  of the delay circuits  52  and  53 . The first example  58  of the delay circuits  52  and  53  includes a P-type MOS transistor  580  and an N-type MOS transistor  581  which make up an inverter  58 A, a P-type MOS transistor  588  and an N-type MOS transistor  589  which make up an inverter  58 B, inverters  585 ,  586 , and  587 , and capacitances  582 ,  583 , and  584  made up of a gate electrode-insulating film-substrate electrode-type transistor, i.e., a so-called MOS-type transistor. The inverter  58 A outputs a signal obtained by inverting the logic of the signal L to the inverter  58 B. The inverter  58 B receives the inverted signal of the signal L, and outputs the signal LPA having further inverted logic thereof (the signal LPA at the time of the first example  58  corresponding to the delay circuit  52 , and the signal LMA when the first example corresponding to the delay circuit  53 ). The capacitances  582 ,  583 , and  584  are connected to a signal line which connects the inverter  58 A and the inverter  58 B by the gate electrode. When the potential of the substrate electrode is high, and the threshold voltage of the MOS-type transistor is equal to or greater than the ground voltage, the capacitances  582 ,  583 , and  584  have great capacitance according to the thickness of the insulating film, but when the potential of the substrate electrode is low, and the threshold voltage of the MOS-type transistor is less than the ground voltage, have small capacitance. In the event that the electrostatic capacitances of the capacitances  582 ,  583 , and  584  are compared with those at the time of having the same threshold voltage, if we say that the capacitance  582  is 1, the capacitance  583  is the ratio of 2, and the capacitance  584  is the ratio of 4. The outputs of the inverters  585 ,  586 , and  587  are connected to the substrate electrodes of the capacitances  582 ,  583 , and  584 , respectively. The inputs of the inverters  585 ,  586 , and  587  are connected to the digital DCP&lt; 0 &gt; through DCP&lt; 2 &gt;, respectively (note that, when the first example  58  corresponds to the delay circuit  53 , these are connected to the DCM&lt; 0 &gt; through DCM&lt; 2 &gt;). Accordingly, the signal from the inverter  58 A to the inverter  58 B is delayed according to the binary numeral represented by the digital signals DCP&lt; 0 &gt; through DCP&lt; 2 &gt;. 
       FIG. 2C  is a circuit diagram illustrating a second example  59  of the delay circuits  52  and  53 . The second example  59  of the delay circuits  52  and  53  includes a P-type MOS transistor  590  and an N-type MOS transistor  591  which make up an inverter  59 A, a P-type MOS transistor  592  and an N-type MOS transistor  593  which make up an inverter  59 B, a variable resistor  594  connected to the P-type MOS transistor  590  and the high-potential VDD power source  60  and serially connected to the P-type MOS transistor  590 , and a variable resistor  595  connected to the N-type MOS transistor  591  and the ground power source and serially connected to the N-type MOS transistor  591 . The inverter  59 A receives the signal L, and outputs an inverted signal thereof to the inverter  59 B. The inverter  59 B further inverts the signal from the inverter  59 A to take this as the signal LPA (the signal LMA when the second example  59  represents the delay circuit  53 ). With the variable resistors  594  and  595 , the resistance varies according to the binary numeral DCP represented by the digital signals DCP&lt; 0 &gt; through DCP&lt; 2 &gt;, and become the maximum resistance when the binary numeral DCP is (111) (when the second example  59  represents the delay circuit  53 , the resistance varies according to the binary numeral DCM represented by the digital signals DCM&lt; 0 &gt; through DCM&lt; 2 &gt;). Accordingly, the potential supplied to the inverter  59 A increases or decreases according to the variable resistors  594  and  595 , and accordingly, the signal to be output from the inverter  59 A is output with delay according the potential thereof from the input signal. 
       FIG. 2D  is a circuit diagram illustrating a specific example of the logic circuit  56  for controlling the delay circuit  51 . The logic circuit  56  includes a DFF  560 , DFF  570 , JKFF  561 , JKFF  562 , JKFF  563 , JKFF  564 , JKFF  571 , JKFF  572 , JKFF  573 , JKFF  574 , AND  565 ,  566 ,  567 ,  568 ,  569 ,  575 ,  576 ,  577 , and  579 . 
     Note that JKFF means a JK flip-flop. Specifically, when J=“H” and K=“H”, signals to be output from terminals Q and /Q are logically inverted each time clock enters a clock terminal CK. Also, when J=“L” and K=“L”, the logics of the signals to be output from the terminals Q and /Q are unchanged. Further, when J=“L” and K=“H”, the terminal Q outputs “H”, and the terminal /Q outputs “L”, and when J=“H” and K=“L”, the terminal Q outputs “L”, and the terminal /Q outputs “H”. Note that according to the definitions of the terminals /Q and Q, the logics of the output signals of the terminals Q and /Q when J=“L” and K=“H”, and the logics of the output signals of the terminals Q and /Q when J=“H” and K=“L” counterchange. In this case, let us say that the signals DCP&lt; 0 &gt; through DCP&lt; 2 &gt;, and DCM&lt; 0 &gt; through DCM&lt; 2 &gt; counterchange. 
     The logic circuit  56  receives signals CS, CD, CK, and OP, and outputs the signals DCP&lt; 0 &gt; through DCP&lt; 2 &gt;, and DCM&lt; 0 &gt; through DCM&lt; 2 &gt;. 
     The AND  568  receives input of the signals CS and OP, and outputs a signal having a logic obtained by logical AND of both signals to terminals D of the DFF  560  and  570 . Accordingly, when the signal CS rises from logic “L” to logic “H”, the signal OP has originally logic “H”, the AND  568  outputs the signal of logic “H”. Subsequently, when the signal OP becomes logic “L”, the AND  568  outputs the signal of logic “L”. 
     The DFF  560  receives a clock signal CK from the terminal CK, latches the logic of the input signal from the terminal D at the leading edge of the clock signal CK, and outputs the latched signal from the terminal Q. 
     The AND  569  receives the signal from the terminal Q of the DFF  560 , and the signal /CE, and outputs a signal having a logic obtained by logical AND of both signals to the JKFF  561 , JKFF  562 , JKFF  563 , and JKFF  564 . Here, the signal /CE is a signal having the inverted logic of the signal CE. Accordingly, when the signal from the terminal Q of the DFF  560  is logic “H”, the output signal of the AND  569  has the same logic as the logic of the signal /CE. On the other hand, when the signal from the terminal Q of the DFF  560  is logic “L”, the signal of logic “L” is output to terminals k of the JKFF  561 , JKFF  562 , JKFF  563 , and JKFF  564 . 
     The JKFF  561  receives the latched signal from the terminal Q of the DFF  560  by a terminal J, receives the output of the AND  569  by the terminal k, and receives the clock signal CK by the terminal CK. The JKFF  562  receives the latched signal from the terminal Q of the DFF  560  by the terminal J, receives the output of the AND  569  by the terminal k, and receives the output signal from the terminal Q of the JKFF  561  by the terminal CK. The JKFF  563  receives the latched signal from the terminal Q of the DFF  560  by the terminal J, receives the output of the AND  569  by the terminal k, and receives the output signal from the terminal Q of the JKFF  562  by the terminal CK. The JKFF  564  receives the latched signal from the terminal Q of the DFF  560  by the terminal J, receives the output of the AND  569  by the terminal k, and receives the output signal from the terminal Q of the JKFF  563  by the terminal CK. The AND  565  receives the signal from the terminal Q of the JKFF  561 , and the signal from the terminal Q of the JKFF  564 , and output a signal obtained by logical AND thereof as the digital signal DCP&lt; 0 &gt;. The AND  566  receives the signal from the terminal Q of the JKFF  562 , and the signal from the terminal Q of the JKFF  564 , and output a signal obtained by logical AND thereof as the digital signal DCP&lt; 1 &gt;. The AND  567  receives the signal from the terminal Q of the JKFF  563 , and the signal from the terminal Q of the JKFF  564 , and output a signal obtained by logical AND thereof as the digital signal DCP&lt; 2 &gt;. 
     The DFF  570  receives the clock signal CK from the terminal CK, latches the logic of the input signal from the terminal D at the leading edge of the clock signal CK, and outputs the latched signal from the terminal Q. 
     The AND  579  receives the signal from the terminal Q of the DFF  570 , and the signal /CE, and outputs a signal having a logic obtained by logical AND of both signals to the JKFF  571 , JKFF  572 , JKFF  573 , and JKFF  574 . Accordingly, when the signal from the terminal Q of the DFF  570  is logic “H”, the output signal of the AND  579  has the same logic value as the logic of the signal /CE. On the other hand, when the signal from the terminal Q of the DFF  570  is logic “H”, the signal of logic “L” is output to the terminals k of the JKFF  571 , JKFF  572 , JKFF  573 , and JKFF  574 . 
     The JKFF  571  receives the latched signal from the terminal Q of the DFF  570  by the terminal J, receives the output of the AND  579  by the terminal k, and receives the clock signal CK by the terminal CK. The JKFF  572  receives the latched signal from the terminal Q of the DFF  570  by the terminal J, receives the output of the AND  579  by the terminal k, and receives the output signal from the terminal Q of the JKFF  571  by the terminal CK. The JKFF  573  receives the latched signal from the terminal Q of the DFF  570  by the terminal J, receives the output of the AND  579  by the terminal k, and receives the output signal from the terminal Q of the JKFF  572  by the terminal CK. The JKFF  574  receives the latched signal from the terminal Q of the DFF  570  by the terminal J, receives the output of the AND  579  by the terminal k, and receives the output signal from the terminal Q of the JKFF  573  by the terminal CK. The AND  575  receives the signal from the terminal Q of the JKFF  571 , and the signal from the terminal Q of the JKFF  574 , and outputs a signal obtained by logical AND thereof as the digital signal DCM&lt; 0 &gt;. The AND  576  receives the signal from the terminal Q of the JKFF  572 , and the signal from the terminal Q of the JKFF  574 , and outputs a signal obtained by logical AND thereof as the digital signal DCM&lt; 1 &gt;. The AND  577  receives the signal from the terminal Q of the JKFF  573 , and the signal from the terminal Q of the JKFF  574 , and outputs a signal obtained by logical AND thereof as the digital signal DCM&lt; 2 &gt;. 
     Thus, when the signal CS and the signal OP are logic “H”, the AND  568  outputs the signal of logic “H” to the terminal D of the DFF  560 . When receiving logic “H” at the terminal D, upon the clock signal CK being input, the DFF  560  outputs the signal of logic “H” by the terminal Q. As a result thereof, the combination of digital signals to be output from the terminal Q of each of the JKFF  561 ,  562 , and  563  is counted down from (1, 1, 1) to (0, 0, 0). While the combination of the above digital signals is counted down from (1, 1, 1) to (0, 0, 0), the signal of logic “H” is output from the terminal Q of the JKFF  564 , and accordingly, the signal logic of “H” is input to one of the inputs of the AND  565 ,  566 , and  567 , and accordingly, the combination from the digital signals DCP&lt; 0 &gt; through DCP&lt; 2 &gt; is counted from (1, 1, 1) to (0, 0, 0). Upon the countdown ending and reaching (0, 0, 0), the signal of logic “L” is output from the terminal Q of the JKFF  564 , and accordingly, regardless of the logic of the signal output from the terminal Q of each of the JKFF  561 ,  562 , and  563 , the combination from the digital signals DCP&lt; 0 &gt; to DCP&lt; 2 &gt; is also held in (0, 0, 0). 
     Note that during the above countdown upon the logic of the signal OP becoming logic “L”, the logic of the signal from the terminal Q of the DFF  570  becomes logic “L”, and accordingly, the countdown of the combination from the digital signals DCP&lt; 0 &gt; to DCP&lt; 2 &gt; is ended, and the values thereof are maintained. 
     On the other hand, the combination of the digital signals output from the terminal of each of the JKFF  571 ,  572 , and  573  is counted up from (0, 0, 0) to (1, 1, 1). However, while the combination from the digital signals DCP&lt; 0 &gt; to DCP&lt; 2 &gt; is counted down from (1, 1, 1) toward (0, 0, 0), the signal of logic “L” is output from the terminal Q of the JKFF  574 , and accordingly, the signal logic “L” is input to one of the inputs of the AND  575 ,  576 , and  577 , the combination from the digital signals DCP&lt; 0 &gt; to DCP&lt; 2 &gt; is held in (0, 0, 0). However, upon the countdown ending, and the combination from the digital signals DCP&lt; 0 &gt; to DCP&lt; 2 &gt; reaching (0, 0, 0), the signal of logic “H” is output from the terminal Q of the JKFF  574 , and accordingly, the combination from the digital signals DCM&lt; 0 &gt; to DCM&lt; 2 &gt; is also counted from (0, 0, 0) toward (1, 1, 1) according to the logic of the signal to be output from the terminal Q of each of the JKFF  571 ,  572 , and  573 . 
     Note that during the above count-up upon the logic of the signal OP becoming logic “L”, the logic of the terminal Q of the DFF  570  becomes logic “L”, and accordingly, the count-up of the combination from the digital signals DCM&lt; 0 &gt; to DCM&lt; 2 &gt; ends, and the values thereof are maintained. 
       FIG. 3  illustrates a flowchart relating to the operation of the logic circuit  56  for controlling the delay circuit  51 . A specific example of the logic circuit  56  is illustrated in  FIG. 2D , but it goes without saying that as long as a circuit which operates in accordance with the flowchart in  FIG. 3 , this circuit may be configured in any wise. 
     With operation OP 1 , determination is made whether or not the logic of the signal CS rises to “1” (logic “H”). When the logic of the signal CS rises to “1”, the logic circuit  56  starts its operation, and accordingly, executes the next operation OP 2 . When the logic of the signal CS is “0” (logic “L”), the logic circuit  56  maintains an idle state. 
     With operation OP 2 , upon the logic of the signal CE being set to “0”, i.e., the logic of the signal /CE being set to “1”, so that the binary numeral DCP represented by the digital signals DCP&lt; 0 &gt; to DCP&lt; 2 &gt; becoming (111), the logic circuit  56  outputs these digital signals. Also, so that the binary numeral DCM represented by the digital signals DCM&lt; 0 &gt; to DCM&lt; 2 &gt; becoming (000), the logic circuit  56  outputs these digital signals. 
     With operation OP 3 , after the logic of the signal CS rises to “1”, determination is made whether or not the logic of the clock signal CK rises to “1”. In the case that the logic of the clock signal CK is “1”, the logic circuit  56  maintains its state. 
     With operation OP 4 , the logic circuit  56  determines whether or not the clock signal CK rises from “0” to “1”. In the case that the clock signal CK does not rise from “0” to “1”, the logic circuit  56  maintains its state. In the case that the clock signal CK rises from “0” to “1”, the logic circuit  56  proceeds to the next operation OP 5 . 
     With operation OP 5 , the logic circuit  56  determines whether or not the logic of the signal OP is “1”. In the case that the logic of the signal OP is “0”, the logic circuit  56  externally outputs a signal indicating that the logic of the signal OP is “0”, and receives the signal CE of which the logic is “1”. In the case that the logic of the signal OP is “1”, the logic circuit  56  proceeds to the next operation OP 6 . 
     With operation OP 6 , the logic circuit  56  determines whether or not the binary numeral DCP represented by the digital signals DCP&lt; 0 &gt; to DCP&lt; 2 &gt; is (000). When the binary numeral DCP represented by the digital signals DCP&lt; 0 &gt; to DCP&lt; 2 &gt; is (000), the logic circuit  56  proceeds to the next operation OP 8 . When the binary numeral DCP represented by the digital signals DCP&lt; 0 &gt; to DCP&lt; 2 &gt; is not (000), the logic circuit  56  proceeds to the next operation OP 7 . 
     With operation OP 7 , the logic circuit  56  executes operation wherein 1 is subtracted from the binary numeral DCP represented by the digital signals DCP&lt; 0 &gt; to DCP&lt; 2 &gt;. That is to say, the logic circuit  56  executes the countdown operation of the binary numeral represented by the digital signals DCP&lt; 0 &gt; to DCP&lt; 2 &gt;. Subsequently, the logic circuit  56  proceeds to operation OP 4 . 
     With operation OP 8 , the logic circuit  56  adds 1 to the binary numeral DCM represented by the digital signals DCM&lt; 0 &gt; to DCM&lt; 2 &gt;, i.e., counts up this binary numeral. 
     With operation OP 9 , the logic circuit  56  determines whether or not the clock signal CK rises from “0” to “1”. In the case that the clock signal CK does not rise from “0” to “1”, the logic circuit  56  maintains its state. In the case that the clock signal CK rises from “0” to “1”, the logic circuit  56  proceeds to the next operation OP 10 . 
     With operation OP 10 , the logic circuit  56  determines whether or not the logic of the signal OP is “1”. In the case that the logic of the signal OP is “0”, the logic circuit  56  externally outputs a signal indicating that the logic of the signal OP is “0”, and receives the signal CE of which the logic is “1”. In the case that the logic of the signal OP is “1”, the logic circuit  56  proceeds to the next operation OP 11 . 
     With operation OP 11 , the logic circuit  56  determines whether or not the binary numeral DCM represented by the digital signals DCM&lt; 0 &gt; to DCM&lt; 2 &gt; is (111). In the case that the binary numeral DCM is (111), the logic circuit  56  externally outputs a signal indicating that the logic of the signal OP is “0”, and receives the signal CE of which the logic is “1”. In the case that the binary DCM is not (111), the logic circuit  56  proceeds to operation OP 12 . 
     With operation OP 12 , the logic circuit  56  adds 1 to the binary numeral DCM represented by the digital signals DCM&lt; 0 &gt; to DCM&lt; 2 &gt;, i.e., counts up this binary numeral. 
       FIG. 4  illustrates a timing chart for describing the operation of the logic circuit  56 . 
     The logic of the signal CS changes from logic “L” to logic “H” at point-in-time T 0 , and maintains logic “H” even at point-in-time Tm+2. 
     The signal CE changes from logic “H” to logic “L” at point-in-time T 0 , and changes from logic “L” to logic “H” at point-in-time Tm+2. 
     (VIP−VIM) indicates the voltage difference between the signals VIP and VIM, (VIP−VIM) is 0 while the logic of the signal CS is “L”. 
     The clock signal CK is a signal which repeats the periods of logic “H” and logic “L”, and the periods of logic “H” and logic “L” have generally the same length. 
     The signal L is synchronized with the reversed phase of the clock signal CK, and also the periods of logic “H” and logic “L” are generally the same as with the clock signal CK. 
     The binary numeral DCP is made up of the logic combination of the digital signals DCP&lt; 0 &gt; through DCP&lt; 2 &gt;, and is (111) at point-in-time T 0 , and is counted down toward (000) from point-in-time T 1 . 
     The binary numeral DCM is made up of the logic combination of the digital signals DCM&lt; 0 &gt; through DCM&lt; 2 &gt;, and is (000) at point-in-time T 0 , and is counted up toward (111) when the binary numeral DCM reaches (000). 
     The signal LP is a signal synchronized with the signal L, wherein the leading-edge point-in-time from logic “L” to logic “H” is delayed according to the binary numeral DCP as compared to the signal L. The degree of delay thereof is the maximum when the binary numeral DCP is (111), and is the minimum when the binary numeral DCP is (000). 
     The signal LM is a signal synchronized with the signal L, wherein the leading-edge point-in-time from logic “L” to logic “H” is delayed according to the binary numeral DCM as compared to the signal L. The degree of delay thereof is the maximum when the binary numeral DCM is (000), and is the minimum when the binary numeral DCM is (111). 
     The signal OP is a signal indicating the comparison result of the potential of the signal VIP and the potential of the signal VIM when both of the signals LP and LM are logic “H”. At point-in-time Tm+2, the logic of the signal OP is logic “L”. As a result thereof, the count-up of the binary numeral DCM ends, and the values thereof are maintained. 
       FIGS. 5A and 5B  are diagrams representing change in the signal potentials of the signals OP and OM when the leading-edge point-in-time of the signal LP or signal LM is delayed. 
       FIG. 5A  is a diagram representing change in the signal potentials of the signals OP and OM when the leading-edge period of the signal LM is fixed, and the leading-edge delay amount of the signal LP decreases. (VIP−VIM) indicates the voltage difference between the signals VIP and VIM, and (VIP−VIM) is 0. The signal L is synchronized with the reversed phase of the clock signal CK, and also the period of logic “H”, and the period of logic “L” are generally the same as with the clock signal CK. 
     The logic leading-edge delay amount of the signal LP varies according to the magnitude of the binary numeral DCP made up of the digital signals DCP&lt; 0 &gt; through DCP&lt; 2 &gt; as to the logic leading edge of the signal L. The signal OP is the output signal to be output from the latch unit  20  of the comparison circuit  10 . 
     Now, let us say that the logic leading edge of the signal LP is delayed such as the waveform of the signal LP shown by a dotted line, and the signals L and LM simultaneously logically rise. 
     In this case, the potential of the node A deteriorates toward a potential lower than the potential of the high-potential VDD power source  60  by an amount corresponding to the multiplying of the potential of the high-potential VDD power source  60  by the result obtained by dividing the on resistance of the P-type MOS transistor  21  by the resistance of the whole first current route, but the deterioration of the potential of the node B to be connected to the signal OP is little such as the signal OP shown by a dotted line. This is because high-potential VDD is applied to the node B by the P-type MOS transistor  22  until the logic of the signal LP rises. 
     Accordingly, with the comparison circuit  10 , the potential of the node A is lower than the potential of the node B. As a result thereof, upon the logic of the signal LP rising, the potential difference between the nodes A and B is expanded by the operation of the latch unit  20 , and the potential of the signal OM becomes equal to or smaller than a predetermined threshold from the potential VDD of the high-potential VDD power source  60 . As a result thereof, the logic of the signal OM is determined to be “L”, and the logic of the signal OP is determined to be “H”. 
     Next, upon the binary numeral DCP represented by the digital signals DCP&lt; 0 &gt; through DCP&lt; 2 &gt; being counted down such as the waveform of the signal LP shown by a solid line, the delay amount of the logic leading edge of the signal LP decreases. 
     Thus, the logic of the signal LP rises at a stage wherein the deterioration of the potential of the node A is small. 
     Now, VIP is equal to VIM, and accordingly, if we say that the properties of the on resistances of the N-type MOS transistors  31  and  32  of the input unit  30  are equal as to gate voltage, the similar properties of the P-type MOS transistors  21  and  22  of the latch unit  20  are equal, and further the similar properties of the N-type MOS transistors  23  and  24  of the latch unit  20  are equal, the potential difference between the nodes A and B is amplified as is due to the deterioration of the potential of the node A. 
     However, in any of a case where the properties as to the gate voltage are arranged so that the on resistance of the N-type MOS transistor  31  of the input unit  30  is higher than the on resistance of the N-type MOS transistor  32 , a case where the properties are arranged so that the on resistance of the P-type MOS transistor  21  of the latch unit  20  is lower than the on resistance of the P-type MOS transistor  22 , further a case where the properties are arranged so that the on resistance of the N-type MOS transistor  23  of the latch unit  20  is higher than the on resistance of the N-type MOS transistor  24 , upon the logic of the signal LP rising, even if the potential of the node A deteriorates, the potentials of the nodes A and B may be inverted. In this case, upon the potential of the node B to be connected to the signal OP first deteriorating less than a threshold such as the signal OP shown by a solid line, the signal OP is determined to be logic “L”, and the logic of the signal OM is determined to be “H”. 
     However, the delay amount of the logic leading edge of the signal LP is kept great, whereby the potential of the node A, and the potential of the node B can be prevented from being inverted. In this case, the logic of the signal OM is determined to be “L”, and the logic of the signal OP is determined to be “H”. 
       FIG. 5B  is a diagram representing change in the signal potentials of the signals OP and OM when the leading-edge period of the signal LP is fixed, and the leading-edge delay amount of the signal LM increases. (VIP−VIM) indicates the voltage difference between the signals VIP and VIM, and (VIP−VIM) is 0. The signal L is synchronized with the reversed phase of the clock signal CK, and also the period of logic “H”, and the period of logic “L” are generally the same as with the clock signal CK. 
     The logic leading-edge delay amount of the signal LM varies according to the magnitude of the binary numeral DCM made up of the digital signals DCM&lt; 0 &gt; through DCM&lt; 2 &gt; as to the logic leading edge of the signal L. The signal OM is the output signal to be output from the latch unit  20  of the comparison circuit  10 . 
     First, let us say that that the logic leading edge of the signal LM is not delayed so much such as the waveform of the signal LM shown by a dotted line, and the signals L and LP simultaneously logically rise. 
     In this case, the potential of the node B deteriorates toward a potential lower than the potential of the high-potential VDD power source  60  by an amount corresponding to multiplying of the potential of the high-potential VDD power source  60  by the result obtained by dividing the on resistance of the P-type MOS transistor  22  by the resistance of the whole second current route, and the deterioration of the potential of the node A to be connected to the signal OM occurs such as the signal OM shown by a dotted line. 
     Now, VIP is equal to VIM, and accordingly, if we say that the properties of the on resistances of the N-type MOS transistors  31  and  32  of the input unit  30  are equal as to gate voltage, the similar properties of the P-type MOS transistors  21  and  22  of the latch unit  20  are equal, and further the similar properties of the N-type MOS transistors  23  and  24  of the latch unit  20  are equal, the potential difference between the nodes A and B is amplified as is due to the deterioration of the potential of the node B. 
     However, in any case of a case where the properties as to the gate voltage are arranged so that the on resistance of the N-type MOS transistor  31  of the input unit  30  is lower than the on resistance of the N-type MOS transistor  32 , a case where the properties are arranged so that the on resistance of the P-type MOS transistor  21  of the latch unit  20  is higher than the on resistance of the P-type MOS transistor  22 , and further a case where the properties are arranged so that the on resistance of the N-type MOS transistor  23  of the latch unit  20  is lower than the on resistance of the N-type MOS transistor  24 , upon the logic of the signal LM rising, even if the potential of the node B deteriorates, the potentials of the nodes A and B may be inverted such as the signal OM indicated with a dotted line. In this case, the signal OP is determined to be logic “L” such as the signal OM shown by a dotted line, and the logic of the signal OM is determined to be “H” such as the signal OP shown by a dotted line. 
     However, upon the logic leading-edge delay amount of the signal LM increasing such as the signal LM shown by a solid line, the deterioration of the potential of the node B to be connected to the signal OP shown by a solid line becomes great, whereby the potential of the node A, and the potential of the node B is prevented from being inverted. As a result thereof, according to the latch unit  30 , the logic of the signal OM becomes logic “H”, and the logic of the signal OP becomes logic “L”. 
     Thus, in any case of a case where the properties as to the gate voltage between the on resistance of the N-type MOS transistor  31  of the input unit  30  and the on resistance of the N-type MOS transistor  32  differ, a case where the similar properties of the on resistance of the P-type MOS transistor  21  of the latch unit  20 , and the on resistance of the P-type MOS transistor  22  differ, and further a case where the similar properties of the on resistance of the N-type MOS transistor  23  of the latch unit  20 , and the on resistance of the N-type MOS transistor  24  differ, the logic leading-edge delay amount of the signal LM or signal LP is adjusted, whereby the comparison circuit  10  can be caused to execute the same operation as in the case that the on resistance properties of each MOS resistor, and the transistor corresponding thereto are matched. As a result thereof, with the comparison circuit  10 , in the event of executing comparison between the voltage of the signal VIP and the voltage of the signal VIM, error can be prevented from occurring due to the difference of the properties of the MOS transistors making up the comparison circuit  10 , or the amplification and held properties of the latch unit  20 . 
     Thus, the comparison circuit  10  according to the first embodiment is a comparison circuit including an input unit  30  made up of a first MOS transistor (N-type MOS transistor  31 ) configured to receive a first signal at the gate electrode, and a second MOS transistor configured to receive a second signal at the gate; a latch circuit  20  configured to amplify potential difference between a first current route where the current is controlled by the first MOS transistor according to the voltage of the first signal, and a second current route where the current is controlled by the second MOS transistor according to the voltage of the second signal; a comparative operation control unit including a first switch configured to execute supply or blocking of supply of high-potential VDD to the drain of the first MOS transistor by a third current route different from the first current route, and a second switch configured to execute supply or blocking of supply of the high-potential VDD to the drain of the second MOS transistor by a fourth current route different from the second current route, and a third switch configured to execute supply or blocking of supply of a low potential to the first current route and second current route; and a comparative operation setting unit configured to set the period of supply or blocking of supply of the first switch, second switch, and third switch. 
     The comparative operation setting unit (comparative operation setting unit  50 ) includes a delay circuit configured to determine a period between a blocking period of supply of high-potential VDD by the first switch, and a blocking period of supply of high-potential VDD by the second switch, and a setting circuit configured to execute setting of a period. 
     Thus, with the comparison circuit  10  according to the first embodiment, timing of supply or blocking of supply of high-potential VDD to the third current route or the fourth current route by the first switch and the second switch, whereby the potentials of the first current route and the second current route before comparison can be controlled. 
     As a result thereof, the potentials of the first current route and the second current route before comparison are controlled, whereby comparison error along with difference of the properties relating to the gate voltage of the on resistances of the first MOS transistor and the second MOS transistor which receive the input signals VIP and VIM can be controlled. 
     Second Embodiment 
       FIG. 6  is a circuit diagram illustrating a comparison circuit  100  according to the second embodiment. The comparison circuit  100  includes a P-type MOS transistor latch unit  120 , an input  130 , an N-type MOS transistor latch unit  125 , a comparative operation control circuit  140 , and a comparative operation setting circuit  150 . 
     The comparative operation control unit  140  includes P-type MOS transistors  143 ,  144 ,  145 , and  146 , and N-type MOS transistors  141 ,  142 ,  147 , and  148 . 
     The N-type MOS transistor  141  has a drain to be connected to the source of an N-type MOS transistor  131  of the input  130 , a source to be connected to the ground VSS  70 , and a gate for receiving a signal LM 2 . 
     The N-type MOS transistor  142  has a drain to be connected to the source of an N-type MOS transistor  132  of the input  130 , a source to be connected to the ground VSS  70 , and a gate for receiving a signal LP 2 . 
     The N-type MOS transistor  141  supplies the ground potential from the ground VSS  70  to the input unit  130  when the logic of the signal LM 2  is “H”, and blocks supply of the ground potential from the ground VSS  70  to the input unit  130  when the logic is “H”. The N-type MOS transistor  141  serves as a switch for connecting or blocking between the input unit  130  and the ground VSS  70 . 
     The N-type MOS transistor  142  supplies the ground potential from the ground VSS  70  to the input unit  130  when the logic of the signal LP 2  is “H”, and blocks supply of the ground potential from the ground VSS  70  to the input unit  130  when the logic is “L”. The N-type MOS transistor  142  serves as a switch for connecting or blocking between the input unit  130  and the ground VSS  70 . 
     The P-type MOS transistor  143  has a source to be connected to the high-potential VDD power source  60 , a drain to be connected to the drain of the P-type MOS transistor  121  of the P-type MOS transistor latch unit  120 , and a gate for receiving a signal LM 1 . 
     The P-type MOS transistor  143  blocks supply of the high-potential VDD from the high-potential VDD power source  60  to the P-type MOS transistor latch unit  120  when the logic of the signal LM 1  is “H”, and supplies the high-potential VDD from the high-potential VDD power source  60  to the P-type MOS transistor latch unit  120  when the logic is “L”. The P-type MOS transistor  143  serves as a switch for connecting or blocking between the P-type MOS transistor latch unit  120  and the high-potential VDD power source  60 . 
     The P-type MOS transistor  144  has a source to be connected to the high-potential VDD power source  60 , a drain to be connected to the drain of the N-type MOS transistor  126  of the N-type MOS transistor latch unit  125 , and a gate for receiving a signal LM 0 . 
     The P-type MOS transistor  144  blocks supply of the high-potential VDD from the high-potential VDD power source  60  to the N-type MOS transistor latch unit  125  when the logic of the signal LM 0  is “H”, and supplies the high-potential VDD from the high-potential VDD power source  60  to the N-type MOS transistor latch unit  125  when the logic is “L”. The P-type MOS transistor  144  serves as a switch for connecting or blocking between the N-type MOS transistor latch unit  125  and the high-potential VDD power source  60 . 
     The P-type MOS transistor  145  has a source to be connected to the high-potential VDD power source  60 , a drain to be connected to the drain of the N-type MOS transistor  127  of the N-type MOS transistor latch unit  125 , and a gate for receiving a signal LP 0 . 
     The P-type MOS transistor  145  supplies the high-potential VDD from the high-potential VDD power source  60  to the N-type MOS transistor latch unit  125  when the logic of the signal LM 0  is “H”, and blocks supply of the high-potential VDD from the high-potential VDD power source  60  to the N-type MOS transistor latch unit  125  when the logic is “L”. The P-type MOS transistor  145  serves as a switch for connecting or blocking between the N-type MOS transistor latch unit  125  and the high-potential VDD power source  60 . 
     The P-type MOS transistor  146  has a source to be connected to the high-potential VDD power source  60 , a drain to be connected to the drain of the P-type MOS transistor  122  of the P-type MOS transistor latch unit  120 , and a gate for receiving a signal LP 1 . 
     The P-type MOS transistor  146  blocks supply of the high-potential VDD from the high-potential VDD power source  60  to the P-type MOS transistor latch unit  120  when the logic of the signal LP 1  is “H”, and supplies the high-potential VDD from the high-potential VDD power source  60  to the P-type MOS transistor latch unit  120  when the logic is “L”. The P-type MOS transistor  146  serves as a switch for connecting or blocking between the P-type MOS transistor latch unit  120  and the high-potential VDD power source  60 . 
     The N-type MOS transistor  147  has a drain to be connected to the drain of the P-type MOS transistor  121  of the P-type MOS transistor latch unit  120 , a source to be connected to the drain of the N-type MOS transistor  126  of the N-type MOS transistor latch unit  125 , and a gate for receiving a signal LM 2 . 
     The N-type MOS transistor  148  has a drain to be connected to the drain of the P-type MOS transistor  122  of the P-type MOS transistor latch unit  120 , a source to be connected to the drain of the N-type MOS transistor  127  of the N-type MOS transistor latch unit  125 , and a gate for receiving a signal LP 2 . 
     The input unit  130  includes the N-type MOS transistors  131  and  132 . The N-type MOS transistor  131  has a drain to be connected to the source of the N-type MOS transistor  126  of the N-type MOS transistor latch unit  125 , a source to be connected to the drain of the N-type MOS transistor  141  of the comparative operation control circuit  140 , and a gate for receiving the input signal VIP. The on resistance value of the N-type MOS transistor  131  varies depending on the potential of the input signal VIP. 
     The N-type MOS transistor  132  has a drain to be connected to the source of the N-type MOS transistor  127  of the N-type MOS transistor latch unit  125 , a source to be connected to the drain of the N-type MOS transistor  142  of the comparative operation control circuit  140 , and a gate for receiving the signal VIM. The on resistance value of the N-type MOS transistor  132  varies depending on the potential of the signal VIM. 
     The P-type MOS transistor latch unit  120  includes the P-type MOS transistors  121  and  122 . The N-type MOS transistor latch unit  125  includes the N-type MOS transistors  126  and  127 . 
     The P-type MOS transistor  121  has a drain to be connected to the drain of the N-type MOS transistor  147 , a gate to be connected to the drain of the P-type MOS transistor  122 , and a source to be connected to the high-potential VDD power source  60 . 
     The P-type MOS transistor  122  has a drain to be connected to the drain of the N-type MOS transistor  148 , a gate to be connected to the drain of the P-type MOS transistor  121 , and a source to be connected to the high-potential VDD power source  60 . 
     The N-type MOS transistor  126  has a drain to be connected to the source of the N-type MOS transistor  147 , a gate to be connected to the drain of the N-type MOS transistor  127 , and a source to be connected to the drain of the N-type MOS transistor  131  of the input unit  130 . 
     The N-type MOS transistor  127  has a drain to be connected to the source of the N-type MOS transistor  148 , a gate to be connected to the drain of the N-type MOS transistor  126 , and a source to be connected to the drain of the N-type MOS transistor  132  of the input unit  130 . 
     The output signals OM and OP are output from the P-type MOS transistor latch unit  120 . The output signal OM is connected to the node A between the drain of the P-type MOS transistor  121  and the drain of the N-type MOS transistor  147 . The output signal OP is connected to the node B between the drain of the P-type MOS transistor  122  and the drain of the N-type MOS transistor  148 . 
     The gate of the P-type MOS transistor  121  of the P-type MOS transistor latch unit  120  and the drain of the P-type MOS transistor  122  are connected to the node B, and the gate of the P-type MOS transistor  122  and the drain of the P-type MOS transistor  121  are connected to the node A. That is to say, the P-type MOS transistor  121  and the P-type MOS transistor  122  are connected crosswise to the nodes A and B, and accordingly, the P-type MOS transistor  121  and the P-type MOS transistor  122  amplify potential difference between the nodes A and B. 
     The gate of the N-type MOS transistor  126  and the drain of the N-type MOS transistor  127  of the N-type MOS transistor latch unit  125  are connected to the drain (node D) of the N-type MOS Transistor  148 , and the gate of the N-type MOS transistor  127  and the drain of the N-type MOS transistor  126  are connected to the drain (node C) of the N-type MOS transistor  147 . That is to say, the N-type MOS transistor  126  and the N-type MOS transistor  127  are connected crosswise to the nodes C and D, and accordingly, the N-type MOS transistor  126  and the N-type MOS transistor  127  amplify potential difference between the nodes C and D. 
     Thus, the P-type MOS transistor  121  of the P-type MOS transistor latch unit  120 , the N-type MOS transistor  147 , the N-type MOS transistor  126 , the N-type MOS transistor  131 , and the N-type MOS transistor  141  are serially connected between the high-potential VDD power source  60  and the ground VSS  70 , and make up a first current route including the nodes A and C. The P-type MOS transistor  122  of the P-type MOS transistor latch unit  120 , the N-type MOS transistor  148 , the N-type MOS transistor  127 , the N-type MOS transistor  132 , and the N-type MOS transistor  142  are serially connected between the high-potential VDD power source  60  and the ground VSS  70 , and make up a second current route including the nodes B and D. 
     Therefore, when the signals LM 0 , LM 1 , LM 2 , LP 0 , LP 1 , and LP 2  are “H”, supply of the high-potential VDD to the P-type MOS transistor latch unit  120  and the N-type MOS transistor latch unit  125  is blocked by the P-type MOS transistors  143 ,  144 ,  145 , and  146 , and the ground potential VSS is supplied to the input unit  130 , the P-type MOS transistor latch unit  120 , and the N-type MOS transistor latch unit  125  by the N-type MOS transistors  141  and  142 . 
     In the above case, the on resistance of the N-type MOS transistor  131  varies according to the potential of the input signal VIP, and accordingly, upon the potential of the input signal VIP decreasing, the potentials of the nodes A and C increase, and the on resistance of the N-type MOS transistor  132  varies according to the potential of the input signal VIM, and accordingly, upon the potential of the input signal VIM increasing contrary to the input signal VIP, the potentials of the nodes B and D decrease. On the other hand, conversely, upon the potential of the input signal VIP increasing, the potentials of the nodes A and C decrease, and upon the potential of the input signal VIM decreasing, the potentials of the nodes B and D increase. 
     Note that, when the signals LM 0 , LM 1 , LM 2 , LP 0 , LP 1 , and LP 2  are “L”, the high-potential VDD is supplied to the P-type MOS transistor latch unit  120 , and the N-type MOS transistor latch unit  125 , and the input unit  130  by the P-type MOS transistors  143 ,  144 ,  145 , and  146 , and the supply of the ground potential to the input unit  130  is blocked by the N-type MOS transistors  141  and  142 . 
     As a result thereof, potential difference between the nodes A and B is 0, or almost eliminated. 
     The comparative operation setting circuit  150  is made up of a circuit similar to the comparative operation setting circuit  50  according to the first embodiment for driving the signals LM 0  and LP 0 , a circuit similar to the comparative operation setting circuit  50  for driving the signals LM 1  and LP 1 , and a circuit similar to the comparative operation setting circuit  50  for driving the signals LM 2  and LP 2 . Also, the circuits similar to the comparative operation setting circuit  50  include a delay circuit  151 , and a logic circuit  156  for controlling the delay circuit  151 . The delay circuit  151  and the logic circuit  156  are the same circuits as the delay circuit  51  and the logic circuit  56  according to the first embodiment. Now, it goes without saying that delay between the signals LM 0  and LP 0 , delay between the signals LM 1  and LP 1 , and delay between the signals LM 2  and LP 2  may be set to the same delay amount, or may be set separately. It is further needless to say that an arrangement may be made where just one of the delays is set. 
     Thus, the comparison circuit  100  according to the second embodiment includes an input unit ( 130 ) made up of a first MOS transistor (N-type MOS transistor  131 ) for receiving a first signal (signal VIP) at the gate electrode, and a second MOS transistor (N-type MOS transistor  132 ) for receiving a second signal (signal VIM) at the gate electrode; a latch circuit (P-type MOS transistor latch circuit  120 , N-type MOS transistor latch circuit  125 ) for amplifying potential difference between a node within a first current route where an electric current is controlled by the first MOS transistor according to the voltage of the first signal, and a node within a second current route where an electric current is controlled by the second MOS transistor according to the voltage of the second signal; a comparative operation setting unit (comparative operation setting circuit  150 ) including a first switch (P-type MOS transistor  144 ) for executing supply or blocking of supply of the high-potential VDD to the drain of the first MOS transistor by a third current route different from the first current route, a second switch (P-type MOS transistor  145 ) for executing supply or blocking of supply of the high-potential VDD to the drain of the second MOS transistor by a fourth current route different from the second current route, and a third switch (N-type MOS transistors  141  and  147 ) and a fourth switch (N-type MOS transistors  142  and  148 ) which execute supply or blocking of supply of a low potential to the first current route and the second current route; a fifth switch (P-type MOS transistor  143 ) to be connected to the latch circuit; a sixth switch (P-type MOS transistor  146 ) to be connected to the latch circuit; and a comparative operation control unit (comparative operation control circuit  140 ) for controlling supply or blocking of the first through sixth switches. 
     Thus, supply or blocking of any of the first through sixth switches is controlled, whereby the potentials of the first current route and the second current route before start of comparison can also be controlled at the comparison circuit  100  according to the second embodiment. 
     As a result thereof, the potentials of the first current route and the second current route before start of comparison are controlled, whereby comparison error along with different properties relating to the gate voltages of on resistances of the first and second MOS transistors which receive input signals VIP and VIM can be controlled. Note that it goes without saying that comparison error can be controlled by controlling any of the first through sixth switches. 
     Further, with the comparison circuit  100  according to the second embodiment, the latch circuit unit is made up of the N-type MOS transistor latch circuit  120  and the P-type MOS transistor latch circuit  125 , and accordingly, control of supply or blocking of supply of the high-potential VDD to the latch circuits is ensured by providing the fifth switch (P-type MOS transistor  143 ) and the sixth switch (P-type MOS transistor  146 ) to be connected to the latch circuits. 
     Third Embodiment 
       FIG. 7  is a circuit diagram illustrating a comparison circuit  200  according to the third embodiment. The comparison circuit  200  includes a P-type MOS transistor latch unit  220 , an input unit  230 , an N-type MOS transistor latch unit  225 , a comparative operation control circuit  240 , and a comparative operation setting circuit  250 . 
     The comparative operation control circuit  240  includes N-type MOS transistors  243 ,  244 ,  245 , and  246 , and P-type MOS transistors  241 ,  242 ,  247 , and  248 . 
     The P-type MOS transistor  241  has a drain to be connected to the source of a P-type MOS transistor  231  of the input unit  230 , a source to be connected to the high-potential VDD power source  60 , and a gate for receiving a signal LM 2 . 
     The P-type MOS transistor  242  has a drain to be connected to the source of a P-type MOS transistor  232  of the input unit  230 , a source to be connected to the high-potential VDD power source  60 , and a gate for receiving a signal LP 2 . 
     The P-type MOS transistor  241  supplies the high-potential VDD from the high-potential VDD power source  60  to the input unit  230  when the logic of the signal LM 2  is “L”, and blocks supply of the high-potential VDD from the high-potential VDD power source  60  to the input unit  230  when the logic is “H”. The P-type MOS transistor  241  serves as a switch for connecting or blocking between the input unit  230  and the high-potential VDD power source  60 . 
     The P-type MOS transistor  242  supplies the high-potential VDD from the high-potential VDD power source  60  to the input unit  230  when the logic of the signal LP 2  is “L”, and blocks supply of the high-potential VDD from the high-potential VDD power source  60  to the input unit  230  when the logic is “H”. The P-type MOS transistor  242  serves as a switch for connecting or blocking between the input unit  230  and the high-potential VDD power source  60 . 
     The N-type MOS transistor  243  has a source to be connected to a ground VSS power source  70 , a drain to be connected to the drain of an N-type MOS transistor  226  of the N-type MOS transistor latch unit  225 , and a gate for receiving the signal LM 1 . 
     The N-type MOS transistor  243  supplies the high-potential VDD from the high-potential VDD power source  60  to the N-type MOS transistor latch unit  225  when the logic of the signal LM 1  is “H”, and blocks supply of the high-potential VDD from the high-potential VDD power source  60  to the N-type MOS transistor latch unit  225  when the logic is “L”. The N-type MOS transistor  243  serves as a switch for connecting or blocking between the N-type MOS transistor latch unit  225  and the high-potential VDD power source  60 . 
     The N-type MOS transistor  244  has a source to be connected to the ground VSS power source  70 , a drain to be connected to the source of a P-type MOS transistor  221  of the P-type MOS transistor latch unit  220 , and a gate for receiving the signal LM 0 . 
     The N-type MOS transistor  244  supplies the ground VSS potential from the ground VSS power source  70  to the P-type MOS transistor latch unit  220  when the logic of the signal LM 0  is “H”, and blocks supply of the ground VSS potential from the ground VSS power source  70  to the P-type MOS transistor latch unit  220  when the logic is “L”. The N-type MOS transistor  244  serves as a switch for connecting or blocking between the P-type MOS transistor latch unit  220  and the ground VSS power source  70 . 
     The N-type MOS transistor  245  has a source to be connected to the ground VSS power source  70 , a drain to be connected to the source of a P-type MOS transistor  222  of the P-type MOS transistor latch unit  220 , and a gate for receiving the signal LP 0 . 
     The N-type MOS transistor  245  supplies the ground VSS potential from the ground VSS power source  70  to the P-type MOS transistor latch unit  220  when the logic of the signal LP 0  is “H”, and blocks supply of the ground VSS potential from the ground VSS power source  70  to the P-type MOS transistor latch unit  220  when the logic is “L”. The N-type MOS transistor  245  serves as a switch for connecting or blocking between the P-type MOS transistor latch unit  220  and the ground VSS power source  70 . 
     The N-type MOS transistor  246  has a source to be connected to the ground VSS power source  70 , a drain to be connected to the drain of an N-type MOS transistor  227  of the N-type MOS transistor latch unit  225 , and a gate for receiving the signal LP 1 . 
     The N-type MOS transistor  246  supplies the ground VSS from the ground VSS power source  70  to the N-type MOS transistor latch unit  225  when the logic of the signal LP 1  is “H”, and blocks supply of the ground VSS from the ground VSS power source  70  to the N-type MOS transistor latch unit  225  when the logic is “L”. The N-type MOS transistor  246  serves as a switch for connecting or blocking between the N-type MOS transistor latch unit  225  and the ground VSS power source  70 . 
     The P-type MOS transistor  247  has a source to be connected to the drain of the P-type MOS transistor  221  of the P-type MOS transistor latch unit  220 , a drain to be connected to the drain of the N-type MOS transistor  226  of the N-type MOS transistor latch unit  225 , and a gate for receiving the signal LM 2 . 
     The P-type MOS transistor  248  has a source to be connected to the drain of the P-type MOS transistor  222  of the P-type MOS transistor latch unit  220 , a drain to be connected to the drain of the N-type MOS transistor  227  of the N-type MOS transistor latch unit  225 , and a gate for receiving the signal LP 2 . 
     The input unit  230  includes the P-type MOS transistors  231  and  232 . The P-type MOS transistor  231  has a drain to be connected to the source of the P-type MOS transistor  221  of the P-type MOS transistor latch unit  220 , a source to be connected to the drain of the P-type MOS transistor  241  of the comparative operation control circuit  240 , and a gate for receiving an input signal VIP. The on resistance value of the P-type MOS transistor  231  varies according to the potential of the input signal VIP. 
     The P-type MOS transistor  232  has a drain to be connected to the source of the P-type MOS transistor  222  of the P-type MOS transistor latch unit  220 , a source to be connected to the drain of the P-type MOS transistor  242  of the comparative operation control circuit  240 , and a gate for receiving a signal VIM. The on resistance value of the P-type MOS transistor  232  varies according to the potential of the signal VIM. 
     The P-type MOS transistor latch unit  220  includes the P-type MOS transistors  221  and  222 . The N-type MOS transistor latch unit  225  includes the N-type MOS transistors  226  and  227 . 
     The P-type MOS transistor  221  has a drain to be connected to the source of the P-type MOS transistor  247 , a source to be connected to the drain of the P-type MOS transistor  231 , and a gate to be connected to the drain of the P-type MOS transistor  222 . 
     The P-type MOS transistor  222  has a drain to be connected to the source of the P-type MOS transistor  248 , a source to be connected to the drain of the P-type MOS transistor  232 , and a gate to be connected to the drain of the P-type MOS transistor  221 . 
     The N-type MOS transistor  226  has a drain to be connected to the drain of the P-type MOS transistor  247 , a gate to be connected to the drain of the N-type MOS transistor  227 , and a source to be connected to the ground VSS  70 . 
     The N-type MOS transistor  227  has a drain to be connected to the drain of the P-type MOS transistor  248 , a gate to be connected to the drain of the N-type MOS transistor  226 , and a source to be connected to the ground VSS  70 . 
     The output signals OM and OP are output from the N-type MOS transistor latch unit  225 . The output signal OM is connected to the node A between the drain of the N-type MOS transistor  226  and the drain of the P-type MOS transistor  247 . The output signal OP is connected to the node B between the drain of the N-type MOS transistor  227  and the drain of the P-type MOS transistor  248 . 
     The gate of the N-type MOS transistor  226  of the N-type MOS transistor latch unit  225  and the drain of the N-type MOS transistor  227  are connected to the node B, and the gate of the N-type MOS transistor  227  and the drain of the N-type MOS transistor  226  are connected to the node A. That is to say, the N-type MOS transistor  226  and the N-type MOS transistor  227  are connected crosswise to the nodes A and B, and accordingly, the N-type MOS transistor  226  and the N-type MOS transistor  227  amplify potential difference between the nodes A and B. 
     The gate of the P-type MOS transistor  221  and the drain of the P-type MOS transistor  222  of the P-type MOS transistor latch unit  220  are connected to the source (node D) of the P-type MOS Transistor  248 , and the gate of the P-type MOS transistor  222  and the drain of the P-type MOS transistor  221  are connected to the source (node C) of the P-type MOS transistor  247 . That is to say, the P-type MOS transistor  226  and the P-type MOS transistor  227  are connected crosswise to the nodes C and D, and accordingly, the P-type MOS transistor  221  and the P-type MOS transistor  222  amplify potential difference between the nodes C and D. 
     Thus, the P-type MOS transistor  221  of the P-type MOS transistor latch unit  220 , the P-type MOS transistor  247 , the N-type MOS transistor  226 , the P-type MOS transistor  231 , and the P-type MOS transistor  241  are serially connected between the high-potential VDD power source  60  and the ground VSS  70 , and make up a first current route including the nodes A and C. The P-type MOS transistor  222  of the P-type MOS transistor latch unit  220 , the P-type MOS transistor  248 , the N-type MOS transistor  227 , the P-type MOS transistor  232 , and the P-type MOS transistor  242  are serially connected between the high-potential VDD power source  60  and the ground VSS  70 , and make up a second current route including the nodes B and D. 
     Therefore, when the signals LM 0 , LM 1 , LM 2 , LP 0 , LP 1 , and LP 2  are “L”, supply of the ground VSS to the P-type MOS transistor latch unit  220  and the N-type MOS transistor latch unit  225  is blocked by the N-type MOS transistors  243 ,  244 ,  245 , and  246 , and the high-potential VDD is supplied to the input unit  230 , the P-type MOS transistor latch unit  220 , and the N-type MOS transistor latch unit  225  by the P-type MOS transistors  241  and  242 . 
     In the above case, the on resistance of the P-type MOS transistor  231  varies according to the potential of the input signal VIP, and accordingly, upon the potential of the input signal VIP decreasing, the potentials of the nodes A and C decrease, and the on resistance of the N-type MOS transistor  232  varies according to the potential of the input signal VIM, and accordingly, upon the potential of the input signal VIM increasing contrary to the input signal VIP, the potentials of the nodes B and D increase. On the other hand, conversely, upon the potential of the input signal VIP increasing, the potentials of the nodes A and C increase, and upon the potential of the input signal VIM decreasing, the potentials of the nodes B and D decrease. 
     Note that, when the signals LM 0 , LM 1 , LM 2 , LP 0 , LP 1 , and LP 2  are “H”, the ground VSS is supplied to the P-type MOS transistor latch unit  220 , the N-type MOS transistor latch unit  225 , and the input unit  230  by the N-type MOS transistors  243 ,  244 ,  245 , and  246 , and the supply of the high-potential VDD to the input unit  230  is blocked by the P-type MOS transistors  241  and  242 . 
     As a result thereof, potential difference between the nodes A and B is 0, or almost eliminated. 
     The comparative operation setting circuit  250  is made up of a circuit similar to the comparative operation setting circuit  50  according to the first embodiment for driving the signals LM 0  and LP 0 , a circuit similar to the comparative operation setting circuit  50  for driving the signals LM 1  and LP 1 , and a circuit similar to the comparative operation setting circuit  50  for driving the signals LM 2  and LP 2 . Also, the circuits similar to the comparative operation setting circuit  50  include a delay circuit  251 , and a logic circuit  256  for controlling the delay circuit  251 . Also, the delay circuit  251  and the logic circuit  256  are the same circuits as the delay circuit  51  and the logic circuit  56 . Now, it goes without saying that delay between the signals LM 0  and LP 0 , delay between the signals LM 1  and LP 1 , and delay between the signals LM 2  and LP 2  may be set to the same delay amount, or may be set separately. It is further needless to say that an arrangement may be made where just one of the delays is set. 
     Thus, the comparison circuit  200  according to the third embodiment includes an input unit ( 230 ) made up of a first MOS transistor (P-type MOS transistor  231 ) for receiving a first signal (signal VIP) at the gate electrode, and a second MOS transistor (P-type MOS transistor  232 ) for receiving a second signal (signal VIM) at the gate electrode; a latch circuit (P-type MOS transistor latch circuit  220 , N-type MOS transistor latch circuit  225 ) for amplifying potential difference between a first current route where an electric current is controlled by the first MOS transistor according to the voltage of the first signal, and a second current route where an electric current is controlled by the second MOS transistor according to the voltage of the second signal; a comparative operation setting unit (comparative operation setting circuit  250 ) including a first switch (N-type MOS transistor  244 ) for executing supply or blocking of supply of the ground VSS to the drain of the first MOS transistor by a third current route different from the first current route, a second switch (N-type MOS transistor  245 ) for executing supply or blocking of supply of the ground VSS to the drain of the second MOS transistor by a fourth current route different from the second current route, and a third switch (P-type MOS transistors  241  and  247 ) and a fourth switch (P-type MOS transistors  242  and  248 ) which execute supply or blocking of supply of the high-potential VDD to the first current route and the second current route; a fifth switch (N-type MOS transistor  243 ) to be connected to the latch circuit; a sixth switch (N-type MOS transistor  246 ) to be connected to the latch circuit; and a comparative operation control unit (comparative operation control circuit  240 ) for controlling supply or blocking of the first through sixth switches. 
     Thus, supply or blocking timing of any of the first through sixth switches is controlled, whereby the potentials of the first current route and the second current route before start of comparison can also be controlled. 
     As a result thereof, the potentials of the first current route and the second current route before start of comparison are controlled, whereby comparison error along with different properties relating to the gate voltages of on resistances of the first and second MOS transistors which receive input signals VIP and VIM can be controlled. Note that it goes without saying that comparison error can be controlled by controlling any of the first through sixth switches. 
     Further, with the comparison circuit  200  according to the third embodiment, the latch circuit portion is made up of the N-type MOS transistor latch circuit  220  and the P-type MOS transistor latch circuit  225 , and accordingly, control of supply or blocking of supply of the ground VSS to the latch circuits is ensured by providing the fifth switch (N-type MOS transistor  243 ) and the sixth switch (N-type MOS transistor  246 ) to be connected to the latch circuits. 
     Fourth Embodiment 
       FIG. 8  is a circuit diagram illustrating a comparison circuit  300  according to the fourth embodiment. The comparison circuit  300  includes a P-type MOS transistor latch unit  320 , an input unit  330 , an N-type MOS transistor latch unit  325 , a comparative operation control circuit  340 , and a comparative operation setting circuit  350 . 
     The comparative operation control circuit  340  includes N-type MOS transistors  343  and  344 , and P-type MOS transistors  341  and  342 . 
     The P-type MOS transistor  341  has a drain to be connected to the drain of an N-type MOS transistor  331  of the input unit  330 , a source to be connected to the high-potential VDD power source  60 , and a gate for receiving a signal LM 1 . 
     Note that the drain of the P-type MOS transistor  341 , the drain of the N-type MOS transistor  331  of the input unit  330 , the drain of the P-type MOS transistor  321  of the P-type MOS transistor latch unit  320 , and the drain of the N-type MOS transistor  326  of the N-type MOS transistor latch unit  325  are all connected to the node A within the comparison circuit  300 . The comparison circuit  300  outputs an output signal OM from the node A. 
     The P-type MOS transistor  342  has a drain to be connected to the drain of an N-type MOS transistor  332  of the input unit  330 , a source to be connected to the high-potential VDD power source  60 , and a gate for receiving a signal LP 1 . 
     Note that the drain of the P-type MOS transistor  342 , the drain of the N-type MOS transistor  332  of the input unit  330 , the drain of the P-type MOS transistor  322  of the P-type MOS transistor latch unit  320 , and the drain of the N-type MOS transistor  327  of the N-type MOS transistor latch unit  325  are all connected to the node B within the comparison circuit  300 . The comparison circuit  300  outputs an output signal OP from the node B. 
     The P-type MOS transistor  341  supplies the high-potential VDD from the high-potential VDD power source  60  to the input unit  330  when the logic of the signal LM 1  is “L”, and blocks supply of the high-potential VDD from the high-potential VDD power source  60  to the input unit  330  when the logic is “H”. The P-type MOS transistor  341  serves as a switch for connecting or blocking between the input unit  330  and the high-potential VDD power source  60 . 
     The P-type MOS transistor  342  supplies the high-potential VDD from the high-potential VDD power source  60  to the input unit  330  when the logic of the signal LP 1  is “L”, and blocks supply of the high-potential VDD from the high-potential VDD power source  60  to the input unit  330  when the logic is “H”. The P-type MOS transistor  342  serves as a switch for connecting or blocking between the input unit  330  and the high-potential VDD power source  60 . 
     The N-type MOS transistor  343  has a source to be connected to the ground VSS power source  70 , a drain to be connected to the source of the N-type MOS transistor  331  of the input unit  330 , and a gate for receiving a signal LM 0 . 
     The N-type MOS transistor  343  supplies the ground VSS from the ground VSS power source  70  to the input unit  330  when the logic of the signal LM 0  is “H”, and blocks supply of the ground VSS from the ground VSS power source  70  to the input unit  330  when the logic is “L”. The N-type MOS transistor  343  serves as a switch for connecting or blocking between the input unit  330  and the ground VSS power source  70 . 
     The N-type MOS transistor  344  has a source to be connected to the ground VSS power source  70 , a drain to be connected to the source of the N-type MOS transistor  332  of the input unit  330 , and a gate for receiving a signal LP 0 . 
     The N-type MOS transistor  344  supplies the ground VSS from the ground VSS power source  70  to the input unit  330  when the logic of the signal LP 0  is “H”, and blocks supply of the ground VSS from the ground VSS power source  70  to the input unit  330  when the logic is “L”. The N-type MOS transistor  344  serves as a switch for connecting or blocking between the input unit  330  and the ground VSS power source  70 . 
     The input unit  330  includes the N-type MOS transistors  331  and  332 . The N-type MOS transistor  331  has a drain to be connected to the drain of the P-type MOS transistor  321  of the P-type MOS transistor latch unit  320 , a source to be connected to the drain of the N-type MOS transistor  343  of the comparative operation control circuit  340 , and a gate for receiving an input signal VIP. The on resistance value of the N-type MOS transistor  331  varies according to the potential of the input signal VIP. 
     The N-type MOS transistor  332  has a drain to be connected to the drain of the P-type MOS transistor  322  of the P-type MOS transistor latch unit  320 , a source to be connected to the drain of the N-type MOS transistor  344  of the comparative operation control circuit  340 , and a gate for receiving a signal VIM. The on resistance value of the N-type MOS transistor  332  varies according to the potential of the signal VIM. 
     The P-type MOS transistor latch unit  320  includes the P-type MOS transistors  321  and  322 . The N-type MOS transistor latch unit  325  includes the N-type MOS transistors  326  and  327 . 
     The P-type MOS transistor  321  has a drain to be connected to the drain of the N-type MOS transistor  326 , a source to be connected to the high-potential VDD power source  60 , and a gate to be connected to the drain of the P-type MOS transistor  322 . 
     The P-type MOS transistor  322  has a drain to be connected to the drain of the N-type MOS transistor  327 , a source to be connected to the high-potential VDD power source  60 , and a gate to be connected to the drain of the P-type MOS transistor  321 . 
     The N-type MOS transistor  326  has a drain to be connected to the drain of the P-type MOS transistor  321 , a gate to be connected to the drain of the N-type MOS transistor  327 , and a source to be connected to the ground VSS  70 . 
     The N-type MOS transistor  327  has a drain to be connected to the drain of the P-type MOS transistor  322 , a gate to be connected to the drain of the N-type MOS transistor  326 , and a source to be connected to the ground VSS  70 . 
     The output signals OM and OP are output from the N-type MOS transistor latch unit  325 . The output signal OM is connected to the node A between the drain of the N-type MOS transistor  326  and the drain of the P-type MOS transistor  321 . The output signal OP is connected to the node B between the drain of the N-type MOS transistor  327  and the drain of the P-type MOS transistor  322 . 
     The gate of the N-type MOS transistor  326  of the N-type MOS transistor latch unit  325  and the drain of the N-type MOS transistor  327  are connected to the node B, and the gate of the N-type MOS transistor  327  and the drain of the N-type MOS transistor  326  are connected to the node A. That is to say, the N-type MOS transistor  326  and the N-type MOS transistor  327  are connected crosswise to the nodes A and B, and accordingly, the N-type MOS transistor  326  and the N-type MOS transistor  327  amplify potential difference between the nodes A and B. 
     Thus, the P-type MOS transistor  321  of the P-type MOS transistor latch unit  320 , the N-type MOS transistor  331 , and the N-type MOS transistor  343  are serially connected between the high-potential VDD power source  60  and the ground VSS  70 , and make up a first current route including the node A. The P-type MOS transistor  322  of the P-type MOS transistor latch unit  320 , the P-type MOS transistor  332 , and the N-type MOS transistor  344  are serially connected between the high-potential VDD power source  60  and the ground VSS  70 , and make up a second current route including the node B. 
     Therefore, when the signals LM 0 , LM 1 , LP 0 , and LP 1  are “H”, supply of the high-potential VDD to the P-type MOS transistor latch unit  320  and the N-type MOS transistor latch unit  325  is blocked by the P-type MOS transistors  341  and  342 , and the ground potential VSS is supplied to the input unit  330 , and the P-type MOS transistor latch unit  320  by the N-type MOS transistors  343  and  344 . 
     In the above case, the on resistance of the P-type MOS transistor  331  varies according to the potential of the input signal VIP, and accordingly, upon the potential of the input signal VIP decreasing, the potential of the node A increases, and the on resistance of the N-type MOS transistor  332  varies according to the potential of the input signal VIM, and accordingly, upon the potential of the input signal VIM increasing contrary to the input signal VIP, the potential of the node B decreases. On the other hand, conversely, upon the potential of the input signal VIP increasing, the potential of the node A decreases, and upon the potential of the input signal VIM decreasing, the potential of the node B increases. 
     Note that, when the signals LM 0 , LM 1 , LP 0 , and LP 1  are “L”, the high-potential VDD is supplied to the P-type MOS transistor latch unit  320 , and the N-type MOS transistor latch unit  325 , and the input unit  330  by the P-type MOS transistors  341  and  342 , and the supply of the ground VSS to the input unit  330  is blocked by the N-type MOS transistors  343  and  344 . 
     As a result thereof, potential difference between the nodes A and B is 0, or almost eliminated. 
     The comparative operation setting circuit  350  is made up of a circuit similar to the comparative operation setting circuit  50  according to the first embodiment for driving the signals LM 0  and LP 0 , and a circuit similar to the comparative operation setting circuit  50  for driving the signals LM 1  and LP 1 . The circuits similar to the comparative operation setting circuit  50  include a delay circuit  351 , and a logic circuit  356  for controlling the delay circuit  351 . The delay circuit  351  and the logic circuit  356  are the same circuits as the delay circuit  51  and the logic circuit  56 . Now, it goes without saying that delay between the signals LM 0  and LP 0 , and delay between the signals LM 1  and LP 1  may be set to the same delay amount, or may be set separately. It is further needless to say that an arrangement may be made where just one of the delays is set. 
     Thus, the comparison circuit  300  according to the fourth embodiment includes an input unit ( 330 ) made up of a first MOS transistor (N-type MOS transistor  331 ) for receiving a first signal (signal VIP) at the gate electrode, and a second MOS transistor (N-type MOS transistor  332 ) for receiving a second signal (signal VIM) at the gate electrode; a latch circuit (P-type MOS transistor latch circuit  320 , N-type MOS transistor latch circuit  325 ) for amplifying potential difference between a first current route where an electric current is controlled by the first MOS transistor according to the voltage of the first signal, and a second current route where an electric current is controlled by the second MOS transistor according to the voltage of the second signal; a comparative operation setting unit (comparative operation setting circuit  350 ) including a first switch (P-type MOS transistor  341 ) for executing supply or blocking of supply of the high-potential VDD to the drain of the first MOS transistor by a third current route different from the first current route, a second switch (P-type MOS transistor  342 ) for executing supply or blocking of supply of the high-potential VDD to the drain of the second MOS transistor by a fourth current route different from the second current route, and a third switch (N-type MOS transistor  343 ) and a fourth switch (N-type MOS transistors  344 ) which execute supply or blocking of supply of the ground VSS to the first current route and the second current route; and a comparative operation control unit (comparative operation control circuit  340 ) for controlling supply or blocking of the first through fourth switches. 
     Thus, supply or blocking timing of any of the first through fourth switches is controlled, whereby the potentials of the first current route and the second current route before start of comparison can also be controlled. 
     As a result thereof, the potentials of the first current route and the second current route before start of comparison are controlled, whereby comparison error along with different properties relating to the gate voltages of on resistances of the first and second MOS transistors which receive input signals VIP and VIM can be controlled. Note that it goes without saying that comparison error can be controlled by controlling any of the first through fourth switches. 
     Further, the latch circuit portion includes the P-type MOS transistor latch circuit  320  made up of a P-type MOS transistor (P-type MOS transistor  321 ) making up the first current route, and a P-type MOS transistor (P-type MOS transistor  322 ) making up the second current route. Note that supply and blocking of supply of the high-potential VDD to the P-type MOS transistor latch unit  320  are executed by the first switch and the second switch. The N-type MOS transistors making up the N-type MOS transistor latch unit are not included in the first current route and the second current route, and accordingly, capacitance parasitizing the first current route and the second current route decreases, and the response speed of the comparison circuit  300  becomes fast. 
     Fifth Embodiment 
       FIG. 9  is a circuit diagram illustrating a comparison circuit  400  according to the fifth embodiment. The comparison circuit  400  includes a P-type MOS transistor latch unit  420 , an input unit  430 , an N-type MOS transistor latch unit  425 , a comparative operation control circuit  440 , and a comparative operation setting circuit  450 . 
     The comparative operation control circuit  440  includes N-type MOS transistors  443  and  444 , and P-type MOS transistors  441  and  442 . 
     The P-type MOS transistor  441  has a drain to be connected to the source of a P-type MOS transistor  431  of the input unit  430 , a source to be connected to the high-potential VDD power source  60 , and a gate for receiving a signal LM 1 . 
     The P-type MOS transistor  442  has a drain to be connected to the source of a P-type MOS transistor  432  of the input unit  430 , a source to be connected to the high-potential VDD power source  60 , and a gate for receiving a signal LP 1 . 
     The P-type MOS transistor  441  supplies the high-potential VDD from the high-potential VDD power source  60  to the input unit  430  when the logic of the signal LM 1  is “L”, and blocks supply of the high-potential VDD from the high-potential VDD power source  60  to the input unit  430  when the logic is “H”. The P-type MOS transistor  441  serves as a switch for connecting or blocking between the input unit  430  and the high-potential VDD power source  60 . 
     The P-type MOS transistor  442  supplies the high-potential VDD from the high-potential VDD power source  60  to the input unit  430  when the logic of the signal LP 1  is “L”, and blocks supply of the high-potential VDD from the high-potential VDD power source  60  to the input unit  430  when the logic is “H”. The P-type MOS transistor  442  serves as a switch for connecting or blocking between the input unit  430  and the high-potential VDD power source  60 . 
     The N-type MOS transistor  443  has a source to be connected to a ground VSS power source  70 , a drain to be connected to the drain of the P-type MOS transistor  431  of the input unit  430 , and a gate for receiving a signal LM 0 . 
     Note that the drain of the N-type MOS transistor  443 , the drain of the P-type MOS transistor  431  of the input unit  430 , the drain of the P-type MOS transistor  421  of the P-type MOS transistor latch unit  420 , and the drain of the N-type MOS transistor  426  of the N-type MOS transistor latch unit  425  are all connected to the node A within the comparison circuit  400 . The comparison circuit  400  outputs an output signal OM from the node A. 
     The N-type MOS transistor  443  supplies the ground VSS from the ground VSS power source  70  to the input unit  430  when the logic of the signal LM 0  is “H”, and blocks supply of the ground VSS from the ground VSS power source  70  to the input unit  430  when the logic is “L”. The N-type MOS transistor  443  serves as a switch for connecting or blocking between the input unit  430  and the ground VSS power source  70 . 
     The N-type MOS transistor  444  has a source to be connected to the ground VSS power source  70 , a drain to be connected to the drain of the P-type MOS transistor  432  of the input unit  430 , and a gate for receiving a signal LP 0 . 
     Note that the drain of the N-type MOS transistor  444 , the drain of the N-type MOS transistor  432  of the input unit  430 , the drain of the P-type MOS transistor  422  of the P-type MOS transistor latch unit  420 , and the drain of the N-type MOS transistor  427  of the N-type MOS transistor latch unit  425  are all connected to the node B within the comparison circuit  400 . The comparison circuit  400  outputs an output signal OP from the node B. 
     The N-type MOS transistor  444  supplies the ground VSS from the ground VSS power source  70  to the input unit  430  when the logic of the signal LP 0  is “H”, and blocks supply of the ground VSS from the ground VSS power source  70  to the input unit  430  when the logic is “L”. The N-type MOS transistor  444  serves as a switch for connecting or blocking between the input unit  430  and the ground VSS power source  70 . 
     The input unit  430  includes the P-type MOS transistors  431  and  432 . The P-type MOS transistor  431  has a drain to be connected to the drain of the P-type MOS transistor  421  of the P-type MOS transistor latch unit  420 , a source to be connected to the drain of the N-type MOS transistor  443  of the comparative operation control circuit  440 , and a gate for receiving an input signal VIP. The on resistance value of the P-type MOS transistor  431  varies according to the potential of the input signal VIP. 
     The P-type MOS transistor  432  has a drain to be connected to the drain of the P-type MOS transistor  422  of the P-type MOS transistor latch unit  420 , a source to be connected to the drain of the N-type MOS transistor  444  of the comparative operation control circuit  440 , and a gate for receiving a signal VIM. The on resistance value of the N-type MOS transistor  432  varies according to the potential of the signal VIM. 
     The P-type MOS transistor latch unit  420  includes the P-type MOS transistors  421  and  422 . The N-type MOS transistor latch unit  425  includes the N-type MOS transistors  426  and  427 . 
     The P-type MOS transistor  421  has a drain to be connected to the drain of the N-type MOS transistor  426 , a source to be connected to the high-potential VDD power source  60 , and a gate to be connected to the drain of the P-type MOS transistor  422 . 
     The P-type MOS transistor  422  has a drain to be connected to the drain of the N-type MOS transistor  427 , a source to be connected to the high-potential VDD power source  60 , and a gate to be connected to the drain of the P-type MOS transistor  421 . 
     The N-type MOS transistor  426  has a drain to be connected to the drain of the P-type MOS transistor  421 , a gate to be connected to the drain of the N-type MOS transistor  427 , and a source to be connected to the ground VSS  70 . 
     The N-type MOS transistor  427  has a drain to be connected to the drain of the P-type MOS transistor  422 , a gate to be connected to the drain of the N-type MOS transistor  426 , and a source to be connected to the ground VSS  70 . 
     The output signals OM and OP are output from the N-type MOS transistor latch unit  425 . The output signal OM is connected to the node A between the drain of the N-type MOS transistor  426  and the drain of the P-type MOS transistor  421 . The output signal OP is connected to the node B between the drain of the N-type MOS transistor  427  and the drain of the P-type MOS transistor  422 . 
     The gate of the N-type MOS transistor  426  of the N-type MOS transistor latch unit  425  and the drain of the N-type MOS transistor  427  are connected to the node B, and the gate of the N-type MOS transistor  427  and the drain of the N-type MOS transistor  426  are connected to the node A. That is to say, the N-type MOS transistor  426  and the N-type MOS transistor  427  are connected crosswise to the nodes A and B, and accordingly, the N-type MOS transistor  426  and the N-type MOS transistor  427  amplify potential difference between the nodes A and B. 
     Thus, the N-type MOS transistor  426  of the N-type MOS transistor latch unit  425 , the P-type MOS transistor  431 , and the P-type MOS transistor  441  are serially connected between the high-potential VDD power source  60  and the ground VSS  70 , and make up a first current route including the node A. The N-type MOS transistor  427  of the N-type MOS transistor latch unit  425 , the P-type MOS transistor  432 , and the P-type MOS transistor  442  are serially connected between the high-potential VDD power source  60  and the ground VSS  70 , and make up a second current route including the node B. 
     Therefore, when the signals LM 0 , LM 1 , LP 0 , and LP 1  are “H”, supply of the high-potential VDD to the input unit  430  is blocked by the P-type MOS transistors  441  and  442 , and the ground VSS is supplied to the input unit  430 , the P-type MOS transistor latch unit  420 , and the N-type MOS transistor latch unit  425  by the N-type MOS transistors  443  and  444 . 
     As a result thereof, potential difference between the nodes A and B is 0, or almost eliminated. 
     On the other hand, when the signals LM 0 , LM 1 , LP 0 , and LP 1  are “L”, the high-potential VDD is supplied to the input unit  430  by the P-type MOS transistors  441  and  442 , and the supply of the ground VSS to the P-type MOS transistor latch unit  420 , the N-type MOS transistor latch unit  425 , and the input unit  430  is blocked by the N-type MOS transistors  443  and  444 . 
     In the above case, the on resistance of the P-type MOS transistor  431  varies according to the potential of the input signal VIP, and accordingly, upon the potential of the input signal VIP decreasing, the potential of the node A increases, and the on resistance of the N-type MOS transistor  432  varies according to the potential of the input signal VIM, and accordingly, upon the potential of the input signal VIM increasing contrary to the input signal VIP, the potential of the node B decreases. 
     On the other hand, conversely, upon the potential of the input signal VIP increasing, the potential of the node A decreases, and upon the potential of the input signal VIM decreasing, the potential of the node B increases. 
     The comparative operation setting circuit  450  is made up of a circuit similar to the comparative operation setting circuit  50  according to the first embodiment for driving the signals LM 0  and LP 0 , and a circuit similar to the comparative operation setting circuit  50  for driving the signals LM 1  and LP 1 . The circuits similar to the comparative operation setting circuit  50  include a delay circuit  451 , and a logic circuit  456  for controlling the delay circuit  451 . The delay circuit  451  and the logic circuit  456  are the same circuits as the delay circuit  51  and the logic circuit  56 . Now, it goes without saying that delay between the signals LM 0  and LP 0 , and delay between the signals LM 1  and LP 1  may be set to the same delay amount, or may be set separately, and further, of the above delays, any one alone may be set. 
     Thus, the comparison circuit  400  according to the fifth embodiment includes an input unit ( 430 ) made up of a first MOS transistor (P-type MOS transistor  431 ) for receiving a first signal (signal VIP) at the gate electrode, and a second MOS transistor (P-type MOS transistor  432 ) for receiving a second signal (signal VIM) at the gate electrode; a latch circuit (P-type MOS transistor latch circuit  420 , N-type MOS transistor latch circuit  425 ) for amplifying potential difference between a first current route where an electric current is controlled by the first MOS transistor according to the voltage of the first signal, and a second current route where an electric current is controlled by the second MOS transistor according to the voltage of the second signal; a comparative operation setting unit (comparative operation setting circuit  450 ) including a first switch (P-type MOS transistor  441 ) for executing supply or blocking of supply of the high-potential VDD to the drain of the first MOS transistor by a third current route different from the first current route, a second switch (P-type MOS transistor  442 ) for executing supply or blocking of supply of the high-potential VDD to the drain of the second MOS transistor by a fourth current route different from the second current route, and a third switch (N-type MOS transistor  443 ) and a fourth switch (N-type MOS transistor  444 ) which execute supply or blocking of supply of the ground VSS to the first current route and the second current route; and a comparative operation control unit (comparative operation control circuit  440 ) for controlling supply or blocking of the first through fourth switches. 
     Thus, supply or blocking timing of any of the first through fourth switches is controlled, whereby the potentials of the first current route and the second current route before start of comparison can also be controlled. 
     As a result thereof, the potentials of the first current route and the second current route before start of comparison are controlled, whereby comparison error along with different properties relating to the gate voltages of on resistances of the first and second MOS transistors which receive input signals VIP and VIM can be controlled. Note that it goes without saying that comparison error can be controlled by controlling any of the first through fourth switches. 
     Further, the latch circuit includes the N-type MOS transistor latch unit  425  made up of an N-type MOS transistor (N-type MOS transistor latch circuit  426 ) making up the first current route, and an N-type MOS transistor (N-type MOS transistor  427 ) making up the second current route. Note that supply and blocking of supply of the ground VSS to the N-type MOS transistor latch unit  425  is executed by the first and second switches. The P-type MOS transistors making up the P-type MOS transistor latch unit are not included in the first current route and the second current route, and accordingly, capacitance parasitizing the first current route and the second current route decreases, and the response speed of the comparison circuit  400  becomes fast. 
     Sixth Embodiment 
       FIG. 10  illustrates an ADC (Analog Digital Converter)  500  according to the sixth embodiment. The ADC  500  according to the sixth embodiment includes the latch unit  20 , input unit  30 , and the comparative operation control circuit  40 , of the comparison circuit  10  according to the first embodiment, and further includes a delay circuit  520 , a successive comparative operation control circuit  530 , and a sample-hold circuit  540 . 
     The ADC  500  according to the sixth embodiment is a successive comparative type analog-to-digital conversion circuit using the latch unit  20 , input unit  30 , and comparative operation control circuit  40 , described in the first embodiment. 
     Accordingly, detailed description will be omitted regarding the latch unit  20 , input unit  30 , and comparative operation control circuit  40 , of the ADC  500  according to the sixth embodiment. 
     The input unit  30  receives a complementary input signal Vi, and generates an inversion complementary signal having inversion logic. The latch unit  20  receives and latches the inversion complementary signal thereof. The comparison control circuit  40  receives the signal from the successive comparative operation control circuit  530 , and connects or disconnects the input unit  30  and the latch unit  20  to or from the high-potential power source Vcc according to the logic thereof. The latch unit  20  and the input unit  30  make up a comparator for comparing the potential of a signal Vi+ and the potential of a signal Vi− which make up the complementary input signal Vi. 
     Note that though the latch unit  20 , the input unit  30 , and the comparative operation control circuit  40  operate according to a signal CNTL 531  output from the successive comparative operation control circuit  530 , details regarding the operation thereof will be described with reference to  FIG. 12 . 
     The delay circuit  520  is configured of inverters  521 ,  522 ,  526 , and  527 , inverters  523  and  525  using variable signal delay time, and an inverter  524  which receives a signal CLK at the input terminal. 
     The inverter  524  receivers the signal CLK, and outputs an output signal A to a switch  43   a  included in the comparative operation control circuit  40  to connect or disconnect the ground potential and the input unit  30  according to the logic of the output signal A. 
     The inverters  523  and  525  receive the output signal A of the inverter  524  at the input terminals thereof. The inverter  522  receives the output signal of the inverter  523  at the input terminal thereof. The inverter  521  receives the output signal of the inverter  522  at the input terminal thereof. The inverter  522  outputs a signal Axp to one switch  41   a  of the comparative operation control circuit  40 . Note that the one switch  41   a  connects or disconnects the input unit  30  and the latch unit  20  from the high-potential power source according to the logic of the signal Axp. With the inverter  523 , signal delay time from the signal having been received at the input terminal until the output signal is output varies according to the signal CNTL 531  made up of multiple digital signals. Examples of the inverter  523  will be shown in  FIGS. 11A and 11B . 
     The inverter  526  receives the output signal of the inverter  525  at the input terminal thereof. The inverter  527  receives the output signal of the inverter  526  at the input terminal thereof. The inverter  527  outputs a signal Axm to the other switch  42   a  of the comparative operation control circuit  40 . Note that the other switch  42   a  connects or disconnects the input unit  30  and the latch unit  20  from the high-potential power source according to the logic of the signal Axm. With the inverter  525 , signal delay time from the signal having been received at the input terminal until the output signal is output varies according to the signal CNTL 531  made up of multiple digital signals. Examples of the inverter  525  will be shown in  FIGS. 11A and 11B . 
     Accordingly, with the binary numeral represented by the signal CNTL 531 , time difference can be set to between the output period of the signal Axm and the output period of the signal Axp. 
     The sample-hold circuit  540  includes a switch  541  for connecting or disconnecting an input terminal which receives one signal Vi+ making up the input signal Vin thereof, and a signal line to be connected to the input unit  30  according to the logic of the signal CLK, and a capacitance  542  to be connected to this signal line. Also, the sample-hold circuit  540  includes a switch  543  for connecting or disconnecting an input terminal which receives the other signal Vi− making up the input signal Vin thereof, and a signal line to be connected to the input unit  30  according to the logic of a control signal CN 532  from the successive comparative operation control circuit  530 , and a capacitance  544  to be connected to this signal line. 
     Accordingly, the sample-hold circuit  540  is a circuit for receiving the complementary input signal Vin, and sampling the voltage of one signal Vi+ making up the input signal Vin thereof, and the voltage of the complementary signal Vi− thereof. 
     The successive comparative operation control circuit  530  receives signals Vo+ and Vo− from the latch unit  20 , and the signal CLK, outputs the signal CNTL 531  to the delay circuit  520 , and outputs the control signal CN 532  to the sample-hold circuit  540 . Note that the operation of the successive comparative operation control circuit  530 , and the signal CNTL 531  will be described in detail with reference to  FIGS. 11 through 16 . Also, the control signal CN 532  will be described in detail with reference to  FIG. 15 . 
       FIGS. 11A and 11B  illustrate examples of the inverters  523  and  525  of the delay circuit  520 .  FIG. 11A  illustrates an inverter  528  which sets delay time from input of an input signal until output of an output signal to be variable using multiple combination circuits made up of a resistor, and a switch for connecting or disconnecting the resistor thereof to or from the ground line. The inverter  528  is configured of n combination circuits from a combination circuit  528   r   1  to a combination circuit  528   rn  (n is a positive integer) made up of a P-type transistor  528   a , an N-type transistor  528   b , a resistor, and a switch, an NOR  528   c , an input terminal  528   d , and an output terminal  528   e.    
     The P-type transistor  528   a  is a MOS transistor wherein the source thereof is connected to the high-potential power source Vcc, the drain thereof is connected to the output terminal  528   e , and the gate thereof is connected to the input terminal  528   d.    
     The P-type transistor  528   b  is a MOS transistor wherein the source thereof is connected to the n combination circuits from the combination circuit  528   r   1  to the combination circuit  528   rn , the drain thereof is connected to the output terminal  528   e , and the gate thereof is connected to the input terminal  528   d.    
     The n combination circuits from the combination circuit  528   r   1  to the combination circuit  528   rn  are connected between the source of the N-type transistor  528   b  and the ground Vss in parallel. Each of the combination circuits is configured by a resistor, a first switch, and a second switch being connected serially, and one end of the resistor is connected to the source of the N-type transistor  528   b , and the other end of the second switch is connected to the ground Vss. 
     If we say that the resistance value of the resistor of the combination circuit  528   r   1  is 1, the resistance value of the resistor of the combination circuit  528   rn  is the n&#39;th power of 2. The signal CNTL 531  is multiple signals representing n+2 binary codes, and signals representing the first bit to the n&#39;th bit of the binary numeral that the signal CNTL 531  represents are connected to the first switches from the combination circuit  528   r   1  to the combination circuit  528   rn . The first switches from the combination circuit  528   r   1  to the combination circuit  528   rn  are connected when the signals of the signal CNTL 531  are “1”, and are disconnected when the signals of the signal CNTL 531  have an inverse logic. 
     The signal corresponding to the n+1&#39;th bit of the binary numeral that the signal CNTL 531  represents is commonly connected to all the second switches from the combination circuit  528   r   1  to the combination circuit  528   rn . If we say that the second switch of the inverter  528  operating as the inverter  523  operates to be connected when the logic of the signal thereof is “1”, and operates to be disconnected when the logic is “0”, the second switch of the inverter  528  operating as the inverter  525  operates to be connected when the logic of the signal thereof is “0”, and operates to be disconnected when the logic is “1”. 
     Also, the signal corresponding to the n+1&#39;th bit of the binary numeral that the signal CNTL 531  represents is connected to one input terminal of the NOR  528   c . The NOR  528   c  inverts the logic of the signal thereof, and outputs the output signal thereof to the switch  528   f . The switch  528   f  connects or disconnects the source of the N-type transistor  528   b  to or from the ground Vss according to the logic of the output signal from the NOR  528   c.    
     The signal corresponding to the n+2&#39;th bit of the binary numeral that the signal CNTL 531  represents is connected to the other input terminal of the NOR  528   c . The signal corresponding to the n+2&#39;th bit of the signal CNTL 531  is used when both of the signal Axp and the signal Axm are output without delay from the signal A. 
     Thus, with the inverter  528 , operation that makes delay time from input of an input signal to output of an output signal variable according to the signal representing the n+1&#39;th bit of the signal CNTL 531 , and operation that makes the delay time constant are switched. With the inverter  528 , in the case of making the delay time variable, the magnitude of the delay time increases according to the magnitude of the binary numeral represented by from bit  1  to bit n of the signal CNTL 531 . This is because the greater the magnitude of the binary numeral represented by from bit  1  to bit n of the CNTL 531  is, the smaller the electric current from the ground Vss flowing to the source of the N-type transistor  528   b  is. 
       FIG. 11B  illustrates an inverter  529  which makes the delay time from input of an input signal to output of an output signal variable using multiple combination circuits made up of a resistor, and a switch for connecting or disconnecting the capacitance thereof to or from a ground line. The inverter  529  is configured of a P-type transistor  529   a , an N-type transistor  529   b , n combination circuits from a combination circuit  529   c   1  to a combination circuit  529   cn  (n is a positive integer) which are made up of capacitance and a switch, an input terminal  529   d , a NOR  529   f , and an output terminal  529   e.    
     The P-type transistor  529   a  is a MOS transistor wherein the source thereof is connected to the high-potential power source Vcc, the drain thereof is connected to the output terminal  529   e , and the gate thereof is connected to the input terminal  529   d.    
     The N-type transistor  529   b  is a MOS transistor wherein the source thereof is connected to the ground Vss, the drain thereof is connected to the output terminal  529   e , and the gate thereof is connected to the input terminal  528   d.    
     The n combination circuits from the combination circuit  529   c   1  to the combination circuit  529   cn  (n is a positive integer) are connected between the output terminal  529   e  and the ground Vss in parallel. Each of the combination circuits is configured by capacitance, a first switch, and a second switch being serially connected, and one end of the capacitance is connected to the output terminal  529   e , and the other end of the second switch is connected to the ground Vss. 
     If we say that the capacitance value of the capacitance of the combination circuit  529   c   1  is 1, the capacitance value of the capacitance of the combination circuit  529   cn  is the n&#39;th power of 2. The signal CNTL 531  is multiple signals representing binary numeral of n+2 bits, and signals representing the first bit to the n&#39;th bit of the binary numeral that the signal CNTL 531  represents are connected to the first switches from the combination circuit  529   c   1  to the combination circuit  529   cn . The first switches from the combination circuit  529   c   1  to the combination circuit  529   cn  are connected when the signals of the signal CNTL 531  are “1”, and are disconnected when the signals of the signal CNTL 531  have an inverse logic. 
     The signal corresponding to the n+1&#39;th bit of the binary numeral that the signal CNTL 531  represents is connected to one terminal of the NOR  529   f , and the output of the NOR  529   f  is commonly connected to all the second switches from the combination circuit  529   c   1  to the combination circuit  529   cn . If we say that the second switch of the inverter  529  operating as the inverter  523  operates to be connected when the logic of the signal thereof is “1”, and operates to be disconnected when the logic is “0”, the second switch of the inverter  529  operating as the inverter  525  operates to be connected when the logic of the signal thereof is “0”, and operates to be disconnected when the logic is “1”. 
     The signal corresponding to the n+2&#39;th bit of the binary numeral that the signal CNTL 531  represents is connected to the other input terminal of the NOR  529   f . The signal corresponding to the n+2&#39;th bit of the signal CNTL 531  is used when both of the signal Axp and the signal Axm are output without delay from the signal A. 
     Thus, with the inverter  529 , operation that makes delay time from input of an input signal to output of an output signal variable according to the signal representing the most significant bit of the signal CNTL 531 , and operation that makes the delay time constant are switched. With the inverter  529 , in the case of making the delay time variable, the magnitude of the delay time increases according to the magnitude of the binary numeral represented by bit  1  to bit n of the signal CNTL 531 . This is because the greater the magnitude of the binary numeral represented by bit  1  to bit n of the CNTL 531  is, the greater the capacitance value of the capacitance to be connected to the output terminal  529   e  of the N-type transistor is. 
       FIGS. 12A and 12B  are diagrams for describing the operation according to the sixth embodiment regarding the latch unit  20 , input unit  30 , comparative operation control circuit  40 , and delay circuit  520 . 
     With the first embodiment, the latch unit  20 , input unit  30 , comparative operation control circuit  40 , and delay circuit  520  make up a comparator which operates so as to compare the magnitude between the potential of the signal Vi+ and the potential of the signal Vi− making up the complementary input signal Vi input to the input unit  30 . 
     However, with the sixth embodiment, operation for comparing the magnitude of the potential of an input signal with the first embodiment is a basic operation, and further, this basic operation is repeated while shifting the leading or trailing-edge period of the signals Axp and Axm, and accordingly, the latch unit  20 , input unit  30 , comparative operation control circuit  40 , and delay circuit  520  operate as a circuit for detecting the complementary input signal input to the input unit  30 , i.e., the potential difference of the complementary signal made up of the signal Vi+ and the signal Vi−. 
       FIG. 12A  is a diagram illustrating the potential waveforms of the principal signals of the latch unit  20 , input unit  30 , comparative operation control circuit  40 , and delay circuit  520 , corresponding to the operation for shifting the trailing-edge period of the signals Axp and Axm. 
     The successive comparative operation control circuit  530  outputs the signal CNTL 5312  having a logic representing time difference between the leading or trailing-edge periods of the signals Axp and Axm during point-in-time T 1  through point-in-time T 5  to cause the latch unit  20 , input unit  30 , comparative operation control circuit  40 , and delay circuit  520  to execute one-time basic operation. Such as described in  FIGS. 11A and 11B , whether the signal Axm or signal Axp is delayed is determined according to the logic of the most significant bit of the signal CNTL 531 .  FIG. 12A  illustrates an example wherein the signal Axm is delayed. 
     Thus, the delay circuit  520  delays and outputs the signal A and the signal A synchronized with the signal CLK, and also outputs the signal Axm according to the logic of the signal CNTL 531 . The leading edge of the signal A is point-in-time T 2 , and the trailing edges thereof are point-in-time T 1  and point-in-time T 5 . The leading edges of the signal Axp are point-in-time T 1  and point-in-time T 5 , and the trailing edges thereof are point-in-time T 3 . Also, the leading edge of the leading edges of the signal Axm are point-in-time T 1  and point-in-time T 5 , and the trailing edge thereof is point-in-time T 3 . Here, time difference between the point-in-time T 2  and point-in-time T 3  indicates trailing-edge time difference between the signals Axp and Axm. 
     Note that  FIG. 12A  illustrates the operation of each circuit for one cycle of the signal CLK, and illustrates the operation of each circuit in the case that the trailing-edge time difference between the signals Axp and Axm represented by the signal CNTL is 0.21 ns. However, before the point-in-time T 1 , the successive comparative operation control circuit  530  increases the trailing-edge time difference between the signals Axp and Axm represented by the logic of the signal CNTL in increments of 0.01 ns for each two cycle. 
     The signal Vi to be input to the input unit  30  is a complementary signal, and in  FIG. 12A , the difference between the signal Vi+ and signal Vi− making up the complementary input signal Vi is kept in 100 mv. 
     At the point-in-time T 1 , upon the logic of the signal A turning to “L”, and the logics of the signals Axp and Axm turning to “H”, the switch of the comparative operation control circuit  40  connects the latch unit  20  and the high-potential power source Vcc, and connects the input unit  30  and the ground Vss, and accordingly, the logics of the signals Vp and Vm latched by the latch unit  20  both turn to “H”. 
     Subsequently, at the point-in-time T 2 , upon the logic of the signal A turning to “H”, and the logic of the signal Axp turning to “L”, the logic of the signal Vm begins to operate toward “L”. On the other hand, at the point-in-time T 3 , the logic of the signal Axm also turns to “L”, and the logic of the signal Vp also begins to operate toward “L”. Here, the potential of the signal Vi− is lower than the potential of the signal Vi+, and accordingly, speed wherein the logic of the signal Vp proceeds to “L” is faster than speed wherein the logic of the signal Vm proceeds to “L”. However, point-in-time when the logic of the signal Vp begins to proceed to “L” is T 3  that is slower than T 2 , and accordingly, the potential of the signal Vm first exceeds the operation threshold of the latch unit  20 . As a result thereof, at point-in-time T 4 , according to the operation of the latch unit  20 , the logic of the signal Vm turns to “L”, and the logic of the signal Vp turns to “H”. As a result thereof, the logic of an output signal Vo changes at the point-in-time T 3 . 
     Subsequently, at point-in-time T 5 , upon the logic of the signal A turning to “L”, and the logics of the signals Axp and Axm turning to “H” again, the latch unit  20 , input unit  30 , comparative operation control circuit  40 , and delay circuit  520  return to the state at the point-in-time again. 
       FIG. 12B  is a diagram representing change in the signal Vm when repeating the operation from the point-in-time T 1  to point-in-time T 5  while changing trailing-edge time difference td between the signals Axp and Axm from 0.16 ns to 0.25 ns. 
     In  FIG. 12B , the vertical axis represents potentials, and the horizontal axis indicates elapse of time. Thin solid lines indicate the trailing-edge point-in-time of the signal CLK. Also, heavy solid lines indicate change in the potential of the signal Vm. 
     For example, when the time difference td of the trailing edges of the signals Axp and Axm represented by the logic of the signal CNTL is 0.2 ns, this indicates that the potential of the signal Vm has changed between 3 V and 1.8 V. 
     Also, when the time difference td is 0.21 ns, this indicates that the potential of the signal Vm has changed between 3 V and 0 V. 
     In the event that the time difference td is equal to or smaller than 0.20 ns, this indicates that the logic of the signal Vm latched by the latch unit  20  is “H” after comparative operation. Also, upon the time difference td reaching equal to or greater than 0.21 ns, this indicates that the logic of the signal Vm latched by the latch unit  20  is “L” after comparative operation. That is to say, the logic of the signal Vm or signal Vp after latch changes with time difference td=0.21 ns as a border. 
     Thus, in the case that there is potential difference between the signal Vi− and signal Vi+, upon observing change in the logic of the signal Vm or signal Vp latched by the latch unit  20  while changing the time difference td of the trailing edges of the signals Axp and Axm represented by the logic of the signal CNTL, the time difference td to cause change in the logic of the signal Vm or signal Vp as to the above time difference td after comparative operation can be detected. 
     Therefore, it can be found that relationship of the potential difference between the signal Vi+ and signal Vi+ making up the input signal Vi, and the time difference td can be obtained beforehand. 
     Further, relationship of the logic of the signal CNTL, and the time difference td of the trailing edges of the signals Axp and Axm is uniquely determined such as described in  FIGS. 11A and 11B , and accordingly, relationship of the potential difference between the signal Vi− and signal Vi+, and the logic of the signal CNTL is uniquely determined. 
       FIG. 13  is a diagram representing relationship of the logic of the signal CNTL, the time difference td of the trailing edges of the signal Axp and signal Axm, the potential difference between the signal Vi− and signal Vi+ making up the input signal Vi. 
     In  FIG. 13 , the horizontal axis on the lower side represents the time difference td, the vertical axis represents the potential difference between the signal Vi− and signal Vi+ making up the input signal Vi, and the horizontal axis on the upper side indicates the binary numeral that the signal CNTL  531  corresponding to the time difference td represents. 
     According to  FIG. 13 , the relationship of the potential difference between the signal Vi− and signal Vi+, and the time difference td of the trailing edges of the signals Axp and Axm used for changing the logic of the signal Vm or Vp indicates monotone increase. In more detail, for example, if we represent this relationship such as (potential difference, td), relationship of (0.01, 0.15), (0.05, 0.17), (0.1, 0.2), (0.2, 0.23), (0.3, 0.27), (0.4, 0.3), (0.5, 0.33), (0.6, 0.37), (0.7, 0.39), and (0.8, 0.41) is shown. Note that relationship of “the time difference td” and “the potential difference between the signal Vi− and signal Vi+” can be obtained by circuit simulation based on the circuit diagram of the latch unit  20 , input unit  30 , comparative operation control circuit  40 , and delay circuit  520  shown in  FIG. 10 . 
     The binary numerals shown in the upper portion of the drawing are disposed in the positions corresponding to “the time difference td of the trailing edges of the signals Axp and Axm” obtained when the signal CNTL 531  representing the binary numerals thereof is input. The upper stage of the binary numerals shown in the upper portion of the drawing indicates binary numerals used for delaying only the trailing-edge period of the signal Axm with the trailing-edge period of the signal Axp being fixed as to the trailing-edge period of the signal A. On the other hand, the binary numerals on the lower stage indicate binary numerals used for delaying only the trailing-edge period of the signal Axp with the trailing-edge period of the signal Axm being fixed as to the trailing-edge period of the signal A. 
     If we describe the correspondence as (binary, tpd), for example, this yields correspondences of (00000, 0.12 ns), (00001, 0.12 ns), (10000, 0.14 ns), (10001, 0.14 ns), (01000, 0.16 ns), (01001, 0.16 ns), (11000, 0.18 ns), (11001, 0.18 ns), (00100, 0.20 ns), (00101, 0.20 ns), (10100, 0.22 ns), (10101, 0.22 ns), (01100, 0.24 ns), (01101, 0.24 ns), (11100, 0.26 ns), (11101, 0.26 ns), (00010, 0.28 ns), (00011, 0.28 ns), (10010, 0.30 ns), (10011, 0.30 ns), (01010, 0.32 ns), (01011, 0.32 ns), (11010, 0.34 ns), (11011, 0.34 ns), (00110, 0.36 ns), (00111, 0.36 ns), (10110, 0.38 ns), (10111, 0.38 ns), (01110, 0.40 ns), (01111, 0.40 ns), (11110, 0.42 ns), and (11111, 0.42 ns). 
     Note that the above correspondences can be obtained by executing circuit simulation regarding the inverters  528  and  529  shown in  FIGS. 11A and 11B . 
       FIG. 14  illustrates an operation waveform when detecting the potential difference between the complementary signal made up of the signal Vi+ and signal Vi− by the ADC circuit  500  according to the sixth embodiment. 
     The signal CLK is a clock signal which repeats logic “H” and “L” in a constant cycle. The control signal CN 532  is a control signal to be output from the successive comparative operation control circuit  530 . When the logic of the control signal CN 532  of the sample-hold circuit  540  is “H”, i.e., during from the point-in-time T 1  to the point-in-time T 3 , when the switches  541  and  543  connect the input terminal and the capacitances  542  and  544 , sample the potentials of the signal Vi+ and signal Vi−, and the logic of the control signal CNT 532  is “L”, i.e., during from the point-in-time T 3  to the point-in-time T 8 , and during from the point-in-time T 10  to the point-in-time T 12 , the potentials of the signal Vi+ and signal Vi− are held. 
     The signal A is a logic inversion signal of the signal CLK synchronized with the signal CLK, to be output from the inverter  524  which received the signal CLK. The signal Axp is output from the inverter  524  which received the signal CLK to the switch  41   a  of the comparative control circuit  40  via the inverters  523 ,  522 , and  521 . The signal Axm is output from the inverter  524  which received the signal CLK to the switch  42   a  of the comparative control circuit  40  via the inverters  525 ,  526 , and  527 . 
     If we say that signals making up the complementary signal Vo are one signal Vo+ and the other signal Vo−, these signals have the same voltage when the signal A falls. On the other hand, when the signal A rises, the latch unit  20  latches the logic of the signal Vm and the logic of the signal Vp, and outputs these to the successive comparative operation control circuit  530  as signals Vo+ and Vo−. 
     The signal CNTL 531  is output from the successive comparative operation control circuit  530  to the inverter  523  or inverter  525  in sync with the signal CLK (i.e., from the input signal Vi having been sampled and held until point-in-time T 12  from point-in-time T 3  for each point-in-time). The signal CNTL 531  represents a binary numeral determining td=|Axp−Axm|. Accordingly, the latch unit  20 , input unit  30 , comparative operation control circuit  40 , and delay circuit  520  executes comparison between the potential of the signal Vi+ and the potential of the signal Vi− using td=|Axp−Axm| controlled by the successive comparative operation control circuit  530 . Note that description will be made regarding a method for detecting and digitizing difference between the potential of the signal Vi+ and the potential of the signal Vi− according to the control of the successive comparative operation control circuit  530 , and control thereof, with reference to the flowchart in  FIG. 15 . 
     After the operations of a series of the latch unit  20 , input unit  30 , comparative operation control circuit  40 , and delay circuit  520 , a digital signal Dout representing the difference between the potential of the signal Vi+ and the potential of the signal Vi− detected by the successive comparative operation control circuit  530  is output by the successive comparative operation control circuit  530  at the point-in-time T 2  and point-in-time T 9 . 
       FIG. 15  is a flowchart for describing the control of td=|Axp−Axm| executed by the successive comparative operation control circuit  530 , and a method for detecting the difference between the potential of the signal Vi+ and the potential of the signal Vi− executed by the control thereof. 
     s 600 : The successive comparative operation control circuit  530  outputs a control signal  532  to the sample-hold circuit  540 , and samples and holds the input signal Vi. Next, the flow proceeds to s 605 . 
     s 605 : The successive comparative operation control circuit  530  outputs the signal CNTL 531  to the delay circuit  520 , and sets td=|Axp−Axm|=0. That is to say, delays of the signals Axp and Axm from the signal A are 0 ns. Next, the flow proceeds to s 610 . 
     s 610 : The delay circuit  520  executes signal output operation for receiving the signal CLK, and outputting the signal A, signal Axp, and signal Axm. Subsequently, the successive comparative operation control circuit  530  receives the signal Vo+ and signal Vo− output from the latch unit  20 , and when the logic of the signal Vo+ is “H”, determines that the potential of the signal Vi+ is higher than the potential of the signal Vi−, sets the most significant bit to “1”, and the flow proceeds to s 615 . On the other hand, when the logic of the signal Vo+ is “L”, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is lower than the potential of the signal Vi−, sets the most significant bit to “0”, and the flow proceeds to s 685 . 
     s 615 : The successive comparative operation control circuit  530  outputs the signal CNTL 531  to the delay circuit  520 , sets delay of the signal Axp from the signal A to 0 ns, sets delay of the signal Axm from the signal A to 0.3 ns, and sets td=|Axp−Axm|=0.3 ns. Next, the flow proceeds to s 620 . 
     s 620 : The delay circuit  520  executes signal output operation similar to s 610 . Next, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is higher than the potential +0.4 V of the signal Vi− when the logic of the signal Vo+ is “H”, sets the second bit to “1”, and the flow proceeds to s 625 . On the other hand, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is lower than the potential +0.4 V of the signal Vi− when the logic of the signal Vo+ is “L”, sets the second bit to “0”, and the flow proceeds to s 630 . 
     s 625 : The successive comparative operation control circuit  530  outputs the signal CNTL 531  to the delay circuit  520 , sets delay of the signal Axp from the signal A to 0 ns, sets delay of the signal Axm from the signal A to 0.36 ns, and sets td=|Axp−Axm|=0.36 ns. Next, the flow proceeds to s 635 . 
     s 635 : The delay circuit  520  executes signal output operation similar to s 610 . Next, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is higher than the potential +0.6 V of the signal Vi− when the logic of the signal Vo+ is “H”, sets the third bit to “1”, and the flow proceeds to s 645 . On the other hand, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is lower than the potential +0.6 V of the signal Vi− when the logic of the signal Vo+ is “L”, sets the third bit to “0”, and the flow proceeds to s 650 . 
     s 645 : The successive comparative operation control circuit  530  outputs the signal CNTL 531  to the delay circuit  520 , sets delay of the signal Axp from the signal A to 0 ns, sets delay of the signal Axm from the signal A to 0.38 ns, and sets td=|Axp−Axm|=0.38 ns. Next, the flow proceeds to s 665 . 
     s 665 : The delay circuit  520  executes signal output operation similar to s 610 . Next, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is higher than the potential +0.7 V of the signal Vi− when the logic of the signal Vo+ is “H”, sets the fourth bit to “1”, and outputs the digital signal Dout representing a binary numeral (1111). On the other hand, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is lower than the potential +0.7 V of the signal Vi− when the logic of the signal Vo+ is “L”, sets the fourth bit to “0”, and outputs the digital signal Dout representing a binary numeral (1110). 
     s 650 : The successive comparative operation control circuit  530  outputs the signal CNTL 531  to the delay circuit  520 , sets delay of the signal Axp from the signal A to 0 ns, sets delay of the signal Axm from the signal A to 0.34 ns, and sets td=|Axp−Axm|=0.34 ns. Next, the flow proceeds to s 670 . 
     s 670 : The delay circuit  520  executes signal output operation similar to s 610 . Next, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is higher than the potential +0.5 V of the signal Vi− when the logic of the signal Vo+ is “H”, sets the fourth bit to “1”, and outputs the digital signal Dout representing a binary numeral (1101). On the other hand, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is lower than the potential +0.5 V of the signal Vi− when the logic of the signal Vo+ is “L”, sets the fourth bit to “0”, and outputs the digital signal Dout representing a binary numeral (1100). 
     s 630 : The successive comparative operation control circuit  530  outputs the signal CNTL 531  to the delay circuit  520 , sets delay of the signal Axp from the signal A to 0 ns, sets delay of the signal Axm from the signal A to 0.22 ns, and sets td=|Axp−Axm|=0.22 ns. Next, the flow proceeds to s 640 . 
     s 640 : The delay circuit  520  executes signal output operation similar to s 610 . Next, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is higher than the potential +0.2 V of the signal Vi− when the logic of the signal Vo+ is “H”, sets the third bit to “1”, and the flow proceeds to s 660 . On the other hand, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is lower than the potential +0.2 V of the signal Vi− when the logic of the signal Vo+ is “L”, sets the third bit to “0”, and the flow proceeds to s 655 . 
     s 660 : The successive comparative operation control circuit  530  outputs the signal CNTL 531  to the delay circuit  520 , sets delay of the signal Axp from the signal A to 0 ns, sets delay of the signal Axm from the signal A to 0.26 ns, and sets td=|Axp−Axm|=0.26 ns. Next, the flow proceeds to s 680 . 
     s 680 : The delay circuit  520  executes signal output operation similar to s 610 . Next, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is higher than the potential +0.3 V of the signal Vi− when the logic of the signal Vo+ is “H”, sets the fourth bit to “1”, and outputs the digital signal Dout representing a binary numeral (1011). On the other hand, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is lower than the potential +0.3 V of the signal Vi− when the logic of the signal Vo+ is “L”, sets the fourth bit to “0”, and outputs the digital signal Dout representing a binary numeral (1010). 
     s 655 : The successive comparative operation control circuit  530  outputs the signal CNTL 531  to the delay circuit  520 , sets delay of the signal Axp from the signal A to 0 ns, sets delay of the signal Axm from the signal A to 0.18 ns, and sets td=|Axp−Axm|=0.18 ns. Next, the flow proceeds to s 675 . 
     s 675 : The delay circuit  520  executes signal output operation similar to s 610 . Next, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is higher than the potential +0.1 V of the signal Vi− when the logic of the signal Vo+ is “H”, sets the fourth bit to “1”, and outputs the digital signal Dout representing a binary numeral (1001). On the other hand, the successive comparative operation control circuit  530  determines that the potential of the signal Vi+ is lower than the potential +0.1 V of the signal Vi− when the logic of the signal Vo+ is “L”, sets the fourth bit to “0”, and outputs the digital signal Dout representing a binary numeral (1000). 
     s 685 : The successive comparative operation control circuit  530  executes the same operation as with s 615 , and sets delay of the signal Axp from the signal A to 0.3 ns. Next, the flow proceeds to s 690 . 
     s 690 : The successive comparative operation control circuit  530  and the delay circuit  520  execute the same operation as with s 620 , the successive comparative operation control circuit  530  determines that the potential of the signal Vi− is higher than the potential +0.4 V of the signal Vi+ when the logic of the signal Vo+ is “L”, sets the second bit to “0”, and the flow proceeds to s 695 . On the other hand, the successive comparative operation control circuit  530  determines that the potential of the signal Vi− is lower than the potential +0.4 V of the signal Vi+ when the logic of the signal Vo+ is “H”, sets the second bit to “1”, and the flow proceeds to s 700 . 
     s 695 : The successive comparative operation control circuit  530  executes the same operation as with s 625 , and sets delay of the signal Axp from the signal A to 0.36 ns. Next, the flow proceeds to s 705 . 
     s 705 : The successive comparative operation control circuit  530  and the delay circuit  520  execute the same operation as with s 635 , the successive comparative operation control circuit  530  determines that the potential of the signal Vi− is higher than the potential +0.6 V of the signal Vi+ when the logic of the signal Vo+ is “L”, sets the third bit to “0”, and the flow proceeds to s 715 . On the other hand, the successive comparative operation control circuit  530  determines that the potential of the signal Vi− is lower than the potential +0.6 V of the signal Vi+ when the logic of the signal Vo+ is “H”, sets the third bit to “1”, and the flow proceeds to s 720 . 
     s 715 : The successive comparative operation control circuit  530  executes the same operation as with s 645 , and sets delay of the signal Axm from the signal A to 0.38 ns. Next, the flow proceeds to s 735 . 
     s 735 : The successive comparative operation control circuit  530  and the delay circuit  520  execute the same operation as with s 665 , the successive comparative operation control circuit  530  determines that the potential of the signal Vi− is higher than the potential +0.7 V of the signal Vi+ when the logic of the signal Vo+ is “L”, sets the fourth bit to “0”, and outputs the digital signal Dout representing a binary numeral (0000). On the other hand, the successive comparative operation control circuit  530  determines that the potential of the signal Vi− is lower than the potential +0.7 V of the signal Vi+ when the logic of the signal Vo+ is “H”, sets the fourth bit to “1”, and outputs the digital signal Dout representing a binary numeral (0001). 
     s 720 : The successive comparative operation control circuit  530  executes the same operation as with s 650 , and sets delay of the signal Axp from the signal A to 0.34 ns. Next, the flow proceeds to s 740 . 
     s 740 : The successive comparative operation control circuit  530  and the delay circuit  520  execute the same operation as with s 670 , the successive comparative operation control circuit  530  determines that the potential of the signal Vi− is higher than the potential +0.5 V of the signal Vi+ when the logic of the signal Vo+ is “L”, sets the fourth bit to “0”, and outputs the digital signal Dout representing a binary numeral (0010). On the other hand, the successive comparative operation control circuit  530  determines that the potential of the signal Vi− is lower than the potential +0.5 V of the signal Vi+ when the logic of the signal Vo+ is “H”, sets the fourth bit to “1”, and outputs the digital signal Dout representing a binary numeral (0011). 
     s 700 : The successive comparative operation control circuit  530  executes the same operation as with s 630 , and sets delay of the signal Axp from the signal A to 0.22 ns. Next, the flow proceeds to s 710 . 
     s 710 : The successive comparative operation control circuit  530  and the delay circuit  520  execute the same operation as with s 660 , the successive comparative operation control circuit  530  determines that the potential of the signal Vi− is higher than the potential +0.2 V of the signal Vi+ when the logic of the signal Vo+ is “L”, sets the third bit to “0”, and the flow proceeds to s 730 . On the other hand, the successive comparative operation control circuit  530  determines that the potential of the signal Vi− is lower than the potential +0.2 V of the signal Vi+ when the logic of the signal Vo+ is “H”, sets the third bit to “1”, and the flow proceeds to s 725 . 
     s 730 : The successive comparative operation control circuit  530  executes the same operation as with s 660 , and sets delay of the signal Axp from the signal A to 0.26 ns. Next, the flow proceeds to s 750 . 
     s 750 : The successive comparative operation control circuit  530  and the delay circuit  520  execute the same operation as with s 680 , the successive comparative operation control circuit  530  determines that the potential of the signal Vi− is higher than the potential +0.3 V of the signal Vi+ when the logic of the signal Vo+ is “L”, sets the fourth bit to “0”, and outputs the digital signal Dout representing a binary numeral (0100). On the other hand, the successive comparative operation control circuit  530  determines that the potential of the signal Vi− is lower than the potential +0.3 V of the signal Vi+ when the logic of the signal Vo+ is “H”, sets the fourth bit to “1”, and outputs the digital signal Dout representing a binary numeral (0101). 
     s 725 : The successive comparative operation control circuit  530  executes the same operation as with s 675 , and sets delay of the signal Axp from the signal A to 0.18 ns. Next, the flow proceeds to s 745 . 
     s 745 : The successive comparative operation control circuit  530  and the delay circuit  520  execute the same operation as with s 645 , the successive comparative operation control circuit  530  determines that the potential of the signal Vi− is higher than the potential +0.1 V of the signal Vi+ when the logic of the signal Vo+ is “L”, sets the fourth bit to “0”, and outputs the digital signal Dout representing a binary numeral (0110). On the other hand, the successive comparative operation control circuit  530  determines that the potential of the signal Vi− is lower than the potential +0.1 V of the signal Vi+ when the logic of the signal Vo+ is “H”, sets the fourth bit to “1”, and outputs the digital signal Dout representing a binary numeral (0111). 
       FIGS. 16A and 16B  are tables for describing a method for deriving relationship between the binary numeral that the signal CNTL 531  represents, and the binary numeral that the digital signal Dout to be output by the analog-to-digital conversion by the ADC  500  represents in the case that there is no linearity with correlation with difference between the potential of the signal Vi− and the potential of the signal Vi+. 
       FIG. 16A  is a table indicating the signal CNTL 531 , correlation with the difference between the potential of the signal Vi− and the potential of the signal Vi+, and relationship between the signal CNTL 531  and the binary numeral that the digital signal Dout represents. 
     With the table shown in  FIG. 16A , a first column indicates binary numerals that the signal CNTL 531  represents, a second column indicates time difference td between the trailing edge of the signal Axp and the trailing edge of the signal Axm, a third column indicates difference ΔVi between the potential of the signal Vi− and the potential of the signal Vi+, and a fourth column indicates binary numeral codes (binary numerals) that the signal Dout represents, i.e., indicates results obtained by converting the analog value indicating the difference ΔVi between the potential of the signal Vi− and the potential of the signal Vi+ into a digital value by the ADC  500 . 
     Therefore, such as the following, in the case that there is no linearity with correlation between the binary numeral that the signal CNTL 531  represents, and the difference between the potential of the signal Vi− and the potential of the signal Vi+, analog-to-digital conversion is executed. 
     First, with regard to the inverter circuit shown in  FIGS. 11A and 11B , according to circuit simulation, the td in the second column is obtained while changing the binary numeral represented by the signal CNTL 531  in the first column from (11110) to (00000), and further from (00001) to (11111). 
     Next, circuit simulation is executed such as shown in  FIGS. 12A and 12B , and when the input signal Vi is input so as to obtain the ΔVi in the third column, tPD is obtained whereby the logics of the signals Vm and Vp are inverted, and such as shown in  FIG. 13 , correlation data of the difference between the potential of the signal Vi− and the potential of the signal Vi+, and the binary numeral represented by the signal CNTL 531  is obtained. 
     Next, such as the table in  FIG. 16A , the signal CNTL 531  corresponding to an equal interval point of the difference value between the potential of the signal Vi− and the potential of the signal Vi+ is obtained form the table such as 0.1, 0.2, and so on through 0.8. 
     Next, in accordance with the flowchart in  FIG. 15 , analog-to-digital conversion is executed using the signal CNTL 531  corresponding to the equal interval point to obtain the digital value represented by the signal Dout shown in the fourth column, i.e., binary numeral. 
       FIG. 16B  is a table indicating correlation between the signal CNTL 531 , and the difference between the potential of the signal Vi− and the potential of the signal Vi+, and relationship between the signal CNTL 531  and the binary numerals that the digital signal Dout represents. 
     With the table shown in  FIG. 16B , a first column indicates the difference ΔVi between the potential of the signal Vi− and the potential of the signal Vi+, a second column indicates time difference td between the trailing edge of the signal Axp and the trailing edge of the signal Axm, a third column indicates the binary numerals represented by the signal CNTL 531 , and a fourth column indicates binary codes (binary numerals) that the signal Dout represents, i.e., indicates results obtained by converting the analog value indicating the difference ΔVi between the potential of the signal Vi− and the potential of the signal Vi+ into a digital value by the ADC  500 . 
     Therefore, such as the following, in the case that there is no linearity with correlation between the binary numeral that the signal CNTL 531  represents, and the difference between the potential of the signal Vi− and the potential of the signal Vi+, analog-to-digital conversion is executed. 
     First, with regard to the inverter circuit shown in  FIGS. 11A and 11B , according to the same method as with the description in  FIG. 16A , correlation data between the difference between the potential of the signal Vi− and the potential of the signal Vi+, and the binary numeral represented with the signal CNTL 531  is obtained. 
     Next, in accordance with the flowchart in  FIG. 15 , three times of additions or deletions are changed using the signal CNTL 531 , and the last signal CNTL 531  is stored. 
     Next, the binary numeral of the signal Dout corresponding to the last signal CNTL 531  is determined using the table in  FIG. 16B . 
     Thus, as in  FIG. 13 , even in the event that correlation between the signal CNTL 531 , and the difference between the potential of the signal Vi− and the potential of the signal Vi+ has no proportionality relation, analog-to-digital conversion can be executed with the successive comparative operation in  FIG. 15 . 
     Thus, the ADC  500  according to the sixth embodiment is a successive comparative type analog-to-digital conversion device including a comparator including an input unit (input unit  30 ) for receiving a complementary input signal to generate an inversion complementary signal having the inversion logic of the complementary input signal, and a latch unit (latch unit  20 ) for latching the inversion complementary signal; a comparative operation control circuit  40  including a first switch (switch  41   a ) for connecting or disconnecting the latch unit and the input unit to or from a high-potential power source according to the logic of a first signal (signal Axp or signal Axm), and a second switch (switch  42   a ) for connecting or disconnecting the latch unit and the input unit to or from the high-potential power source according to the logic of a second signal (signal Axp or signal Axm); a delay circuit  520  for outputting the first signal (signal Axp or signal Axm), and the second signal (signal Axp or signal Axm); and a successive operation control circuit for outputting a control signal for controlling the period of logic change of the first signal for controlling disconnection of the first switch (switch  41   a ), and the period of logic change of the second signal for controlling disconnection of the second switch (switch  42   a ) based on the logic of a signal latched by the latch unit. 
     With a common successive comparative type analog-to-digital conversion device, in order to convert potential difference between the potential of the signal Vi+ and the potential of the signal Vi− making up the complementary input signal Vi, the difference between the potential of the signal Vi+ and the potential of the signal Vi− has to be determined according to a series of operation for comparing both potentials while changing the degree of potential increase/decrease as to one of the potential of the sampled signal Vi+, and the potential of the signal Vi−. 
     When executing increase/decrease in potential, a method has to be used wherein electric charge secured at the time of sampling of a signal is stored. Therefore, with a common successive comparative type analog-to-digital conversion device, one electrode of capacitance is connected to the node in which the electric charge secured at the time of sampling of a signal is sealed, and voltage applied to the other electrode is changed, thereby changing the degree of increase/decrease in potential. 
     Thus, with a common successive comparative type analog-to-digital conversion device, capacitance to be connected to the node in which the electric charge secured at the time of sampling of a signal is sealed results in increase in circuit area. 
     However, with the ADC  500  according to the sixth embodiment, the same effects as with a case where increase/decrease in potential is added to one of the potential of the signal Vi+ and the potential of the signal Vi− with potential comparative operation are created by providing difference between the disconnection period of the first switch to be connected to the drain of an NMOS transistor which receives the signal Vi+ of the input unit  30 , and the disconnection period of the second switch to be connected to the drain of an NMOS transistor which receives the signal Vi− of the input unit  30 . 
     Thus, capacitance to be connected to the node in which the electric charge secured at the time of sampling of a signal is sealed can be eliminated from the ADC  500  according to the sixth embodiment. Accordingly, the circuit area of the ADC  500  according to the sixth embodiment can be reduced as compared to a common successive comparative type analog-to-digital circuit. 
     The ADC  500  according to the sixth embodiment is further an analog-to-digital conversion device including a circuit for sampling and holding one signal and the other signal of the complementary input signal. 
     The delay circuit  520  of the ADC  500  according to the sixth embodiment is an analog-to-digital conversion device including a first circuit for outputting the first signal, and a second circuit for outputting the second signal, the first circuit and the second circuit include an inverter circuit, and the inverter circuit includes a logic inversion circuit for inverting the logic of a signal to be input, and a circuit for adding load capacitance to the output signal line of the logic inversion circuit according to the binary numeral represented by the control signal. 
     The delay circuit  520  of the ADC  500  according to the sixth embodiment is an analog-to-digital conversion device including a first circuit for outputting the first signal, and a second circuit for outputting the second signal, the first circuit and the second circuit include an inverter circuit, and the inverter circuit includes a logic inversion circuit for inverting the logic of a signal to be input, and a circuit for making resistance between the logic inversion circuit and a ground power source line variable according to the binary numeral represented by the control signal. 
     Seventh Embodiment 
     With the sixth embodiment, the switches  41   a  and  42   a  of the comparative operation control circuit  40  have been single switches. However, the switches  41   a  and  42   a  of an ADC  600  according to the seventh embodiment include multiple switches. 
       FIG. 17  illustrates the ADC  600  according to the seventh embodiment. The ADC  600  according to the seventh embodiment includes a latch unit  20 , an input unit  30 , a comparative operation control circuit  640 , a delay circuit  620 , a successive comparative operation control circuit  630 , and a sample-hold circuit  540 . 
     The latch unit  20 , input unit  30 , and sample-hold circuit  540  are the same circuits described in the sixth embodiment, and description regarding the configuration and operation thereof will be omitted. 
     The successive comparative operation control circuit  630  causes the latch unit  20 , input unit  30 , comparative operation control circuit  640 , and delay circuit  620  to execute operation shown in the flowchart in  FIG. 15  for comparing the potential of the signal Vi+ and the potential of the signal Vi− making up the complementary input signal Vi in sync with the signal CLK. As described later, in order to execute selection of one of signals Axp 1  and Axp 2 , and selection of one of signals Axm 1  and Axm 2 , to be output from the delay circuit  620 , the number of bits of the binary numeral represented by a signal CNTL 631  that the successive comparative operation control circuit  630  outputs is greater than the number of bits of the signal CNTL 531  according to the sixth embodiment by one bit. 
     The delay circuit  620  includes inverters  621   a ,  622   a , and  623   a  which output the signal Axp 1 , inverters  621   b ,  622   b , and  623   b  which output the signal Axp 2 , inverters  624   a ,  625   a , and  626   a  which output the signal Axm 1 , and inverters  624   b ,  625   b , and  626   b  which output the signal Axm 2 . Here, time difference between the logic trailing edge of the signal A and the trailing edge of the signal Axp 1 , and time difference between the logic trailing edge of the signal A and the trailing edge of the signal Axp 2  are changed according to the binary numeral represented by the signal CNTL 631 . 
     Also, time difference between the logic trailing edge of the signal A and the trailing edge of the signal Axm 1 , and time difference between the logic trailing edge of the signal A and the trailing edge of the signal Axm 2  are changed according to the binary numeral represented by the signal CNTL 631 . 
     The comparative operation control circuit  640  includes switches  41   b  and  41   c  for connecting or disconnecting the latch unit  20  and the input unit  30  to or from the high-potential power source Vcc according to the logics of the signals Axp 1  and Axp 2 , switches  42   b  and  42   c  for connecting or disconnecting the latch unit  20  and the input unit  30  to or from the high-potential power source Vcc according to the logics of the signals Axm 1  and Axm 2 , and a switch  43   a  for connecting or disconnecting the input unit  30  to or from the ground power source Vss according to the logic of the signal A. 
     Note that, with the above configuration, the number of switches used for connecting or disconnecting the high-potential power source Vcc to or from the source of the N-type transistor of which the gate is connected to the signal Vi− has been two, and the number of switches used for connecting or disconnecting the high-potential power source Vcc to or from the source of the N-type transistor of which the gate is connected to the signal Vi+ has been two, but each of these is not restricted to two, and rather may be multiple of two or more. In this case, the signal Axpn (n is a positive integer equal to or greater than 2), and the signal Axmn (n is a positive integer equal to or greater than 2), to be connected to each of the switches are each independent, and the trailing-edge period of the logic of each signal is also set independently. 
     Thus, the disconnecting periods of the switches  41   b ,  41   c ,  42   b , and  42   c  are adjusted by the comparative operation control circuit  640 , whereby the lowering speeds of the potentials of the signals Vm and Vp between the latch unit  20  and the input unit  30  can be adjusted at the time of comparing the potentials of the signals Vi+ and Vi−. Thus, the difference between the potential of the signal Vi+ and the potential of the signal Vi−, and correlation between the trailing-edge period of the signal A, and the trailing-edge periods of the signals Axp 1  and Axp 2 , shown in  FIG. 13  can be adjusted. Note that it goes without saying that even when there are multiple switches, multiple signals Axp, and multiple signals Axm, the disconnecting period of each switch can be adjusted. 
     Thus, the ADC  600  according to the seventh embodiment is a successive comparative analog-to-digital conversion device including a comparator including an input unit (input unit  30 ) for receiving a complementary input signal to generate a inversion complementary signal having the inversion logic of the complementary input signal, and a latch unit (latch unit  20 ) for latching the inversion complementary signal; a comparative operation control circuit  640  including multiple first switches (switches  41   b  and  41   c ) for connecting or disconnecting the latch unit and the input unit to or from the high-potential power source according to the logic of each of multiple first signals (signals Axp or signals Axm), and multiple second switches (switches  42   b  and  42   c ) for connecting or disconnecting the latch unit and the input unit to or from the high-potential power source according to the logic of each of multiple second signals (signals Axp or signals Axm); a delay circuit  520  for outputting each of the first signals (signals Axp or signals Axm), and each of the second signals (signals Axp or signals Axm); and a successive operation control circuit for outputting a control signal which controls the period of logic change of each of the first signals for controlling disconnection of each of the multiple first switches (switches  41   a ), and the period of logic change of each of the second signals for controlling disconnection of each of the multiple second switches (switches  42   a ). 
     Eighth Embodiment 
     With the sixth embodiment, the signals Axp and Axm are output from an inverter for corrugating disposed on the subsequent stage of an inverter for generating signal delay at the delay circuit  520 . On the other hand, with the eighth embodiment, the signals Axp and Axm are output from an inverter for generating signal delay at a delay circuit  720 . 
       FIG. 18  illustrates an ADC circuit  700  according to the eighth embodiment. The ADC  600  according to the seventh embodiment includes a latch unit  20 , an input unit  30 , a comparative operation control circuit  40 , a delay circuit  720 , a successive comparative operation control circuit  530 , and a sample-hold circuit  540 . 
     The latch unit  20 , input unit  30 , comparative operation control circuit  40 , delay circuit  720 , successive comparative operation control circuit  530 , and sample-hold circuit  540  are the same circuits as described with the sixth embodiment, and description regarding configuration and operation thereof will be omitted. 
     The delay circuit  720  includes inverters  721 ,  722 ,  723 ,  724 ,  725 ,  726 , and  727 . 
     The inverter  727  receives the signal CLK and outputs the inversion signal thereof. The inverters  723  and  726  receive the signal to be output from the inverter  727 , and output the inversion signal thereof. The inverters  722  and  725  receive the output signals from the inverters  723  and  726  and output the inversion signals thereof, respectively. The inverters  721  and  724  receive the signals to be output from the inverters  722  and  725  and output the inversion signals thereof as the signals Axp and Axm. 
     With the inverters  721  and  724 , delay time from input of the input signal to output of the output signal varies according to the signal CNTL 531 . 
     Thus, with the delay circuit  720 , time difference from the trailing edge of the signal A to the trailing edge of the signal Axp, and time difference from the trailing edge of the signal A to the trailing edge of the signal Axm are the same as with the delay circuit  520 . However, with the delay circuit  720 , the period of potential change at the time of the trailing edges of the logics of the signals Axp and Axm becomes long. This is because the signals output from the inverters  721  and  725  become the signals Axp and Axm as is without passing through an inverter for corrugating, and accordingly, the potential changes of the signals Axp and Axm become moderate. 
     Thus, time from start of disconnecting operation to end thereof between the high-potential power source Vcc, and the latch unit  20  and the input unit  30  by the switch  41   a  of the comparative operation control circuit  40  is longer than time required for disconnecting operation by the switch  41   a  at the ADC  500 . 
     As a result thereof, the falling speeds of the potentials of the signals Vm and Vp between the latch unit  20  and the input unit  30  changes at the time of comparing the potentials of the signals Vi+ and Vi− as compared to the case of the ADC  500  according to the first embodiment. 
     Thus, the difference between the potential of the signal Vi+ and the potential of the signal Vi−, and correlation between the trailing-edge period of the signal A and the trailing-edge periods of the signals Axm 1  and Axm 2  and the signals Axp 1  and Axp 2 , shown in  FIG. 13 , can be adjusted. 
     Thus, the ADC  700  according to the eighth embodiment is a successive comparative type analog-to-digital conversion device including a comparator including an input unit (input unit  30 ) for receiving a complementary input signal to generate a inversion complementary signal having the inversion logic of the complementary input signal, and a latch unit (latch unit  20 ) for latching the inversion complementary signal; a comparative operation control circuit  40  including a first switch (switch  41   a ) for connecting or disconnecting the latch unit and the input unit to or from a high-potential power source according to the logic of a first signal (signal Axp or signal Axm), and a second switch (switch  42   a ) for connecting or disconnecting the latch unit and the input unit to or from the high-potential power source according to the logic of a second signal (signal Axp or signal Axm); a delay circuit  520  including a first inverter (inverter  721 ) for outputting the first signal (signal Axp or signal Axm), and a second inverter (inverter  724 ) for outputting the second signal (signal Axp or signal Axm); and a successive operation control circuit for outputting a control signal for controlling the period of logic change of the first signal for controlling disconnection of the first switch (switch  41   a ), and the period of logic change of the second signal for controlling disconnection of the second switch (switch  42   a ) based on the logic of a signal latched by the latch unit. 
     The first inverter and the second inverter according to the eighth embodiment include a logic inversion circuit for inverting the logic of an input signal, and a circuit for adding load capacitance to an output signal line of the logic inversion circuit according to the binary numeral represented by the control signal. 
     The first inverter and the second inverter according to the eighth embodiment include a logic inversion circuit for inverting the logic of an input signal, and a circuit for varying resistance between the logic inversion circuit and the ground power source line according to the binary numeral represented by the control signal. 
     Ninth Embodiment 
     With the sixth embodiment, the signals Axp and Axm are output from an inverter for corrugating disposed on the subsequent stage of an inverter for generating signal delay at the delay circuit  520 . On the other hand, with the ninth embodiment, the signals Axp and Axm are output from an inverter for generating signal delay at a delay circuit  820 . 
       FIG. 19  illustrates an ADC circuit  800  according to the ninth embodiment. The ADC  800  according to the ninth embodiment includes a latch unit  20 , an input unit  30 , a comparative operation control circuit  40 , a comparative operation control circuit  840 , a delay circuit  820 , a successive comparative operation control circuit  530 , and a sample-hold circuit  540 . 
     The latch unit  20 , input unit  30 , successive comparative operation control circuit  530 , and sample-hold circuit  540  are the same circuits as described with the sixth embodiment, and description regarding configuration and operation thereof will be omitted. 
     The delay circuit  820  includes inverters  821 ,  822 ,  823 ,  824 ,  825 ,  826 , and  827 . The inverter  827  receives the signal CLK and outputs the inversion signal thereof to the inverters  823  and  826 . The inverter  827  outputs the signal A, signal Axp 1 , and signal Axm 1 . The inverters  823  and  826  receive the output signal from the inverter  827  and output the inversion signal thereof to the inverters  822  and  825 . The inverters  822  and  825  receive the output signals from the inverters  823  and  826  and output the inversion signal thereof to the inverters  821  and  824 . The inverter  821  outputs the signal Axp 2  to the comparative operation control circuit  840 , and the inverter  824  outputs the signal Axm 2  to the comparative operation control circuit  840 . 
     The comparative operation control circuit  840  includes switches  41   b  and  41   c  for connecting or disconnecting the latch unit  20  and the input unit  30  to or from the high-potential power source Vcc according to the logics of the signals Axp 1  and Axp 2 , switches  42   b  and  42   c  for connecting or disconnecting the latch unit  20  and the input unit  30  to or from the high-potential power source Vcc according to the logics of the signals Axm 1  and Axm 2 , and a switch  43   a  for connecting or disconnecting the input unit  30  to or from the ground power source Vss according to the logic of the signal A. 
     Thus, disconnection between the latch unit  20  and input unit  30 , and the high-potential power source Vcc by the switches  41   b  and  42   b , and connection between the input unit  30  and the ground Vss by the switch  43   a  are simultaneously executed by the delay circuit  820 . Also, after delay time according to the binary numeral represented by the signal CNTL 531  of the successive comparative operation circuit  530  by the delay circuit  820 , disconnection between the latch unit  20  and input unit  30 , and the high-potential power source Vcc by the switches  41   c  and  42   c  is executed. 
     Thus, the disconnection periods of the switches  41   c  and  42   c  are adjusted by the delay circuit  820 , and thus, when comparing the potentials of the signals Vi+ and Vi−, the falling speed of the potentials of the signals Vm and Vp between the latch unit  20  and the input unit  30  can be adjusted. Thus, the difference between the potential of the signal Vi+ and the potential of the signal Vi−, and correlation between the trailing-edge period of the signal A, and the trailing-edge periods of the signals Axm 1  and Axm 2  and the signals Axp 1  and Axp 2 , shown in  FIG. 13  can be adjusted. 
     Thus, the ADC  800  according to the ninth embodiment is a successive comparative analog-to-digital conversion device including a comparator including an input unit (input unit  30 ) for receiving a complementary input signal to generate a inversion complementary signal having the inversion logic of the complementary input signal, and a latch unit (latch unit  20 ) for latching the inversion complementary signal; a comparative operation control circuit  840  including a first switch (switch  41   c ) for connecting or disconnecting the latch unit and the input unit to or from a high-potential power source according to the logic of a first signal (signal Axp 2  or signal Axm 2 ), a second switch (switch  42   c ) for connecting or disconnecting the latch unit and the input unit to or from the high-potential power source according to the logic of a second signal (signal Axp 2  or signal Axm 2 ), and a third switch (switch  41   b ) and a fourth switch (switch  42   b ) for connecting or disconnecting the latch unit and the input unit to or from the high-potential power source according to the logic of a third signal (signal Axp 1  or signal Axm 1 ); a delay circuit  820  for outputting the first signal (signal Axp 2  or signal Axm 2 ), the second signal (signal Axp 2  or signal Axm 2 ), and the third signal (signal Axp 1  or signal Axm 1 ); and a successive operation control circuit  530  for outputting a control signal for controlling the period of logic change of the first signal for controlling disconnection of the first switch (switch  41   c ), and the period of logic change of the second signal for controlling disconnection of the second switch (switch  42   c ) based on the logic of a signal latched by the latch unit. 
     Tenth Embodiment 
     With the sixth embodiment, the sample-hold circuit  540  samples and holds the potential of the signal Vi+ and the potential of the signal Vi− making up the input signal Vin. On the other hand, an ADC  900  according to the tenth embodiment includes no sample-hold circuit  540 . The reason thereof is because with the tenth embodiment, sample-hold of the potential of the input signal Vi is executed at a circuit for inputting the input signal Vi to the ADC  900  illustrated by the tenth embodiment. 
       FIG. 20  is a diagram illustrating the ADC  900  according to the tenth embodiment. The ADC  900  includes a latch unit  20 , an input unit  30 , a comparative operation control circuit  40 , a delay circuit  520 , and a successive comparative operation control circuit  530 . Accordingly, the ADC  900  differs from the ADC  500  in that the sample-hold circuit  540  is not included. 
     Thus, the ADC  900  includes no sample-hold circuit  540 , whereby the area occupied with circuits can be reduced as compared to the ADC  500  according to the sixth embodiment. 
     Thus, the ADC  900  according to the tenth embodiment is a successive comparative type analog-to-digital conversion device including a comparator including an input unit (input unit  30 ) for receiving a complementary input signal to generate a inversion complementary signal having the inversion logic of the complementary input signal, and a latch unit (latch unit  20 ) for latching the inversion complementary signal; a comparative operation control circuit  40  including a first switch (switch  41   a ) for connecting or disconnecting the latch unit and the input unit to or from a high-potential power source according to the logic of a first signal (signal Axp or signal Axm), and a second switch (switch  42   a ) for connecting or disconnecting the latch unit and the input unit to or from the high-potential power source according to the logic of a second signal (signal Axp or signal Axm); a delay circuit  520  for outputting the first signal (signal Axp or signal Axm) and the second signal (signal Axp or signal Axm); and a successive operation control circuit for outputting a control signal for controlling the period of logic change of the first signal for controlling disconnection of the first switch (switch  41   a ), and the period of logic change of the second signal for controlling disconnection of the second switch (switch  42   a ) based on the logic of a signal latched by the latch unit. 
     Eleventh Embodiment 
       FIG. 21  is a diagram illustrating a signal processing device  1  (receiver  1 ) using the ADC illustrated by the sixth embodiment through tenth embodiment. The signal processing device  1  is a device which includes an antenna  2 , a filter circuit and amplifier  3 , an ADC  4 , and a DSP demodulator  5 , and outputs a signal for propagating audio data or image data before modulation which is available at a display device  6 , audio generating device  7 , or the like. 
     The signal processing device  1  is a device for restoring the modulated signal received by the antenna  2  into the original signal. The filter circuit and amplifier  3  is a circuit for amplifying the modulated signal by attenuating noise thereof. The ADC circuit  4  is a circuit for converting the input modulated signal into a digital signal. Note that the ADC circuit  4  is the ADC circuit according to the sixth through tenth embodiments. The DSP demodulator  5  receives the signal digitized by the ADC circuit  4 , restores the signal before modulation, and outputs this to the display device  6  or audio generating device  7 . Here, the signal before modulation means a signal relating to image data for the display device  6 , a signal relating to audio for the audio generating device  7 , or the like. 
     Thus, with the ADC  4 , when executing successive comparative operation to detect the difference between the potential of the signal Vi+ and the potential of the signal Vi− making up the input signal Vi, a comparative operation control circuit is used, whereby the number of elements making up the circuits can be reduced. Accordingly, with the whole receiver  1 , the circuit area of the ADC  4  can be reduced, and accordingly, the circuit area of the whole receiver  1  can be reduced. 
     Thus, the signal processing device according to the eleventh embodiment is a receiver (receiver  1 ) including a filter circuit (filter circuit and amplifier  3 ) for attenuating noise from an analog reception signal; an amplifier (filter circuit and amplifier  3 ) for amplifying the analog reception signal of which the noise is attenuated; an analog-tot-digital circuit according to one of the sixth through tenth embodiments for converting the analog reception signal of which the noise is attenuated into a digital signal; a DSP demodulator (DSP demodulator  5 ) for restoring the signal before modulation from the reception signal of which the noise is attenuated. 
     All examples and conditional language recited herein are intended for pedagogical purposes to aid the reader in understanding the invention and the concepts contributed by the inventor to furthering the art, and are to be construed as being without limitation to such specifically recited examples and conditions, nor does the organization of such examples in the specification relate to a depicting of the superiority and inferiority of the invention. Although the embodiments of the present invention have been described in detail, it should be understood that the various changes, substitutions, and alterations could be made hereto without departing from the spirit and scope of the invention.