Patent Document

BACKGROUND OF THE INVENTION 
     The invention pertains to an integrated frequency demodulator subcircuit. 
     A subcircuit of this kind is described in the periodical &#34;Archiv der elektronischen ubertragung (AEu)&#34;, 1982, pp. 292 to 298, particularly FIGS. 1 and 2 on pages 293 and 294. This prior art deals with the frequency demodulation of VHF broadcast signals, the arrangement described being designed to also demodulate the stereo signal contained in the VHF broadcast signal in accordance with the European standard. 
     SUMMARY OF THE INVENTION 
     One object of the invention is to optimize and improve the prior art subcircuit in such a way as to make it suitable for use in digital frequency demodulations for SECAM color-television signals. Investigations by the inventors show that the prior art arrangement is suited in principle for SECAM demodulation, but the amount of area required by the arrangement on the integrated-circuit chip would be prohibitively large. In accordance with the invention, the circuit is designed to occupy a small area on the chip. 
    
    
     BRIEF DESCRIPTION OF THE DRAWING 
     The invention will be better understood from a reading of the following detailed description in conjunction with the drawing in which: 
     FIG. 1 is a circuit diagram of a frequency demodulator subcircuit in accordance with the invention; 
     FIG. 2 is a circuit diagram of another, simplified embodiment of the subcircuit of FIG. 1; and 
     FIG. 3 is a circuit diagram of a further, simplified embodiment which is designed to be implemented in two-phase insulated-gate field-effect transistor technology. 
    
    
     DETAILED DESCRIPTION 
     In the circuit diagram of FIG. 1, first and second digital mixers dm1 and dm2 are of conventional design and are interposed in like manner in the two signals paths connected to the input e, the following first and second digital low-pass filters tp1 and tp2, respectively, and the first and second sampling stages as1 and as2, which follow the low-pass filters tp1 and tp2, respectively, and are clocked with the second clock signal f2. At the same sampling instant in each period of the first clock signal f1, binary numbers which are equivalent to the decimal numbers 1, 0, -1, 0, 1 . . . are fed to the first digital mixer dm1, while the second digital mixer dm2 is supplied at the same sampling instants with binary numbers which are equivalent to the decimal numbers 0, 1, 0, -1, 0 . . . As is stated in the publication mentioned above, through the supply of these numerical values, the digital signals at the input e, which are delivered by a suitable analog-to-digital converter (not shown), are mixed with two signals which have the same frequency as the second clock signal, but differ in phase by exactly 90°. 
     In FIG. 1, each of the two low-pass filters tp1, tp2 consists of a cascade of five nonrecursive digital-filter stages each with the transfer function H&#39;(z)=1+z -1 , so that the transfer function of each of the low-pass filters tp1, tp2 is: H(z)=(1+z -1 ) 5 , where z is the complex frequency variable corresponding to the frequency of the first clock signal f1. The individual digital-filter stages are of identical design. Each of them consists of the delay stage v, whose delay is equal to the period of the first clock signal f1, and the adder stage a, one input of which is presented with the undelayed input signal, while the other is supplied with the signal delayed by the delay stage v. Located at the ends of the signal paths and, thus, at the outputs of the digital filters tp1 and tp2 are the first and second sampling stages as1 and as2, respectively, which are clocked with the second clock signal f2 whose frequency is equal to one quarter of the frequency of the first clock signal f1. The output of the first sampling stage as1 is the output x of the first signal path, and that of the second sampling stage as2 is the output y of the second signal path. 
     Although, for simplicity and ease of illustration, only connecting lines are shown in the figures of the drawing between individual subcircuits as if only single conductors were present, the interconnections are buses consisting of many parallel conductors because the digital signals to be processed are present in parallel form and the signal processing in each of the stages takes place and is completed during one period of the first clock signal f1. This is also apparent from the fact that the frequency of the first clock signal f1 is equal to four times the frequency of the chrominance-subcarrier reference of the SECAM color-television signal; accordingly, the second clock frequency f2 is equal to this reference frequency. 
     FIG. 2 shows a simplified embodiment of the arrangement of FIG. 1 which requires only five of the ten delay stages v of FIG. 1, namely the delay stages v1, v2, v3, v4 and v5, which follow the input e in a cascade arrangement, and whose input signals y2, x2, y1, x1, and y0 and the output signal x0 of the fifth delay stage v5 are applied alternately to the two signal paths. In the first signal path, the input signals of the second and fourth delay stages v2 and v4 and the output signal of the fifth delay stage v5 are fed to the first computing circuit r1, which is designed exclusively to calculate the term 2 -6  (x0-10x1+5x2), whereas the input signals y2, y1, and y5 of the first, third, and fifth delay stages v1, v3, and v5 are fed to the second computing circuit r2, which is designed exclusively to calculate the term 2 -6  (5y0-10y1+y2). What was said about the buses used is indicated in FIG. 2 by a diagonal in the lead connected to the input e, which is designated by the reference number 13, and by diagonals in the leads running to the outputs x and y, which are designated by the reference numerals 11. The numerals signify that 13-bit and 11-bit digital words, respectively, are transferred over these buses in parallel. Accordingly, all subcircuits in the arrangement according to the invention handle the signals in parallel. 
     FIG. 3 shows a circuit diagram of a further simplification if the invention is realized using two-phase insulated-gate field-effect transistor circuits. This technique has been known for a long time and is described, for example, in &#34;The Electronic Engineer&#34;, March 1970, pages 56 to 61. 
     In this realization, the five delay stages v1 . . . v5 of FIG. 2, their two associated computing circuits r1, r2, and the two sampling stages as1, as2 of FIG. 1 are functionally united as follws. The input e is followed by the three delay stages v1&#39;, v2&#39;, v3&#39; in a cascade arrangement. The first clock signal f1 is divided into the two clock phases ph1, ph2 of the two-phase clock system, which have the same frequency as the first clock signal f1. On the leading edge of each first clock phase ph1, the respective signal at the input e is transferred to the first input of the multiplier m, whose second input is fed with a binary word corresponding to the decimal factor &#34;5&#34;. On the leading edge of each second clock phase ph2, the output signal of the multiplier m is transferred to the input of the doubler stage vd, i.e., a stage which multiplies the output signal of the multiplier m by a binary word corresponding to the decimal factor 2. This can be done simply by shifting the output signal of the multiplier m one place to the left in the straight binary code, as is well known. 
     On the next leading edge of the first clock phase ph1, the output signal of the doubler stage vd is applied to the subtrahend input of the subtracter sb. The output of the third delay stage v3&#39; is coupled to the first input of the first electronic switch s1, whose output is connected to minuend input of the subtracter sb. The second input of the first electronic switch s1 is connected to the output of the multiplier m via the first delay element vg1, which delays the multiplier&#39;s output signal by 2.5 times the period of the first clock signal f1. The output of the subtractor sb is coupled to the first input of the adder ad. The input e is preceded by the second delay element vg2, which provides a delay equal to half the period of the first clock signal f1. The input thus formed, e&#39;, is connected to the first input of the second electronic switch s2, whose second input is connected to the output of the multiplier m, and whose output is coupled to the second input of the adder ad. 
     Within four periods of the first clock signal f1, the first inputs of the two switches s1, s2 are connected to the outputs of the respective switches during the second and fourth periods, and the second inputs during the first and third periods. The output of the adder ad is connected to the input of the third electronic switch s3. The input is connected to the output x of the first signal path during the third period, and to the output y of the second signal path during the fourth period. 
     As mentioned above, the computing subcircuits, i.e., the multiplier m, the subtracter sb, and the adder ad, perform the computation within a maximum time equal to half the period of the first clock signal f1, i.e., within a maximum period of 28 ns. Such simple computing circuits are realizeable without difficulty. The multiplier m is preferably implemented as a series combination of two adders the first of which shifts the multiplier&#39;s input signal two places to the left, which correspond to a multiplication by the decimal factor 4, and the second of which adds the input signal to the result of the shift. After one period of the first clock signal s1, a signal equal to ten times the signal value present at the input e at the end of the preceding period appears at the subtrahend input of the subtracter sb. Similar time considerations apply to the other input of the subtracter sb and to the two inputs of the adder ad.

Technology Category: 5