Patent Document

CROSS-REFERENCE TO RELATED APPLICATIONS 
   This is a continuation application of application Ser. No. 11/039,633 filed on Jan. 14, 2005 and issued as U.S. Pat. No. 7,042,282 issued on May 9, 2006 which is in turn a continuation application of application Ser. No. 10/431,976 filed on May 7, 2003 issued as U.S. Pat. No. 6,844,776 issued on Jan. 18, 2005 which is in turn a continuation application of application Ser. No. 09/684,497 filed on Oct. 6, 2000 and issued as U.S. Pat. No. 6,734,724 on May 11, 2004. 

   BACKGROUND 
   1. Field of the Invention 
   The present invention relates to power amplifiers, particularly switched-mode power amplifiers. 
   2. Description of Related Art 
   Switched-mode power amplifiers have demonstrated the capability of producing, with high power-added efficiency (PAE), phase-modulated signals that have very high signal quality—i.e., low root-mean-square (RMS) phase error relative to an ideal signal and little or no degradation in power spectral density (PSD). These power amplifiers have also been demonstrated to be highly tolerant of temperature variation, and are believed to be highly tolerant to fabrication-process variation, making them attractive for high-volume applications such as consumer electronics. Such power amplifiers include a switch connected to a resonant network; the output of the resonant network is connected in turn to a load (e.g., the antenna in a radio transmitter). 
   An early switched-mode amplifier is described in U.S. Pat. No. 3,900,823 to Sokal et al., incorporated herein by reference. Sokal et al. describes the problem (created by unavoidable feedthrough from amplifier input to amplifier output) of power control at low power levels and proposes solving the problem by controlling RF input drive magnitude to a final amplifier stage. In particular, the input drive magnitude of the final stage is controlled by using negative feedback techniques to control the DC power supply of one or more stages preceding the final stage. Various other known techniques use variation of amplifier power supply for linearization as described, for example, in the following patents, incorporated herein by reference: U.S. Pat. No. 5,091,919; U.S. Pat. No. 5,142,240, and U.S. Pat. No. 5,745,526. 
   Another type of switched-mode amplifier, that does not require the use of negative feedback as in Sokal, is described in U.S. patent application Ser. Nos. 09/247,095 and 09/247,097 of the present assignee, entitled HIGH-EFFICIENCY MODULATING RF AMPLIFIER and HIGH-EFFICIENCY AMPLIFIER OUTPUT LEVEL AND BURST CONTROL, respectively, filed Feb. 9, 1999 (WO0048306 and WO0048307) and U.S. patent application Ser. No. 09/637,269 now U.S. Pat. No. 6,636,112, entitled HIGH-EFFICIENCY MODULATING RF AMPLIFIER, filed Aug. 10, 2000, all incorporated herein by reference. In the latter switched-mode power amplifiers, the average power is determined by two signals: the switch supply signal and the switch control signal. The switch supply signal is the DC voltage available on one side of the switch; as this voltage increases, the peak voltage of the oscillatory signals developed within the resonant network and subsequently delivered to the load also increases. The switch control signal is typically a phase-modulated signal that controls the switch (i.e., determines whether the switch is on or off). This switch control signal should be strong enough to toggle the switch on and off but should not be excessively strong: unlike a linear amplifier in which the strength of the output signal is determined by the strength of the input signal, in a switched-mode power amplifier, if the switch control signal is too strong, the excess signal merely leaks through the switch and into the resonant network (i.e., feedthrough). When this occurs, a version of the switch control signal that is out-of-phase with respect to the desired signal adds to the desired signal within the resonant network, altering both the phase and the amplitude of the output signal in an undesirable way. 
   French Patent 2,768,574 also describes a switched-mode power amplifier arrangement. Referring to  FIG. 1 , in this arrangement, the power amplifier circuit comprises a DC-to-DC converter  20  and a power amplifier  30 . The DC-to-DC converter  20  includes a pulse-width modulator  22 , a commutator/rectifier  24  and a filter  26 . 
   The pulse-width modulator  22  is coupled to receive a DC-to-DC command input signal from a signal input terminal  21 , and is arranged to apply a pulse-width-modulated signal to the commutator/rectifier  24 . The commutator/rectifier  24  is coupled to receive a DC-to-DC power supply input signal from a signal input terminal  25 , and is also coupled to apply a switched signal to filter  26 . The filter  26  in turn applies a filtered switched signal  28  in common to multiple stages of the power amplifier  30 . 
   A circuit of the foregoing type is substantially limited by the frequency of the pulse-width modulator. In addition, common control of multiple power amplifier stages in the manner described may prove disadvantageous as described more fully hereinafter. 
   It is desirable to achieve more precise control of switched-mode-generated RF signals, including amplitude-modulated signals, such that the aforementioned benefits of switched-mode power amplifiers may be more fully realized. 
   SUMMARY OF THE INVENTION 
   This invention controls and modulates switched-mode power amplifiers to enable the production of signals that include amplitude modulation (and possibly, but not necessarily, phase modulation), the average power of which may be controlled over a potentially wide range. 
   In order to produce amplitude-modulated signals, the DC switch supply voltage is replaced by a time-varying switch supply signal that is related to the desired amplitude modulation. This switch supply signal can be either the desired amplitude modulation signal itself or a pre-distorted version thereof, where the pre-distortion is such that the output signal has the desired amplitude modulation. In the latter case, the pre-distortion corrects for amplitude non-linearity (so-called AM/AM distortion) in the switch and/or the resonant network. 
   The foregoing modification alone, however, may be insufficient to provide as much dynamic range in the output signal as may be desired. Also, the modification may not be sufficient to maintain dynamic range in the amplitude modulation while adjusting the average power of the output signal. Both of these problems are caused by the undesirable leakage signal described previously; its contribution to the output is largely independent of the level of the switch supply signal. That is, the switch supply signal may be reduced to zero volts (the minimum possible amplitude), yet the output signal will still be at a relatively high level; below some point, the amplitude modulation imparted through the switch supply signal is manifest less and less in the output signal. 
   Similarly, the severity of amplitude-dependent phase shift (so-called AM/PM distortion) increases as the switch supply signal decreases. This effect arises because the leakage of the switch control signal is out of phase relative to the desired signal. As the switch supply signal decreases, the desired signal decreases as well, whereas the leakage signal does not; since these two signals are out of phase, the phase of their sum is increasingly dominated by the phase of the leakage signal. This invention, in one aspect thereof, modifies the switched-mode power amplifier by adjusting the amplitude of the switch control signal to reduce the undesirable leakage effect. As a result, it becomes possible to produce output signals having average power anywhere within a wide range, or to greatly increase the dynamic range over which amplitude modulation may be produced at a given average power level, or both. 

   
     BRIEF DESCRIPTION OF THE DRAWING FIGURES 
     The present invention may be further understood from the following description in conjunction with the appended drawing figures. In the figures: 
       FIG. 1  is a block diagram of a known switched-mode power amplifier in a variable power supply voltage is applied in common to multiple stages; 
       FIG. 2  is a block diagram of a switched-mode power amplifier without amplitude modulation capability; 
       FIG. 3  is a diagram comparing AM/PM distortion in a switched-mode power amplifier without a countermeasure of the invention and with a countermeasure of the invention; 
       FIG. 4  is a waveform diagram of waveforms in the circuit of  FIG. 2 ; 
       FIG. 5  is one possible circuit that may be used to control the application of power to one or more power amplifier stages; 
       FIG. 6  is another possible circuit that may be used to control the application of power to one or more power amplifier stages; 
       FIG. 7  is still another possible circuit that may be used to control the application of power to one or more power amplifier stages; 
       FIG. 8  is a block diagram of a generalized efficient power amplifier structure; 
       FIG. 9  is a block diagram of a switched-mode power amplifier having amplitude modulation capability; 
       FIG. 10  is a waveform diagram of waveforms in the circuit of  FIG. 9 ; 
       FIG. 11  is another waveform diagram of waveforms in the circuit of  FIG. 9 ; 
       FIG. 12  is a more detailed diagram of an exemplary embodiment of the switched-mode power amplifier of  FIG. 9 ; and 
       FIG. 13  is a waveform diagram of waveforms in the circuit of  FIG. 12 . 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   Referring now to  FIG. 2 , a block diagram is shown of a switched-mode power amplifier. A switch  201  is coupled to a resonant network  205  and to power control logic  215 , which is coupled in turn to a DC supply  203 . The resonant network is coupled to a load  207 . Control of the switch  201  is accomplished using a control signal  209 , applied to an amplifier  211 . The amplifier  211  produces a switch control signal  219 , which is applied to the switch  201 . As the switch  201  is opened and closed responsive to the control signal  209 , the resonant network  205  shapes the switch voltage to produce a desired output signal  213 . 
   In the amplifier of  FIG. 2 , the signals  209  and  219  are constant-amplitude (CA) signals (i.e., oscillatory signals having a constant peak amplitude) that may be phase-modulated. The amplitude of the switch control signal  219  is set by the power control logic  215 . The power control logic  215  also controls a DC supply voltage  216  produced by the DC supply  203  and supplied to the switch  201 . As the power control logic  215  causes the DC supply voltage  216  to increase, the peak voltage of the oscillatory signals developed within the resonant network  205  and subsequently delivered to the load  207  also increases. Similarly, as the power control logic  215  causes the DC supply voltage  216  to decrease, the peak voltage of the oscillatory signals developed within the resonant network  205  and subsequently delivered to the load  207  also decreases. 
   Further details of the amplifier chain of  FIG. 2  in accordance with an exemplary embodiment of the invention are described in the foregoing copending U.S. patent applications. In addition, a bias control arrangement may be used to achieve optimal bias of the switch  201  under various conditions as described more fully in U.S. patent application Ser. No. 09/684,496 now U.S. Pat. No. 6,323,731, filed on even date herewith and incorporated herein by reference. 
   In accordance with one aspect of the invention, a signal  218  is used to control the amplitude of the switch control signal  219  in a coordinated manner with control of the DC supply voltage  216 , thereby avoiding excess leakage of the switch control signal  219  through the switch  201  and into the resonant network  205 . 
   More particularly, in any physical embodiment, a stray (unintended) capacitance  212  around the switch  201  is unavoidably present. This stray capacitance provides a leakage path for the switch control signal  219  to leak into the resonant network  205 , where it mixes with the desired switch output signal. Since the switch control signal  219  is out-of-phase with the desired switch output signal, a large phase shift will occur at the switch output when the desired output signal magnitude is near to or smaller than that of the leakage signal. This effect is shown in  FIG. 3 , which depicts output phase and output magnitude as parametric functions of desired magnitude (i.e., as desired magnitude decreases, the curves of  FIG. 3  are traced out in the counter-clockwise direction). In the illustrated case, signal leakage is assumed to be 35 dB below the maximum output signal (1.7%), at a relative phase shift of −170 degrees. If the switch control signal is not reduced (line A), then the amplifier output signal suffers severe AM-PM (and AM—AM) distortion when the desired output magnitude is less than 10% of the peak output magnitude. 
   This effect may be counteracted, for lower amplitude output signals (e.g., less than 10% of the peak output magnitude), by correspondingly reducing the switch control signal (e.g., to 10% of its original value). As  FIG. 3  shows, this measure essentially removes the AM-PM and AM—AM distortion from the desired output signal (line B). In principle, this technique can be extended to arbitrarily low desired output signal magnitudes. 
   For illustration purposes, consider the need to produce a constant-amplitude RF signal in a time-slotted network, in which the output power may vary from slot to slot. In the amplifier of  FIG. 2 , this manner of operation may be achieved by holding the supply voltage  216  constant during a given time slot, and by holding the peak amplitude of the control signal constant during the time slot as illustrated in  FIG. 4 . As a result, the peak amplitude of the output signal  213  is constant during a given time slot. Note that when the supply voltage  216  is at a low level, the control signal  219  is also at a correspondingly low level (e.g., time slot (N)). In this manner, the low-distortion characteristic of line B of  FIG. 3  is achieved. 
   Various specific circuits that may be used within the power control logic  215  of  FIG. 2  to control the application of power to the amplifier stages are shown in  FIG. 5 ,  FIG. 6 , and  FIG. 7 , respectively. 
   Referring first to  FIG. 5 , a DC supply voltage V SUPPLY  is applied to the emitter of a PNP bipolar transistor Q in common-emitter configuration. The DC supply voltage may be unregulated or, alternatively, may have been regulated/conditioned to an appropriate DC level for a desired instantaneous output power using, for example, a switching power supply in combination with a linear regulator as described in greater detail in the aforementioned patent applications. The collector of the transistor Q is connected through a resistive divider network R 1 , R 2  to ground. An operational amplifier  501  is connected to receive a power-setting command signal  523  on a negative input and to receive on its positive input a voltage developed at the junction of the resistors R 1  and R 2 . The operational amplifier  501  produces an output signal that is applied to the base of the transistor Q. In operation, the transistor functions as a controlled resistance, under control of the operational amplifier  501 , to deliver a precisely-controlled voltage to multiple amplifier stages, including, for example, a driver stage  503  (responsive to an RF signal  509  analogous to signal  209  of  FIG. 2 ) and a final stage  505 . In the case of the driver stage  503 , the controlled voltage from the transistor Q is applied through a resistor R 3  to account for the sizing of the driver amplifier relative to the final amplifier. The foregoing circuit realizes fast control and may be used in conjunction with or in lieu of separate DC regulation circuitry. 
   One or more additional driver stages may be provided as shown, for example, in  FIG. 6 . In  FIG. 6 , the supply voltage of an initial stage  607  is controlled less stringently. A number of discrete supply voltages (V 1 , V 2 , . . . , V N ) are applied to a switch  609 , which is controlled to select a desired one of the discrete voltages. Control of the final stage  605  and the immediately preceding driver stage  603  may remain as previously described. 
   If a desired output signal has a large dynamic range, common control of the driver and final stages may prove insufficient. Referring to  FIG. 7 , separate control is provided for each of multiple amplifier stages. This arrangement may be extended to any arbitrary number of stages. 
   Referring again to  FIG. 2 , in the case of constant amplitude output signals, the amplifier as shown is effective to provide efficient amplification and power control. However, it does not provide for amplitude modulation. 
   Referring now to  FIG. 8 , a generalized efficient power amplifier structure is shown, enabling control of multiple stages to achieve complex control, including amplitude modulation, of an amplifier output signal. In  FIG. 8 , an RF input signal, RF in , is applied to an amplifier chain including N stages. The amplifier chain produces an RF output signal, RF out . Supply voltages for each of the stages are independently controlled. One or more control blocks receive a DC supply voltage and, responsive to control signals from a controller (not shown), produce separate power supply voltages for each of the N amplifier stages. In the example of  FIG. 8 , two control blocks are shown, a power/burst control block  801  and a modulation control block  803 . However, the functions of the control blocks may be readily consolidated or sub-divided as will be apparent to one of ordinary skill in the art. 
   Optionally, independent bias signals may be supplied to each one of the stages. In one embodiment, possible values of the bias signal include a value that turns the stage off, e.g., places the active element of the stage in a high-impedance state. In addition, each stage may optionally include a controlled bypass element or network, shown in  FIG. 8  as a resistor connecting the input and output terminals of a stage. Such a bypass may allow performance of an amplifier stage at low input signal levels to be more completely characterized and controlled. In particular, since circuit parasitics unavoidably create the effect of a bypass, by explicitly providing a bypass, it may be designed in such a manner as to dominate parasitic effects. 
   A particular case of the generalized amplifier structure of  FIG. 8  will now be described in detail. 
   Referring to  FIG. 9 , an amplifier is shown that provides the advantages of the amplifier of  FIG. 2  and additionally provides for amplitude modulation. In  FIG. 9 , there is provided a switch  901 , a DC supply  903 , a resonant network  905 , a load  907 , a control signal  909 , a control signal amplifier  911 , an output signal  913  and power control logic  915 , corresponding generally to and given like designations as elements in  FIG. 2 . The control signal amplifier  911  is responsive to a drive control signal  918  to produce a switch control signal  919  In  FIG. 9 , however, there is additionally provided an amplitude modulator  917  responsive to an AM signal  923 . Instead of the power control logic  915  controlling the control signal amplifier  911  directly (as in  FIG. 2 ), the power control logic  915  is coupled to the amplitude modulator  917 , which is responsive to the power control logic  915  to control the control signal amplifier  911 . Under the control of the amplitude modulator  917 , the control signal amplifier  911  produces a switch control signal  919  that is applied to the switch  901 . The DC supply  903  is coupled to the amplitude modulator  917 , which is responsive to the AM signal  923  to modify the supply voltage appropriately and apply a resulting switch supply signal  921  to the switch  901 . 
   Two cases of operation of the amplifier of  FIG. 9  may be distinguished. One case is shown in  FIG. 10 , in which amplitude modulation is achieved solely through variation of the switch supply signal  921 , and power control is achieved jointly through variation of the DC supply  903  and variation of the switch control signal  919  (via signal  918 ). During a timeslot (N−1), the peak amplitude of the switch control signal  919  remains constant. During this time, the peak amplitude of the control signal  909  also remains constant. The switch supply signal  921 , on the other hand, has impressed upon it amplitude modulation signal variations. As a result, the output signal  913  exhibits corresponding amplitude variations. During timeslot (N), the amplitudes of the control signal  909  and the switch control signal  919  are constant at a lower level, and a DC supply voltage  904  (not shown in  FIG. 10 ) is also constant at a lower level, indicative of a lower desired output power level. Different amplitude modulation signal variations are impressed upon the switch supply signal  921  and are manifest in the amplitude of the output signal  913 . During timeslot (N+1), the level of the control signal  909  and the switch control signal  919  are raised back up, as is the DC supply voltage  904 , corresponding to a higher desired output power level. The constant peak amplitude of the switch control signal  919  is set higher for higher desired output power levels, and set lower for lower desired output power levels, so that the switch  901  is successfully turned on and off as needed while minimizing the undesirable leakage of the switch control signal  919  through the switch  901  and into the resonant network  905 . 
   At lower power levels, to avoid excess leakage of the switch control signal  919  into the output signal  913 , it may be necessary to achieve amplitude modulation of the output signal through coordinated variation of both the switch supply signal  921  and the switch control signal  919 . This represents the second case of operation previously referred to, and is illustrated in  FIG. 11 . In particular,  FIG. 11  shows examples of different relationships between amplitude modulation of the switch supply signal  921  and amplitude modulation of the switch control signal  919 . Power control and amplitude modulation of both the switch supply signal  921  and the switch control signal  919  are applied as needed to extend the dynamic range of the output signal  913 . In an exemplary embodiment, amplitude modulation of the switch control signal  919  is applied only when the AM signal  923  dips below a threshold that is power-level dependent. 
   Timeslot (N−1) illustrates the case in which the AM signal  923  is below the power-level-dependent threshold (indicated in dashed lines in the upper frame of the  FIG. 11 ) for the duration of the timeslot. Hence, the switch control signal  919  is amplitude modulated along with the switch supply signal  921  throughout the duration of the timeslot. In timeslot (N), during both an initial portion of the timeslot and during a final portion of the timeslot, the AM signal  923  is assumed to be above the threshold. Hence, during these portions of the timeslot, the switch control signal  919  is not amplitude modulated. (In the middle frame of  FIG. 11 , the dashed lines indicate the nominal amplitude of the switch control signal  919  when the AM signal  923  is above the threshold.) During an intermediate portion of the timeslot, however, the AM signal  923  is assumed to be below the threshold. During this portion of the timeslot, the switch control signal  919  is amplitude modulated along with the switch supply signal  921 . Finally, in timeslot (N+1), the AM signal  923  is assumed to be above the threshold throughout the duration of the timeslot. The amplitude (peak-to-peak) of the switch control signal  919  is therefore held constant throughout the duration of the timeslot. Note that the actual amplitude modulation is still solely impressed on the output signal  913  by switch supply signal  921 . Variation of signal  918  and the resulting variation of signal  919  in concert with signal  921  is performed soley to reduce leakage. As such, the precision required of signal  918  is greatly reduced from that required of signal  921 . 
   Referring now to  FIG. 12 , a more detailed diagram is shown of an amplifier in accordance with an exemplary embodiment of the invention, in which like elements are assigned like reference numerals as in  FIG. 9 . In the embodiment of  FIG. 12 , the control signal amplifier  1211  and the switch  1201  are provided as first and second amplifier stages, a “gain” stage and a “switch” stage, respectively. The gain stage  211  may be implemented in a variety of ways. One implementation is a conventional gain-controlled linear CCS (controlled current source) amplifier of widely-understood classes A, AB, B and C. An alternative implementation is a smaller-scale switch-mode stage of a type described in the aforementioned copending U.S. applications. 
   Within dashed line block  917  are shown further details of one embodiment of the amplitude modulator  917  of  FIG. 9 . In response to AM signal samples  1223  and to a signal  1232  from the power control logic  1215 , the AM logic  1231  calculates appropriate supply levels for the first amplifier stage  1211  and the second amplifier stage  1201 . 
   In the case of the first amplifier stage  1211 , a DC supply voltage is supplied through a transistor  1235 - 1 . Base drive to the transistor  1235 - 1  is controlled by the AM logic  1231  through a DAC (digital to analog converter)  1233 - 1 . Hence the DAC  1233 - 1  sets the level of the switch control signal  1219  seen by the second amplifier stage  1201 . Similarly, in the case of the second amplifier stage  1201 , a DC supply voltage is supplied through a transistor  1235 - 2 . Base drive to the transistor  1235 - 2  is controlled by the AM logic  1231  through a DAC  1233 - 2 . 
   In an exemplary embodiment, the output of the DAC  1233 - 1  is given by the following rule: 
                       DAC   1     ⁡     (   t   )       =       ⁢     v   ⁢     (   p   )         ,               ⁢       for   ⁢           ⁢     a   ⁡     (   t   )         ≥     m   ⁡     (   p   )                       =       ⁢       v   ⁡     (   p   )       ·       a   ⁡     (   t   )         m   ⁡     (   p   )             ,               ⁢       for   ⁢           ⁢     a   ⁡     (   t   )         &lt;     m   ⁡     (   p   )                     
where a(t) is the AM signal at time t, m(p) is a threshold dependent on the power level p, and v(p) is the nominal output voltage of DAC 1 , for power level p.
 
   Operation of the amplifier of  FIG. 12  in accordance with the foregoing rule is illustrated in  FIG. 13 . As seen therein, as the signal a(t) (the amplitude of the AM signal at time t) fluctuates, for a first period of time, the signal exceeds the threshold m(p) for the current power level p. During this period, the voltage DAC 1 (t) is set to the nominal level v(p). Thereafter, the signal a(t) dips below the threshold for a period of time. During this period of time, the voltage DAC 1 (t) is amplitude modulated in accordance with the fluctuations of the signal a(t). When the signal a(t) again rises above the threshold, the voltage DAC 1 (t) is again set to the nominal level. 
   Thus, there has been described an efficient amplifier for RF signals that provides for amplitude modulation over a wide dynamic range. The amplitude of the switch control signal is adjusted to reduce the undesirable leakage effect. As a result, it becomes possible to produce output signals having average power anywhere within a wide-range, or to greatly increase the dynamic range over which amplitude modulation may be produced at a given average power level, or both. 
   It will be apparent to those of ordinary skill in the art that the present invention can be embodied in other specific forms without departing from the spirit or essential character thereof. The described embodiments are therefore intended to be in all respects illustrative and not restrictive. The scope of the invention is indicated by the appended claims, rather than the foregoing description, and all changes which come within the meaning and range of equivalents thereof are intended to be embraced therein.

Technology Category: 5