Patent Document

[0001]     This application claims priority under 35 U.S.C. § 120 as a divisional of U.S. patent application Ser. No. 10/010,601, filed Dec. 6, 2001, entitled “SYSTEMS AND METHODS FOR WIRELESS COMMUNICATION OVER A WIDE BANDWIDTH CHANNEL USING A PLURALITY OF SUB-CHANNELS.” 
         [0002]     This application may be related to the following U.S. patent applications: Ser. No. 10/961,592, filed Oct. 8, 2004; Ser. No. 10/961,614, filed Oct. 8, 2004; Ser. No. 10/963,026, filed Oct. 12, 2004; Ser. No. 10/962,935, filed Oct. 12, 2004; Ser. No. 10/964,482, filed Oct. 13, 2004; Ser. No. 10/964,336, filed Oct. 13, 2004; Ser. No. 10/984,436, filed Nov. 8, 2004; Ser. No. 10/985,977, filed Nov. 9, 2004; Ser. No. 10/985,861, filed Nov. 10, 2004; Ser. No. 10/988,373, filed Nov. 12, 2004; Ser. No. 11/055,525, filed Feb. 9, 2005; Ser. No. 10/120,456, filed Apr. 9, 2002; Ser. No. 10/810,948, filed Mar. 25, 2004; Ser. No. 10/811,223, filed Mar. 26, 2004; Ser. No. 10/934,316, filed Sep. 3, 2004; Ser. No. 10/948,099, filed Sep. 23, 2004; Ser. No. 10/810,410, filed Mar. 26, 2004; Ser. No. 10/952,458, filed Sep. 27, 2004; and Ser. No. 11/332,946, filed Jan. 17, 2006.  
     
    
     BACKGROUND OF THE INVENTION  
       [0003]     1. Field of the Invention  
         [0004]     The invention relates generally to communications, and more particularly to systems and methods for high data rate communications.  
         [0005]     2. Background  
         [0006]     Wireless communication systems are proliferating at the Wide Area Network (WAN), Local Area Network (LAN), and Personal Area Network (PAN) levels. These wireless communication systems use a variety of techniques to allow simultaneous access to multiple users. The most common of these techniques are Frequency Division Multiple Access (FDMA), which assigns specific frequencies to each user, Time Division Multiple Access (TDMA), which assigns particular time slots to each user, and Code Division Multiple Access (CDMA), which assigns specific codes to each user. But these wireless communication systems and various modulation techniques are afflicted by a host of problems that limit the capacity and the quality of service provided to the users. The following paragraphs briefly describe a few of these problems for the purpose of illustration.  
         [0007]     One problem that can exist in a wireless communication system is multipath interference. Multipath interference, or multipath, occurs because some of the energy in a transmitted wireless signal bounces off of obstacles, such as buildings or mountains, as it travels from source to destination. The obstacles in effect create reflections of the transmitted signal and the more obstacles there are, the more reflections they generate. The reflections then travel along their own transmission paths to the destination (or receiver). The reflections will contain the same information as the original signal; however, because of the differing transmission path lengths, the reflected signals will be out of phase with the original signal. As a result, they will often combine destructively with the original signal in the receiver. This is referred to as fading. To combat fading, current systems typically try to estimate the multipath effects and then compensate for them in the receiver using an equalizer. In practice, however, it is very difficult to achieve effective multipath compensation.  
         [0008]     A second problem that can affect the operation of wireless communication systems is interference from adjacent communication cells within the system. In FDMA/TDMA systems, this type of interference is prevent through a frequency reuse plan. Under a frequency reuse plan, available communication frequencies are allocated to communication cells within the communication system such that the same frequency will not be used in adjacent cells. Essentially, the available frequencies are split into groups. The number of groups is termed the reuse factor. Then the communication cells are grouped into clusters, each cluster containing the same number of cells as there are frequency groups. Each frequency group is then assigned to a cell in each cluster. Thus, if a frequency reuse factor of 7 is used, for example, then a particular communication frequency will be used only once in every seven communication cells. Thus, in any group of seven communication cells, each cell can only use 1/7 th  of the available frequencies, i.e., each cell is only able to use 1/7 th  of the available bandwidth.  
         [0009]     In a CDMA communication system, each cell uses the same wideband communication channel. In order to avoid interference with adjacent cells, each communication cell uses a particular set of spread spectrum codes to differentiate communications within the cell from those originating outside of the cell. Thus, CDMA systems preserve the bandwidth in the sense that they avoid reuse planning. But as will be discussed, there are other issues that limit the bandwidth in CDMA systems as well.  
         [0010]     Thus, in overcoming interference, system bandwidth is often sacrificed. Bandwidth is becoming a very valuable commodity as wireless communication systems continue to expand by adding more and more users. Therefore, trading off bandwidth for system performance is a costly, albeit necessary, proposition that is inherent in all wireless communication systems.  
         [0011]     The foregoing are just two examples of the types of problems that can affect conventional wireless communication systems. The examples also illustrate that there are many aspects of wireless communication system performance that can be improved through systems and methods that, for example, reduce interference, increase bandwidth, or both.  
         [0012]     Not only are conventional wireless communication systems effected by problems, such as those described in the preceding paragraphs, but also different types of systems are effected in different ways and to different degrees. Wireless communication systems can be split into three types: 1) line-of-sight systems, which can include point-to-point or point-to-multipoint systems; 2) indoor non-line of sight systems; and 3) outdoor systems such as wireless WANs. Line-of-sight systems are least affected by the problems described above, while indoor systems are more affected, due for example to signals bouncing off of building walls. Outdoor systems are by far the most affected of the three systems. Because these types of problems are limiting factors in the design of wireless transmitters and receivers, such designs must be tailored to the specific types of system in which it will operate. In practice, each type of system implements unique communication standards that address the issues unique to the particular type of system. Even if an indoor system used the same communication protocols and modulation techniques as an outdoor system, for example, the receiver designs would still be different because multipath and other problems are unique to a given type of system and must be addressed with unique solutions. This would not necessarily be the case if cost efficient and effective methodologies can be developed to combat such problems as described above that build in programmability so that a device can be reconfigured for different types of systems and still maintain superior performance.  
       SUMMARY OF THE INVENTION  
       [0013]     In order to combat the above problems, systems and methods of ultra-wideband communication are provided. In one ultra-wideband communication method, a communication channel is divided into a plurality of non-overlapping sub-channels by dividing a single serial message intended for an ultra-wideband communication device into a plurality of parallel messages. Each of the plurality of parallel messages are then encoded onto at least some of the plurality of sub-channels, and then transmitted.  
         [0014]     Other aspects, advantages, and novel features of the invention will become apparent from the following Detailed Description of Preferred Embodiments, when considered in conjunction with the accompanying drawings. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0015]     Preferred embodiments of the present inventions taught herein are illustrated by way of example, and not by way of limitation, in the figures of the accompanying drawings, in which:  
         [0016]      FIG. 1  is a diagram illustrating an example embodiment of a wideband channel divided into a plurality of sub-channels in accordance with the invention;  
         [0017]      FIG. 2  is a diagram illustrating the effects of multipath in a wireless communication system;  
         [0018]      FIG. 3  is a diagram illustrating another example embodiment of a wideband communication channel divided into a plurality of sub-channels in accordance with the invention;  
         [0019]      FIG. 4  is a diagram illustrating the application of a roll-off factor to the sub-channels of  FIGS. 1 and 2 ;  
         [0020]      FIG. 5A  is a diagram illustrating the assignment of sub-channels for a wideband communication channel in accordance with the invention;  
         [0021]      FIG. 5B  is a diagram illustrating the assignment of time slots for a wideband communication channel in accordance with the invention;  
         [0022]      FIG. 6  is a diagram illustrating an example embodiment of a wireless communication in accordance with the invention;  
         [0023]      FIG. 7  is a diagram illustrating the use of synchronization codes in the wireless communication system of  FIG. 5  in accordance with the invention;  
         [0024]      FIG. 8  is a diagram illustrating a correlator that can be used to correlate synchronization codes in the wireless communication system of  FIG. 5 ;  
         [0025]      FIG. 9  is a diagram illustrating synchronization code correlation in accordance with the invention;  
         [0026]      FIG. 10  is a diagram illustrating the cross-correlation properties of synchronization codes configured in accordance with the invention;  
         [0027]      FIG. 11  is a diagram illustrating another example embodiment of a wireless communication system in accordance with the invention;  
         [0028]      FIG. 12A  is a diagram illustrating how sub-channels of a wideband communication channel according to the present invention can be grouped in accordance with the present invention;  
         [0029]      FIG. 12B  is a diagram illustrating the assignment of the groups of sub-channels of  FIG. 12A  in accordance with the invention;  
         [0030]      FIG. 13  is a diagram illustrating the group assignments of  FIG. 12B  in the time domain;  
         [0031]      FIG. 14  is a flow chart illustrating the assignment of sub-channels based on SIR measurements in the wireless communication system of  FIG. 11  in accordance with the invention;  
         [0032]      FIG. 15  is a logical block diagram of an example embodiment of transmitter configured in accordance with the invention;  
         [0033]      FIG. 16  is a logical block diagram of an example embodiment of a modulator configured in accordance with the present invention for use in the transmitter of  FIG. 15 ;  
         [0034]      FIG. 17  is a diagram illustrating an example embodiment of a rate controller configured in accordance with the invention for use in the modulator of  FIG. 16 ;  
         [0035]      FIG. 18  is a diagram illustrating another example embodiment of a rate controller configured in accordance with the invention for use in the modulator of  FIG. 16 ;  
         [0036]      FIG. 19  is a diagram illustrating an example embodiment of a frequency encoder configured in accordance with the invention for use in the modulator of  FIG. 16 ;  
         [0037]      FIG. 20  is a logical block diagram of an example embodiment of a TDM/FDM block configured in accordance with the invention for use in the modulator of  FIG. 16 ;  
         [0038]      FIG. 21  is a logical block diagram of another example embodiment of a TDM/FDM block configured in accordance with the invention for use in the modulator of  FIG. 16 ;  
         [0039]      FIG. 22  is a logical block diagram of an example embodiment of a frequency shifter configured in accordance with the invention for use in the modulator of  FIG. 16 ;  
         [0040]      FIG. 23  is a logical block diagram of a receiver configured in accordance with the invention;  
         [0041]      FIG. 24  is a logical block diagram of an example embodiment of a demodulator configured in accordance with the invention for use in the receiver of  FIG. 23 ;  
         [0042]      FIG. 25  is a logical block diagram of an example embodiment of an equalizer configured in accordance with the present invention for use in the demodulator of  FIG. 24 ; and  
         [0043]      FIG. 26  is a logical block diagram of an example embodiment of a wireless communication device configured in accordance with the invention. 
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0000]     1. Introduction  
         [0044]     In the following paragraphs, the present invention will be described in detail by way of example with reference to the attached drawings. While this invention is capable of embodiment in many different forms, there is shown in the drawings and will herein be described in detail specific embodiments, with the understanding that the present disclosure is to be considered as an example of the principles of the invention and not intended to limit the invention to the specific embodiments shown and described. That is, throughout this description, the embodiments and examples shown should be considered as exemplars, rather than as limitations on the present invention. As used herein, the “present invention” refers to any one of the embodiments of the invention described herein, and any equivalents. Furthermore, reference to various feature(s) of the “present invention” throughout this document does not mean that all claimed embodiments or methods must include the referenced feature(s).  
         [0045]     In order to improve wireless communication system performance and allow a single device to move from one type of system to another, while still maintaining superior performance, the systems and methods described herein provide various communication methodologies that enhance performance of transmitters and receivers with regard to various common problems that afflict such systems and that allow the transmitters and/or receivers to be reconfigured for optimal performance in a variety of systems. Accordingly, the systems and methods described herein define a channel access protocol that uses a common wideband communication channel for all communication cells. The wideband channel, however, is then divided into a plurality of sub-channels. Different sub-channels are then assigned to one or more users within each cell. But the base station, or service access point, within each cell transmits one message that occupies the entire bandwidth of the wideband channel. Each user&#39;s communication device receives the entire message, but only decodes those portions of the message that reside in sub-channels assigned to the user. For a point-to-point system, for example, a single user may be assigned all sub-channels and, therefore, has the full wide band channel available to them. In a wireless WAN, on the other hand, the sub-channels may be divided among a plurality of users.  
         [0046]     In the descriptions of example embodiments that follow, implementation differences, or unique concerns, relating to different types of systems will be pointed out to the extent possible. But it should be understood that the systems and methods described herein are applicable to any type of communication systems. In addition, terms such as communication cell, base station, service access point, etc. are used interchangeably to refer to the common aspects of networks at these different levels.  
         [0047]     To begin illustrating the advantages of the systems and methods described herein, one can start by looking at the multipath effects for a single wideband communication channel  100  of bandwidth B as shown in  FIG. 1 . Communications sent over channel  100  in a traditional wireless communication system will comprise digital data bits, or symbols, that are encoded and modulated onto a RF carrier that is centered at frequency f c  and occupies bandwidth B. Generally, the width of the symbols (or the symbol duration) T is defined as 1/B. Thus, if the bandwidth B is equal to 100 MHz, then the symbol duration T is defined by the following equation: 
 
 T= 1/B= 1/100 megahertz(MHZ)=10 nanoseconds(ns).  (1) 
 
         [0048]     When a receiver receives the communication, demodulates it, and then decodes it, it will recreate a stream  104  of data symbols  106  as illustrated in  FIG. 2 . But the receiver will also receive multipath versions  108  of the same data stream. Because multipath data streams  108  are delayed in time relative to the data stream  104  by delays d 1 , d 2 , d 3 , and d 4 , for example, they may combine destructively with data stream  104 .  
         [0049]     A delay spread d s  is defined as the delay from reception of data stream  104  to the reception of the last multipath data stream  108  that interferes with the reception of data stream  104 . Thus, in the example illustrated in  FIG. 2 , the delay spread d s  is equal to delay d 4 . The delay spread d s  will vary for different environments. An environment with a lot of obstacles will create a lot of multipath reflections. Thus, the delay spread d s  will be longer. Experiments have shown that for outdoor WAN type environments, the delay spread d s  can be as long as 20 microseconds. Using the 10 ns symbol duration of equation (1), this translates to 2000 symbols. Thus, with a very large bandwidth, such as 100 MHz, multipath interference can cause a significant amount of interference at the symbol level for which adequate compensation is difficult to achieve. This is true even for indoor environments. For indoor LAN type systems, the delay spread d s  is significantly shorter, typically about 1 microsecond. For a 10 ns symbol duration, this is equivalent to 100 symbols, which is more manageable but still significant.  
         [0050]     By segmenting the bandwidth B into a plurality of sub-channels  202 , as illustrated in  FIG. 2 , and generating a distinct data stream for each sub-channel, the multipath effect can be reduced to a much more manageable level. For example, if the bandwidth b of each sub-channel  202  is 500 KHz, then the symbol duration is 2 microseconds. Thus, the delay spread d s  for each sub-channel is equivalent to only 10 symbols (outdoor) or half a symbol (indoor). Thus, by breaking up a message that occupies the entire bandwidth B into discrete messages, each occupying the bandwidth b of sub-channels  202 , a very wideband signal that suffers from relatively minor multipath effects is created.  
         [0051]     Before discussing further features and advantages of using a wideband communication channel segmented into a plurality of sub-channels as described, certain aspects of the sub-channels will be explained in more detail. Referring back to  FIG. 3 , the overall bandwidth B is segmented into N sub-channels center at frequencies f o  to f N-1 . Thus, the sub-channel  202  that is immediately to the right of f c  is offset from f c  by b/2, where b is the bandwidth of each sub-channel  202 . The next sub-channel  202  is offset by 3b/2, the next by 5b/2, and so on. To the left of fc, each sub-channel  202  is offset by −b/2, −3b/ 2 , −5b/2, etc.  
         [0052]     Preferably, sub-channels  202  are non-overlapping as this allows each sub-channel to be processed independently in the receiver. To accomplish this, a roll-off factor is preferably applied to the signals in each sub-channel in a pulse-shaping step. The effect of such a pulse-shaping step is illustrated in  FIG. 2  by the non-rectangular shape of the pulses in each sub-channel  202 . Thus, the bandwidth b of each sub-channel can be represented by an equation such as the following: 
 
 b =(1 +r )/ T;   (2) 
        Where r=the roll-off factor; and 
            T=the symbol duration.    
               
 
         [0055]     Without the roll-off factor, i.e., b=1/T, the pulse shape would be rectangular in the frequency domain, which corresponds to a (sin x)/x function in the time domain. The time domain signal for a (sin x)/x signal  400  is shown in  FIG. 4  in order to illustrate the problems associated with a rectangular pulse shape and the need to use a roll-off factor.  
         [0056]     As can be seen, main lobe  402  comprises almost all of signal  400 . But some of the signal also resides in side lobes  404 , which stretch out indefinitely in both directions from main lobe  402 . Side lobes  404  make processing signal  400  much more difficult, which increases the complexity of the receiver. Applying a roll-off factor r, as in equation ( 2 ), causes signal  400  to decay faster, reducing the number of side lobes  404 . Thus, increasing the roll-off factor decreases the length of signal  400 , i.e., signal  400  becomes shorter in time. But including the roll-off factor also decreases the available bandwidth in each sub-channel  202 . Therefore, r must be selected so as to reduce the number of side lobes  404  to a sufficient number, e.g., 15, while still maximizing the available bandwidth in each sub-channel  202 .  
         [0057]     Thus, the overall bandwidth B for communication channel  200  is given by the following equation: 
 
 B=N (1 +r )/ T;   (3) 
 
or 
 
 B=M/T;   (4) 
 
Where 
 
 M =(1 +r ) N.   (5) 
 
         [0058]     For efficiency purposes related to transmitter design, it is preferable that r is chosen so that M in equation (5) is an integer. Choosing r so that M is an integer allows for more efficient transmitters designs using, for example, Inverse Fast Fourier Transform (IFFT) techniques. Since M=N+N(r), and N is always an integer, this means that r must be chosen so that N(r) is an integer. Generally, it is preferable for r to be between 0.1 and 0.5. Therefore, if N is 16, for example, then 0.5 could be selected for r so that N(r) is an integer. Alternatively, if a value for r is chosen in the above example so that N(r) is not an integer, B can be made slightly wider than M/T to compensate. In this case, it is still preferable that r be chosen so that N(r) is approximately an integer.  
         [0000]     2. Example Embodiment of a Wireless Communication System  
         [0059]     With the above in mind,  FIG. 6  illustrates an example communication system  600  comprising a plurality of cells  602  that each use a common wideband communication channel to communicate with communication devices  604  within each cell  602 . The common communication channel is a wideband communication channel as described above. Each communication cell  602  is defined as the coverage area of a base station, or service access point,  606  within the cell. One such base station  606  is shown for illustration in  FIG. 6 . For purposes of this specification and the claims that follow, the term base station will be used generically to refer to a device that provides wireless access to the wireless communication system for a plurality of communication devices, whether the system is a line of sight, indoor, or outdoor system.  
         [0060]     Because each cell  602  uses the same communication channel, signals in one cell  602  must be distinguishable from signals in adjacent cells  602 . To differentiate signals from one cell  602  to another, adjacent base stations  606  use different synchronization codes according to a code reuse plan. In  FIG. 6 , system  600  uses a synchronization code reuse factor of 4, although the reuse factor can vary depending on the application.  
         [0061]     Preferably, the synchronization code is periodically inserted into a communication from a base station  606  to a communication device  604  as illustrated in  FIG. 6 . After a predetermined number of data packets  702 , in this case two, the particular synchronization code  704  is inserted into the information being transmitted by each base station  606 . A synchronization code is a sequence of data bits known to both the base station  606  and any communication devices  604  with which it is communicating. The synchronization code allows such a communication device  604  to synchronize its timing to that of base station  606 , which, in turn, allows device  604  to decode the data properly. Thus, in cell  1  (see lightly shaded cells  602  in  FIG. 6 ), for example, synchronization code  1  (SYNC 1 ) is inserted into data stream  706 , which is generated by base station  606  in cell  1 , after every two packets  702 ; in cell  2  SYNC 2  is inserted after every two packets  702 ; in cell  3  SYNC 3  is inserted; and in cell  4  SYNC 4  is inserted. Use of the synchronization codes is discussed in more detail below.  
         [0062]     In  FIG. 5A , an example wideband communication channel  500  for use in communication system  600  is divided into 16 sub-channels  502 , centered at frequencies f O  to f 15 . A base station  606  at the center of each communication cell  602  transmits a single packet occupying the whole bandwidth B of wideband channel  500 . Such a packet is illustrated by packet  504  in  FIG. 5B . Packet  504  comprises sub-packets  506  that are encoded with a frequency offset corresponding to one of sub-channels  502 . Sub-packets  506  in effect define available time slots in packet  504 . Similarly, sub-channels  502  can be said to define available frequency bins in communication channel  500 . Therefore, the resources available in communication cell  602  are time slots  506  and frequency bins  502 , which can be assigned to different communication devices  604  within each cell  602 .  
         [0063]     Thus, for example, frequency bins  502  and time slots  506  can be assigned to 4 different communication devices  604  within a cell  602  as shown in  FIG. 5 . Each communication device  604  receives the entire packet  504 , but only processes those frequency bins  502  and/or timeslots  506  that are assigned to it. Preferably, each device  604  is assigned non-adjacent frequency bins  502 , as in  FIG. 5A . This way, if interference corrupts the information in a portion of communication channel  500 , then the effects are spread across all devices  604  within a cell  602 . Hopefully, by spreading out the effects of interference in this manner the effects are minimized and the entire information sent to each device  604  can still be recreated from the unaffected information received in other frequency bins. For example, if interference, such as fading, corrupted the information in bins f o -f 4 , then each user  1 - 4  loses one packet of data. But each user potentially receives three unaffected packets from the other bins assigned to them. Hopefully, the unaffected data in the other three bins provides enough information to recreate the entire message for each user. Thus, frequency diversity can be achieved by assigning non-adjacent bins to each of multiple users.  
         [0064]     Ensuring that the bins assigned to one user are separated by more than the coherence bandwidth ensures frequency diversity. As discussed above, the coherence bandwidth is approximately equal to 1/d s . For outdoor systems, where d s  is typically 1 microsecond, 1/d s =1/1 microsecond=1 Mega Hertz (MHz). Thus, the non-adjacent frequency bands assigned to a user are preferably separated by at least 1 MHz. It is even more preferable, however, if the coherence bandwidth plus some guard band to ensure sufficient frequency diversity separate the non-adjacent bins assigned to each user. For example, it is preferable in certain implementations to ensure that at least 5 times the coherence bandwidth, or 5 MHz in the above example, separates the non-adjacent bins.  
         [0065]     Another way to provide frequency diversity is to repeat blocks of data in frequency bins assigned to a particular user that are separated by more than the coherence bandwidth. In other words, if 4 sub-channels  202  are assigned to a user, then data block a can be repeated in the first and third sub-channels  202  and data block b can be repeated in the second and fourth sub-channels  202 , provided the sub-channels are sufficiently separated in frequency. In this case, the system can be said to be using a diversity length factor of 2. The system can similarly be configured to implement other diversity lengths, e.g., 3, 4, . . . , 1.  
         [0066]     It should be noted that spatial diversity can also be included depending on the embodiment. Spatial diversity can comprise transmit spatial diversity, receive spatial diversity, or both. In transmit spatial diversity, the transmitter uses a plurality of separate transmitters and a plurality of separate antennas to transmit each message. In other words, each transmitter transmits the same message in parallel. The messages are then received from the transmitters and combined in the receiver. Because the parallel transmissions travel different paths, if one is affected by fading, the others will likely not be affected. Thus, when they are combined in the receiver, the message should be recoverable even if one or more of the other transmission paths experienced severe fading.  
         [0067]     Receive spatial diversity uses a plurality of separate receivers and a plurality of separate antennas to receive a single message. If an adequate distance separates the antennas, then the transmission path for the signals received by the antennas will be different. Again, this difference in the transmission paths will provide imperviousness to fading when the signals from the receivers are combined.  
         [0068]     Transmit and receive spatial diversity can also be combined within a system such as system  600  so that two antennas are used to transmit and two antennas are used to receive. Thus, each base station  606  transmitter can include two antennas, for transmit spatial diversity, and each communication device  604  receiver can include two antennas, for receive spatial diversity. If only transmit spatial diversity is implemented in system  600 , then it can be implemented in base stations  606  or in communication devices  604 . Similarly, if only receive spatial diversity is included in system  600 , then it can be implemented in base stations  606  or communication devices  604 .  
         [0069]     The number of communication devices  604  assigned frequency bins  502  and/or time slots  506  in each cell  602  is preferably programmable in real time. In other words, the resource allocation within a communication cell  602  is preferably programmable in the face of varying external conditions, i.e., multipath or adjacent cell interference, and varying requirements, i.e., bandwidth requirements for various users within the cell. Thus, if user  1  requires the whole bandwidth to download a large video file, for example, then the allocation of bins  502  can be adjust to provide user  1  with more, or even all, of bins  502 . Once user  1  no longer requires such large amounts of bandwidth, the allocation of bins  502  can be readjusted among all of users  1 - 4 .  
         [0070]     It should also be noted that all of the bins assigned to a particular user can be used for both the forward and reverse link. Alternatively, some bins  502  can be assigned as the forward link and some can be assigned for use on the reverse link, depending on the implementation.  
         [0071]     To increase capacity, the entire bandwidth B is preferably reused in each communication cell  602 , with each cell  602  being differentiated by a unique synchronization code (see discussion below). Thus, system  600  provides increased immunity to multipath and fading as well as increased bandwidth due to the elimination of frequency reuse requirements.  
         [0000]     3. Synchronization  
         [0072]      FIG. 8  illustrates an example embodiment of a synchronization code correlator  800 . When a device  604  in cell  1  (see  FIG. 6 ), for example, receives an incoming communication from the cell  1  base station  606 , it compares the incoming data with SYNC 1  in correlator  800 . Essentially, the device scans the incoming data trying to correlate the data with the known synchronization code, in this case SYNC 1 . Once correlator  800  matches the incoming data to SYNC 1  it generates a correlation peak  804  at the output. Multipath versions of the data will also generate correlation peaks  806 , although these peaks  806  are generally smaller than correlation peak  804 . The device can then use the correlation peaks to perform channel estimation, which allows the device to adjust for the multipath using an equalizer. Thus, in cell  1 , if correlator  800  receives a data stream comprising SYNC 1 , it will generate correlation peaks  804  and  806 . If, on the other hand, the data stream comprises SYNC 2 , for example, then no peaks will be generated and the device will essentially ignore the incoming communication.  
         [0073]     Even though a data stream that comprises SYNC 2  will not create any correlation peaks, it can create noise in correlator  800  that can prevent detection of correlation peaks  804  and  806 . Several steps can be taken to prevent this from occurring. One way to minimize the noise created in correlator  800  by signals from adjacent cells  602 , is to configure system  600  so that each base station  606  transmits at the same time. This way, the synchronization codes can preferably be generated in such a manner that only the synchronization codes  704  of adjacent cell data streams, e.g., streams  708 ,  710 , and  712 , as opposed to packets  702  within those streams, will interfere with detection of the correct synchronization code  704 , e.g., SYNC 1 . The synchronization codes can then be further configured to eliminate or reduce the interference.  
         [0074]     For example, the noise or interference caused by an incorrect synchronization code is a function of the cross correlation of that synchronization code with respect to the correct code. The better the cross correlation between the two, the lower the noise level. When the cross correlation is ideal, then the noise level will be virtually zero as illustrated in  FIG. 9  by noise level  902 . Therefore, a preferred embodiment of system  600  uses synchronization codes that exhibit ideal cross correlation, i.e., zero. Preferably, the ideal cross correlation of the synchronization codes covers a period l that is sufficient to allow accurate detection of multipath  906  as well as multipath correlation peaks  904 . This is important so that accurate channel estimation and equalization can take place. Outside of period  1 , the noise level  908  goes up, because the data in packets  702  is random and will exhibit low cross correlation with the synchronization code, e.g., SYNC 1 . Preferably, period  1  is actually slightly longer then the multipath length in order to ensure that the multipath can be detected.  
         [0075]     a. Synchronization code generation  
         [0076]     Conventional systems use orthogonal codes to achieve cross correlation in correlator  800 . In system  600  for example, SYNC 1 , SYNC 2 , SYNC 3 , and SYNC 4 , corresponding to cells  1 - 4  (see lightly shaded cells  602  of  FIG. 5 ) respectively, will all need to be generated in such a manner that they will have ideal cross correlation with each other. In one embodiment, if the data streams involved comprise high and low data bits, then the value “1” can be assigned to the high data bits and “−1” to the low data bits. Orthogonal data sequences are then those that produce a “0” output when they are exclusively ORed (XORed) together in correlator  800 . The following example illustrates this point for orthogonal sequences 1 and 2:  
               sequence   ⁢           ⁢   1   ⁢     :               1   ⁢           ⁢   1     -     1   ⁢           ⁢   1                 sequence   ⁢           ⁢   2   ⁢     :               1   ⁢           ⁢   1   ⁢           ⁢   1     -   1           _       
               ⁢                                                                   ⁢         1   ⁢           ⁢   1     -   1   -   1     =   0                 
 
         [0077]     Thus, when the results of XORing each bit pair are added, the result is “0”.  
         [0078]     But in system  600 , for example, each code must have ideal, or zero, cross correlation with each of the other codes used in adjacent cells  602 . Therefore, in one example embodiment of a method for generating synchronization codes exhibiting the properties described above, the process begins by selecting a “perfect sequence” to be used as the basis for the codes. A perfect sequence is one that when correlated with itself produces a number equal to the number of bits in the sequence. For example:  
             Perfect   ⁢           ⁢   sequence   ⁢           ⁢   1   ⁢     :                       1   ⁢           ⁢   1     -     1   ⁢           ⁢   1                         ⁢       1   ⁢           ⁢   1     -     1   ⁢           ⁢   1               _             
               ⁢       1   ⁢           ⁢   1   ⁢           ⁢   1   ⁢           ⁢   1     =   4         
 
         [0079]     But each time a perfect sequence is cyclically shifted by one bit, the new sequence is orthogonal with the original sequence. Thus, for example, if perfect sequence 1 is cyclically shifted by one bit and then correlated with the original, the correlation produces a “0” as in the following example;  
             Perfect   ⁢           ⁢   sequence   ⁢           ⁢   1   ⁢     :                       1   ⁢           ⁢   1     -     1   ⁢           ⁢   1                         ⁢       1   ⁢           ⁢   1   ⁢           ⁢   1     -   1             _             
               ⁢         1   ⁢           ⁢   1     -   1   -   1     =   0         
 
         [0080]     If the perfect sequence 1 is again cyclically shifted by one bit, and again correlated with the original, then it will produce a “0”. In general, you can cyclically shift a perfect sequence by any number of bits up to its length and correlate the shifted sequence with the original to obtain a “0”.  
         [0081]     Once a perfect sequence of the correct length is selected, the first synchronization code is preferably generated in one embodiment by repeating the sequence 4 times. Thus, if perfect sequence 1 is being used, then a first synchronization code y would be the following: 
        y=1 1-11 11-11 11-11 11-11.        
 
         [0083]     Or in generic form: 
        y=x(0)x(1)x(2)x(3) x(0)x(1)x(2)x(3) x(0)x(1)x(2)x(3) x(0)x(1)x(2)x(3).        
 
         [0085]     For a sequence of length L:  
         [0086]     y=x(0)x(1) . . . x(L)x(0)x(1) . . . x(L)x(0)x(1) . . . x(L)x(0)x(1) . . . x(L)  
         [0087]     Repeating the perfect sequence allows correlator  800  a better opportunity to detect the synchronization code and allows generation of other uncorrelated frequencies as well. Repeating has the effect of sampling in the frequency domain. This effect is illustrated by the graphs in  FIG. 10 . Thus, in TRACE  1 , which corresponds to synchronization code y, a sample  1002  is generated every fourth sample bin  1000 . Each sample bin is separated by 1/(4L×T), where T is the symbol duration. Thus, in the above example, where L=4, each sample bin is separated by 1/(16×T) in the frequency domain. TRACES  2 - 4  illustrate the next three synchronization codes. As can be seen, the samples for each subsequent synchronization code are shifted by one sample bin relative to the samples for the previous sequence. Therefore, none of the sequences interfere with each other.  
         [0088]     To generate the subsequent sequences, corresponding to TRACES  2 - 4 , sequence y must be shifted in frequency. This can be accomplished using the following equation: 
 
 z   r ( m )= y ( m )*exp( j* 2*π*r*m/( n*L )),  (5) 
 
         [0089]     for r=1 to L (# of sequences) and m=0 to 4*L-1 (time); and  
         [0090]     where: z r (m)=each subsequent sequence; 
        y(m)=the first sequence; and     n=the number of times the sequence is repeated.        
 
         [0093]     It will be understood that multiplying by an exp(j2π(r*m/N)) factor, where N is equal to the number of times the sequence is repeated n multiplied by the length of the underlying perfect sequence L, in the time domain results in a shift in the frequency domain. Equation (5) results in the desired shift as illustrated in  FIG. 10  for each of synchronization codes  2 - 4 , relative to synchronization code  1 . The final step in generating each synchronization code is to append the copies of the last M samples, where M is the length of the multipath, to the front of each code. This is done to make the convolution with the multipath cyclic and to allow easier detection of the multipath.  
         [0094]     It should be noted that synchronization codes can be generated from more than one perfect sequence using the same methodology. For example, a perfect sequence can be generated and repeated four times and then a second perfect sequence can be generated and repeated four times to get a n factor equal to eight. The resulting sequence can then be shifted as described above to create the synchronization codes.  
         [0095]     b. Signal Measurements Using Synchronization Codes  
         [0096]     Therefore, when a communication device is at the edge of a cell, it will receive signals from multiple base stations and, therefore, will be decoding several synchronization codes at the same time. This can be illustrated with the help of  FIG. 11 , which illustrates another example embodiment of a wireless communication system  1100  comprising communication cells  1102 ,  1104 , and  1106  as well as communication device  1108 , which is in communication with base station  1110  of cell  1102  but also receiving communication from base stations  1112  and  1114  of cells  1104  and  1106 , respectively.  
         [0097]     If communications from base station  1110  comprise synchronization code SYNC 1  and communications from base station  1112  and  1114  comprise SYNC 2  and SYNC 3  respectively, then device  1108  will effectively receive the sum of these three synchronization codes. This is because, as explained above, base stations  1110 ,  1112 , and  1114  are configured to transmit at the same time. Also, the synchronization codes arrive at device  1108  at almost the same time because they are generated in accordance with the description above.  
         [0098]     Again as described above, the synchronization codes SYNC 1 , SYNC 2 , and SYNC 3  exhibit ideal cross correlation. Therefore, when device  1108  correlates the sum x of codes SYNC 1 , SYNC 2 , and SYNC 3 , the latter two will not interfere with proper detection of SYNC 1  by device  1108 . Importantly, the sum x can also be used to determine important signal characteristics, because the sum x is equal to the sum of the synchronization code signal in accordance with the following equation: 
 
 x =SYNC1+SYNC2+SYNC3.  (6) 
 
         [0099]     Therefore, when SYNC 1  is removed, the sum of SYNC 2  and SYNC 3  is left, as shown in the following: 
 
 x −SYNC1=SYNC2+SYNC3.  (7) 
 
         [0100]     The energy computed from the sum (SYNC 2 +SYNC 3 ) is equal to the noise or interference seen by device  1108 . Since the purpose of correlating the synchronization code in device  1106  is to extract the energy in SYNC 1 , device  1108  also has the energy in the signal from base station  1110 , i.e., the energy represented by SYNC 1 . Therefore, device  1106  can use the energy of SYNC 1  and of (SYNC 2 +SYNC 3 ) to perform a signal-to-interference measurement for the communication channel over which it is communicating with base station  1110 . The result of the measurement is preferably a signal-to-interference ratio (SIR). The SIR measurement can then be communicated back to base station  1110  for purposes that will be discussed below.  
         [0101]     The ideal cross correlation of the synchronization codes, also allows device  1108  to perform extremely accurate determinations of the Channel Impulse Response (CIR), or channel estimation, from the correlation produced by correlator  800 . This allows for highly accurate equalization using low cost, low complexity equalizers, thus overcoming a significant draw back of conventional systems.  
         [0000]     4. Sub-Channel Assignments  
         [0102]     As mentioned, the SIR as determined by device  1108  can be communicated back to base station  1110  for use in the assignment of channels  502 . In one embodiment, due to the fact that each sub-channel  502  is processed independently, the SIR for each sub-channel  502  can be measured and communicated back to base station  1110 . In such an embodiment, therefore, sub-channels  502  can be divided into groups and a SIR measurement for each group can be sent to base station  1110 . This is illustrated in  FIG. 12A , which shows a wideband communication channel  1200  segmented into sub-channels fo to f 5 . Sub-channels fo to f 15  are then grouped into 8 groups G 1  to G 8 . Thus, in one embodiment, device  1108  and base station  1110  communicate over a channel such as channel  1200 .  
         [0103]     Sub-channels in the same group are preferably separated by as many sub-channels as possible to ensure diversity. In  FIG. 12A  for example, sub-channels within the same group are 7 sub-channels apart, e.g., group G 1  comprises f 0  and f 8 . Device  1102  reports a SIR measurement for each of the groups G 1  to G 8 . These SIR measurements are preferably compared with a threshold value to determine which sub-channels groups are useable by device  1108 . This comparison can occur in device  1108  or base station  1110 . If it occurs in device  1108 , then device  1108  can simply report to base station  1110  which sub-channels groups are useable by device  1108 .  
         [0104]     SIR reporting will be simultaneously occurring for a plurality of devices within cell  1102 . Thus,  FIG. 12B  illustrates the situation where two communication devices corresponding to User  1  and User  2  report SIR levels above the threshold for groups G 1 , G 3 , G 5 , and G 7 . Base station  1110  preferably then assigns sub-channel groups to User  1  and User  2  based on the SIR reporting as illustrated in  FIG. 12B . When assigning the “good” sub-channel groups to User  1  and User  2 , base station  1110  also preferably assigns them based on the principles of frequency diversity. In  FIG. 12B , therefore, User  1  and User  2  are alternately assigned every other “good” sub-channel.  
         [0105]     The assignment of sub-channels in the frequency domain is equivalent to the assignment of time slots in the time domain. Therefore, as illustrated in  FIG. 13 , two users, User  1  and User  2 , receive packet  1302  transmitted over communication channel  1200 .  FIG. 13  also illustrated the sub-channel assignment of  FIG. 12B . While  FIGS. 12 and 13  illustrate sub-channel/time slot assignment based on SIR for two users, the principles illustrated can be extended for any number of users. Thus, a packet within cell  1102  can be received by 3 or more users. Although, as the number of available sub-channels is reduced due to high SIR, so is the available bandwidth. In other words, as available channels are reduced, the number of users that can gain access to communication channel  1200  is also reduced.  
         [0106]     Poor SIR can be caused for a variety of reasons, but frequently it results from a device at the edge of a cell receiving communication signals from adjacent cells. Because each cell is using the same bandwidth B, the adjacent cell signals will eventually raise the noise level and degrade SIR for certain sub-channels. In certain embodiments, therefore, sub-channel assignment can be coordinated between cells, such as cells  1102 ,  1104 , and  1106  in  FIG. 10 , in order to prevent interference from adjacent cells.  
         [0107]     Thus, if communication device  1108  is near the edge of cell  1102 , and device  1118  is near the edge of cell  1106 , then the two can interfere with each other. As a result, the SIR measurements that device  1108  and  1118  report back to base stations  1110  and  1114 , respectively, will indicate that the interference level is too high. Base station  1110  can then be configured to assign only the odd groups, i.e., G 1 , G 3 , G 5 , etc., to device  1108 , while base station  1114  can be configured to assign the even groups to device  1118 . The two devices  1108  and  1118  will then not interfere with each other due to the coordinated assignment of sub-channel groups.  
         [0108]     Assigning the sub-channels in this manner reduces the overall bandwidth available to devices  1108  and  1118 , respectively. In this case the bandwidth is reduced by a factor of two. But it should be remembered that devices operating closer to each base station  1110  and  1114 , respectively, will still be able to use all channels if needed. Thus, it is only devices, such as device  1108 , that are near the edge of a cell that will have the available bandwidth reduced. Contrast this with a CDMA system, for example, in which the bandwidth for all users is reduced, due to the spreading techniques used in such systems, by approximately a factor of 10 at all times. It can be seen, therefore, that the systems and methods for wireless communication over a wide bandwidth channel using a plurality of sub-channels not only improves the quality of service, but can also increase the available bandwidth significantly.  
         [0109]     When there are three devices  1108 ,  1118 , and  1116  near the edge of their respective adjacent cells  1102 ,  1104 , and  1106 , the sub-channels can be divided by three. Thus, device  1108 , for example, can be assigned groups G 1 , G 4 , etc., device  1118  can be assigned groups G 2 , G 5 , etc., and device  1116  can be assigned groups G 3 , G 6 , etc. In this case the available bandwidth for these devices, i.e., devices near the edges of cells  1102 ,  1104 , and  1106 , is reduced by a factor of 3, but this is still better than a CDMA system, for example.  
         [0110]     The manner in which such a coordinated assignment of sub-channels can work is illustrated by the flow chart in  FIG. 14 . First in step  1402 , a communication device, such as device  1108 , reports the SIR for all sub-channel groups G 1  to G 8 . The SIRs reported are then compared, in step  1404 , to a threshold to determine if the SIR is sufficiently low for each group. Alternatively, device  1108  can make the determination and simply report which groups are above or below the SIR threshold. If the SIR levels are good for each group, then base station  1110  can make each group available to device  1108 , in step  1406 . Periodically, device  1108  preferably measures the SIR level and updates base station  1110  in case the SIR as deteriorated. For example, device  1108  may move from near the center of cell  1102  toward the edge, where interference from an adjacent cell may affect the SIR for device  1108 .  
         [0111]     If the comparison in step  1404  reveals that the SIR levels are not good, then base station  1110  can be preprogrammed to assign either the odd groups or the even groups only to device  1108 , which it will do in step  1408 . Device  1108  then reports the SIR measurements for the odd or even groups it is assigned in step  1410 , and they are again compared to a SIR threshold in step  1412 .  
         [0112]     It is assumed that the poor SIR level is due to the fact that device  1108  is operating at the edge of cell  1102  and is therefore being interfered with by a device such as device  1118 . But device  1108  will be interfering with device  1118  at the same time. Therefore, the assignment of odd or even groups in step  1408  preferably corresponds with the assignment of the opposite groups to device  1118 , by base station  1114 . Accordingly, when device  1108  reports the SIR measurements for whichever groups, odd or even, are assigned to it, the comparison in step  1410  should reveal that the SIR levels are now below the threshold level. Thus, base station  1110  makes the assigned groups available to device  1108  in step  1414 . Again, device  1108  preferably periodically updates the SIR measurements by returning to step  1402 .  
         [0113]     It is possible for the comparison of step  1410  to reveal that the SIR levels are still above the threshold, which should indicate that a third device, e.g., device  1116  is still interfering with device  1108 . In this case, base station  1110  can be preprogrammed to assign every third group to device  1108  in step  1416 . This should correspond with the corresponding assignments of non-interfering channels to devices  1118  and  1116  by base stations  1114  and  1112 , respectively. Thus, device  1108  should be able to operate on the sub-channel groups assigned, i.e., G 1 , G 4 , etc., without undue interference. Again, device  1108  preferably periodically updates the SIR measurements by returning to step  1402 . Optionally, a third comparison step (not shown) can be implemented after step  1416 , to ensure that the groups assigned to device  1408  posses an adequate SIR level for proper operation. Moreover, if there are more adjacent cells, i.e., if it is possible for devices in a 4 th  or even a 5 th  adjacent cell to interfere with device  1108 , then the process of  FIG. 14  would continue and the sub-channel groups would be divided even further to ensure adequate SIR levels on the sub-channels assigned to device  1108 .  
         [0114]     Even though the process of  FIG. 14  reduces the bandwidth available to devices at the edge of cells  1102 ,  1104 , and  1106 , the SIR measurements can be used in such a manner as to increase the data rate and therefore restore or even increase bandwidth. To accomplish this, the transmitters and receivers used in base stations  1102 ,  1104 , and  1106 , and in devices in communication therewith, e.g., devices  1108 ,  1114 , and  1116  respectively, must be capable of dynamically changing the symbol mapping schemes used for some or all of the sub-channel. For example, in some embodiments, the symbol mapping scheme can be dynamically changed among BPSK, QPSK, 8PSK, 16QAM, 32QAM, etc. As the symbol mapping scheme moves higher, i.e., toward 32QAM, the SIR level required for proper operation moves higher, i.e., less and less interference can be withstood. Therefore, once the SIR levels are determined for each group, the base station, e.g., base station  1110 , can then determine what symbol mapping scheme can be supported for each sub-channel group and can change the modulation scheme accordingly. Device  1108  must also change the symbol mapping scheme to correspond to that of the base stations. The change can be effected for all groups uniformly, or it can be effected for individual groups. Moreover, the symbol mapping scheme can be changed on just the forward link, just the reverse link, or both, depending on the embodiment.  
         [0115]     Thus, by maintaining the capability to dynamically assign sub-channels and to dynamically change the symbol mapping scheme used for assigned sub-channels, the systems and methods described herein provide the ability to maintain higher available bandwidths with higher performance levels than conventional systems. To fully realize the benefits described, however, the systems and methods described thus far must be capable of implementation in a cost effect and convenient manner. Moreover, the implementation must include reconfigurability so that a single device can move between different types of communication systems and still maintain optimum performance in accordance with the systems and methods described herein. The following descriptions detail example high level embodiments of hardware implementations configured to operate in accordance with the systems and methods described herein in such a manner as to provide the capability just described above.  
         [0000]     5. Sample Transmitter Embodiments  
         [0116]      FIG. 15  is logical block diagram illustrating an example embodiment of a transmitter  1500  configured for wireless communication in accordance with the systems and methods described above. The transmitter could, for example be within a base station, e.g., base station  606 , or within a communication device, such as device  604 . Transmitter  1500  is provided to illustrate logical components that can be included in a transmitter configured in accordance with the systems and methods described herein. It is not intended to limit the systems and methods for wireless communication over a wide bandwidth channel using a plurality of sub-channels to any particular transmitter configuration or any particular wireless communication system.  
         [0117]     With this in mind, it can be seen that transmitter  1500  comprises a serial-to-parallel converter  1504  configured to receive a serial data stream  1502  comprising a data rate R. Serial-to-parallel converter  1504  converts data stream  1502  into N parallel data streams  1504 , where N is the number of sub-channels  202 . It should be noted that while the discussion that follows assumes that a single serial data stream is used, more than one serial data stream can also be used if required or desired. In any case, the data rate of each parallel data stream  1504  is then R/N. Each data stream  1504  is then sent to a scrambler, encoder, and interleaver block  1506 . Scrambling, encoding, and interleaving are common techniques implemented in many wireless communication transmitters and help to provide robust, secure communication. Examples of these techniques will be briefly explained for illustrative purposes.  
         [0118]     Scrambling breaks up the data to be transmitted in an effort to smooth out the spectral density of the transmitted data. For example, if the data comprises a long string of “1”s, there will be a spike in the spectral density. This spike can cause greater interference within the wireless communication system. By breaking up the data, the spectral density can be smoothed out to avoid any such peaks. Often, scrambling is achieved by XORing the data with a random sequence.  
         [0119]     Encoding, or coding, the parallel bit streams  1504  can, for example, provide Forward Error Correction (FEC). The purpose of FEC is to improve the capacity of a communication channel by adding some carefully designed redundant information to the data being transmitted through the channel. The process of adding this redundant information is known as channel coding. Convolutional coding and block coding are the two major forms of channel coding. Convolutional codes operate on serial data, one or a few bits at a time. Block codes operate on relatively large (typically, up to a couple of hundred bytes) message blocks. There are a variety of useful convolutional and block codes, and a variety of algorithms for decoding the received coded information sequences to recover the original data. For example, convolutional encoding or turbo coding with Viterbi decoding is a FEC technique that is particularly suited to a channel in which the transmitted signal is corrupted mainly by additive white gaussian noise (AWGN) or even a channel that simply experiences fading.  
         [0120]     Convolutional codes are usually described using two parameters: the code rate and the constraint length. The code rate, k/n, is expressed as a ratio of the number of bits into the convolutional encoder (k) to the number of channel symbols (n) output by the convolutional encoder in a given encoder cycle. A common code rate is ½, which means that 2 symbols are produced for every 1-bit input into the coder. The constraint length parameter, K, denotes the “length” of the convolutional encoder, i.e. how many k-bit stages are available to feed the combinatorial logic that produces the output symbols. Closely related to K is the parameter m, which indicates how many encoder cycles an input bit is retained and used for encoding after it first appears at the input to the convolutional encoder. The m parameter can be thought of as the memory length of the encoder.  
         [0121]     Interleaving is used to reduce the effects of fading. Interleaving mixes up the order of the data so that if a fade interferes with a portion of the transmitted signal, the overall message will not be affected. This is because once the message is de-interleaved and decoded in the receiver, the data lost will comprise non-contiguous portions of the overall message. In other words, the fade will interfere with a contiguous portion of the interleaved message, but when the message is de-interleaved, the interfered with portion is spread throughout the overall message. Using techniques such as FEC, the missing information can then be filled in, or the impact of the lost data may just be negligible.  
         [0122]     After blocks  1506 , each parallel data stream  1504  is sent to symbol mappers  1508 . Symbol mappers  1508  apply the requisite symbol mapping, e.g., BPSK, QPSK, etc., to each parallel data stream  1504 . Symbol mappers  1508  are preferably programmable so that the modulation applied to parallel data streams can be changed, for example, in response to the SIR reported for each sub-channel  202 . It is also preferable, that each symbol mapper  1508  be separately programmable so that the optimum symbol mapping scheme for each sub-channel can be selected and applied to each parallel data stream  1504 .  
         [0123]     After symbol mappers  1508 , parallel data streams  1504  are sent to modulators  1510 . Important aspects and features of example embodiments of modulators  1510  are described below. After modulators  1510 , parallel data streams  1504  are sent to summer  1512 , which is configured to sum the parallel data streams and thereby generate a single serial data stream  1518  comprising each of the individually processed parallel data streams  1504 . Serial data stream  1518  is then sent to radio module  1512 , where it is modulated with an RF carrier, amplified, and transmitted via antenna  1516  according to known techniques.  
         [0124]     The transmitted signal occupies the entire bandwidth B of communication channel  100  and comprises each of the discrete parallel data streams  1504  encoded onto their respective sub-channels  102  within bandwidth B. Encoding parallel data streams  1504  onto the appropriate sub-channels  102  requires that each parallel data stream  1504  be shifted in frequency by an appropriate offset. This is achieved in modulator  1510 .  
         [0125]      FIG. 16  is a logical block diagram of an example embodiment of a modulator  1600  in accordance with the systems and methods described herein. Importantly, modulator  1600  takes parallel data streams  1602  performs Time Division Modulation (TDM) or Frequency Division Modulation (FDM) on each data stream  1602 , filters them using filters  1612 , and then shifts each data stream in frequency using frequency shifter  1614  so that they occupy the appropriate sub-channel. Filters  1612  apply the required pulse shaping, i.e., they apply the roll-off factor described in section  1 . The frequency shifted parallel data streams  1602  are then summed and transmitted. Modulator  1600  can also include rate controller  1604 , frequency encoder  1606 , and interpolators  1610 . All of the components shown in  FIG. 15  are described in more detail in the following paragraphs and in conjunction with  FIGS. 16-22 .  
         [0126]      FIG. 17  illustrates one example embodiment of a rate controller  1700  in accordance with the systems and methods described herein. Rate control  1700  is used to control the data rate of each parallel data stream  1602 . In rate controller  1700 , the data rate is halved by repeating data streams d( 0 ) to d( 7 ), for example, producing streams a( 0 ) to a( 15 ) in which a( 0 ) is the same as a( 8 ), a( 1 ) is the same as a( 9 ), etc.  FIG. 17  also illustrates that the effect of repeating the data streams in this manner is to take the data streams that are encoded onto the first 8 sub-channels  1702 , and duplicate them on the next 8 sub-channels  1702 . As can be seen, 7 sub-channels separate sub-channels  1702  comprising the same, or duplicate, data streams. Thus, if fading effects one sub-channel  1702 , for example, the other sub-channels  1702  carrying the same data will likely not be effected, i.e., there is frequency diversity between the duplicate data streams. So by sacrificing data rate, in this case half the data rate, more robust transmission is achieved. Moreover, the robustness provided by duplicating the data streams d( 0 ) to d( 7 ) can be further enhanced by applying scrambling to the duplicated data streams via scramblers  1708 .  
         [0127]     It should be noted that the data rate can be reduced by more than half, e.g., by four or more. Alternatively, the data rate can also be reduced by an amount other than half. For example if information from n data stream is encoded onto m sub-channels, where m&gt;n. Thus, to decrease the rate by ⅔, information from one data stream can be encoded on a first sub-channel, information from a second data stream can be encoded on a second data channel, and the sum or difference of the two data streams can be encoded on a third channel. In which case, proper scaling will need to be applied to the power in the third channel. Otherwise, for example, the power in the third channel can be twice the power in the first two.  
         [0128]     Preferably, rate controller  1700  is programmable so that the data rate can be changed responsive to certain operational factors. For example, if the SIR reported for sub-channels  1702  is low, then rate controller  1700  can be programmed to provide more robust transmission via repetition to ensure that no data is lost due to interference. Additionally, different types of wireless communication system, e.g., indoor, outdoor, line-of-sight, may require varying degrees of robustness. Thus, rate controller  1700  can be adjusted to provide the minimum required robustness for the particular type of communication system. This type of programmability not only ensures robust communication, it can also be used to allow a single device to move between communication systems and maintain superior performance.  
         [0129]      FIG. 18  illustrates an alternative example embodiment of a rate controller  1800  in accordance with the systems and methods described. In rate controller  1800  the data rate is increased instead of decreased. This is accomplished using serial-to-parallel converters  1802  to convert each data streams d( 0 ) to d( 15 ), for example, into two data streams. Delay circuits  1804  then delay one of the two data streams generated by each serial-to-parallel converter  1802  by ½ a symbol. Thus, data streams d( 0 ) to d( 15 ) are transformed into data streams a( 0 ) to a( 31 ). The data streams generated by a particular serial-to-parallel converter  1802  and associate delay circuit  1804  must then be summed and encoded onto the appropriate sub-channel. For example, data streams a( 0 ) and a( 1 ) must be summed and encoded onto the first sub-channel. Preferably, the data streams are summed subsequent to each data stream being pulsed shaped by a filter  1612 .  
         [0130]     Thus, rate controller  1604  is preferably programmable so that the data rate can be increased, as in rate controller  1800 , or decreased, as in rate controller  1700 , as required by a particular type of wireless communication system, or as required by the communication channel conditions or sub-channel conditions. In the event that the data rate is increased, filters  1612  are also preferably programmable so that they can be configured to apply pulse shaping to data streams a( 0 ) to a( 31 ), for example, and then sum the appropriate streams to generate the appropriate number of parallel data streams to send to frequency shifter  1614 .  
         [0131]     The advantage of increasing the data rate in the manner illustrated in  FIG. 18  is that higher symbol mapping rates can essentially be achieved, without changing the symbol mapping used in symbol mappers  1508 . Once the data streams are summed, the summed streams are shifted in frequency so that they reside in the appropriate sub-channel. But because the number of bits per each symbol has been doubled, the symbol mapping rate has been doubled. Thus, for example, a 4QAM symbol mapping can be converted to a 16QAM symbol mapping, even if the SIR is too high for 16QAM symbol mapping to otherwise be applied. In other words, programming rate controller  1800  to increase the data rate in the manner illustrated in  FIG. 18  can increase the symbol mapping even when channel conditions would otherwise not allow it, which in turn can allow a communication device to maintain adequate or even superior performance regardless of the type of communication system.  
         [0132]     The draw back to increasing the data rate as illustrated in  FIG. 18  is that interference is increased, as is receiver complexity. The former is due to the increased amount of data. The latter is due to the fact that each symbol cannot be processed independently because of the ½ symbol overlap. Thus, these concerns must be balanced against the increase symbol mapping ability when implementing a rate controller such as rate controller  1800 .  
         [0133]      FIG. 19  illustrates one example embodiment of a frequency encoder  1900  in accordance with the systems and methods described herein. Similar to rate encoding, frequency encoding is preferably used to provide increased communication robustness. In frequency encoder  1900  the sum or difference of multiple data streams are encoded onto each sub-channel. This is accomplished using adders  1902  to sum data streams d( 0 ) to d( 7 ) with data streams d( 8 ) to d( 15 ), respectively, while adders  1904  subtract data streams d( 0 ) to d( 7 ) from data streams d( 8 ) to d( 15 ), respectively, as shown. Thus, data streams a( 0 ) to a( 15 ) generated by adders  1902  and  1904  comprise information related to more than one data streams d( 0 ) to d( 15 ). For example, a( 0 ) comprises the sum of d( 0 ) and d( 8 ), i.e., d( 0 )+d( 8 ), while a( 8 ) comprises d( 8 )-d( 0 ). Therefore, if either a( 0 ) or a( 8 ) is not received due to fading, for example, then both of data streams d( 0 ) and d( 8 ) can still be retrieved from data stream a( 8 ).  
         [0134]     Essentially, the relationship between data stream d( 0 ) to d( 15 ) and a( 0 ) to a( 15 ) is a matrix relationship. Thus, if the receiver knows the correct matrix to apply, it can recover the sums and differences of d( 0 ) to d( 15 ) from a( 0 ) to a( 15 ). Preferably, frequency encoder  1900  is programmable, so that it can be enabled and disabled in order to provided robustness when required. Preferable, adders  1902  and  1904  are programmable also so that different matrices can be applied to d( 0 ) to d( 15 ).  
         [0135]     After frequency encoding, if it is included, data streams  1602  are sent to TDM/FDM blocks  1608 . TDM/FDM blocks  1608  perform TDM or FDM on the data streams as required by the particular embodiment.  FIG. 20  illustrates an example embodiment of a TDM/FDM block  2000  configured to perform TDM on a data stream. TDM/FDM block  2000  is provided to illustrate the logical components that can be included in a TDM/FDM block configured to perform TDM on a data stream. Depending on the actual implementation, some of the logical components may or may not be included. TDM/FDM block  2000  comprises a sub-block repeater  2002 , a sub-block scrambler  2004 , a sub-block terminator  2006 , a sub-block repeater  2008 , and a sync inserter  2010 .  
         [0136]     Sub-block repeater  2002  is configured to receive a sub-block of data, such as block  2012  comprising bits a( 0 ) to a( 3 ) for example. Sub-block repeater is then configured to repeat block  2012  to provide repetition, which in turn leads to more robust communication. Thus, sub-block repeater  2002  generates block  2014 , which comprises 2 blocks  2012 . Sub-block scrambler  2004  is then configured to receive block  2014  and to scramble it, thus generating block  2016 . One method of scrambling can be to invert half of block  2014  as illustrated in block  2016 . But other scrambling methods can also be implemented depending on the embodiment.  
         [0137]     Sub-block terminator  2006  takes block  2016  generated by sub-block scrambler  2004  and adds a termination block  2034  to the front of block  2016  to form block  2018 . Termination block  2034  ensures that each block can be processed independently in the receiver. Without termination block  2034 , some blocks may be delayed due to multipath, for example, and they would therefore overlap part of the next block of data. But by including termination block  2034 , the delayed block can be prevented from overlapping any of the actual data in the next block.  
         [0138]     Termination block  2034  can be a cyclic prefix termination  2036 . A cyclic prefix termination  2036  simply repeats the last few symbols of block  2018 . Thus, for example, if cyclic prefix termination  2036  is three symbols long, then it would simply repeat the last three symbols of block  2018 . Alternatively, termination block  2034  can comprise a sequence of symbols that are known to both the transmitter and receiver. The selection of what type of block termination  2034  to use can impact what type of equalizer is used in the receiver. Therefore, receiver complexity and choice of equalizers must be considered when determining what type of termination block  2034  to use in TDM/FDM block  2000 .  
         [0139]     After sub-block terminator  2006 , TDM/FDM block  2000  can include a sub-block repeater  2008  configured to perform a second block repetition step in which block  2018  is repeated to form block  2020 . In certain embodiments, sub-block repeater can be configured to perform a second block scrambling step as well. After sub-block repeater  2008 , if included, TDM/FDM block  2000  comprises a sync inserter  210  configured to periodically insert an appropriate synchronization code  2032  after a predetermined number of blocks  2020  and/or to insert known symbols into each block. The purpose of synchronization code  2032  is discussed in section  3 .  
         [0140]      FIG. 21 , on the other hand, illustrates an example embodiment of a TDM/FDM block  2100  configured for FDM, which comprises sub-block repeater  2102 , sub-block scrambler  2104 , block coder  2106 , sub-block transformer  2108 , sub-block terminator  2110 , and sync inserter  2112 . As with TDM/FDM block  2000 , sub-block repeater  2102  repeats block  2114  and generates block  2116 . Sub-block scrambler then scrambles block  2116 , generating block  2118 . Sub-block coder  2106  takes block  2118  and codes it, generating block  2120 . Coding block correlates the data symbols together and generates symbols b. This requires joint demodulation in the receiver, which is more robust but also more complex. Sub-block transformer  2108  then performs a transformation on block  2120 , generating block  2122 . Preferably, the transformation is an IFFT of block  2120 , which allows for more efficient equalizers to be used in the receiver. Next, sub-block terminator  2110  terminates block  2122 , generating block  2124  and sync inserter  2112  periodically inserts a synchronization code  2126  after a certain number of blocks  2124  and/or insert known symbols into each block. Preferably, sub-block terminator  2110  only uses cyclic prefix termination as described above. Again this allows for more efficient receiver designs.  
         [0141]     TDM/FDM block  2100  is provided to illustrate the logical components that can be included in a TDM/FDM block configured to perform FDM on a data stream. Depending on the actual implementation, some of the logical components may or may not be included. Moreover, TDM/FDM block  2000  and  2100  are preferably programmable so that the appropriate logical components can be included as required by a particular implementation. This allows a device that incorporates one of blocks  2000  or  2100  to move between different systems with different requirements. Further, it is preferable that TDM/FDM block  1608  in  FIG. 16  be programmable so that it can be programmed to perform TDM, such as described in conjunction with block  2000 , or FDM, such as described in conjunction with block  2100 , as required by a particular communication system.  
         [0142]     After TDM/FDM blocks  1608 , in  FIG. 16 , the parallel data streams are preferably passed to interpolators  1610 .  
         [0143]     After Interpolators  1610 , the parallel data streams are passed to filters  1612 , which apply the pulse shaping described in conjunction with the roll-off factor of equation (2) in section  1 . Then the parallel data streams are sent to frequency shifter  1614 , which is configured to shift each parallel data stream by the frequency offset associated with the sub-channel to which the particular parallel data stream is associated.  
         [0144]      FIG. 22  illustrates an example embodiment of a frequency shifter  2200  in accordance with the systems and methods described herein. As can be seen, frequency shifter  2200  comprises multipliers  2202  configured to multiply each parallel data stream by the appropriate exponential to achieve the required frequency shift. Each exponential is of the form: exp(j2πd c nT/rM), where c is the corresponding sub-channel, e.g., c=0 to N−1, and n is time. Preferably, frequency shifter  1614  in  FIG. 16  is programmable so that various channel/sub-channel configurations can be accommodated for various different systems. Alternatively, an IFFT block can replace shifter  1614  and filtering can be done after the IFFT block. This type of implementation can be more efficient depending on the implementation.  
         [0145]     After the parallel data streams are shifted, they are summed, e.g., in summer  1512  of  FIG. 15 . The summed data stream is then transmitted using the entire bandwidth B of the communication channel being used. But the transmitted data stream also comprises each of the parallel data streams shifted in frequency such that they occupy the appropriate sub-channel. Thus, each sub-channel may be assigned to one user, or each sub-channel may carry a data stream intended for different users. The assignment of sub-channels is described in section  3   b . Regardless of how the sub-channels are assigned, however, each user will receive the entire bandwidth, comprising all the sub-channels, but will only decode those sub-channels assigned to the user.  
         [0000]     6. Sample Receiver Embodiments  
         [0146]      FIG. 23  illustrates an example embodiment of a receiver  2300  that can be configured in accordance with the present invention. Receiver  2300  comprises an antenna  2302  configured to receive a message transmitted by a transmitter, such as transmitter  1500 . Thus, antenna  2302  is configured to receive a wide band message comprising the entire bandwidth B of a wide band channel that is divided into sub-channels of bandwidth b. As described above, the wide band message comprises a plurality of messages each encoded onto each of a corresponding sub-channel. All of the sub-channels may or may not be assigned to a device that includes receiver  2300 ; Therefore, receiver  2300  may or may not be required to decode all of the sub-channels.  
         [0147]     After the message is received by antenna  2300 , it is sent to radio receiver  2304 , which is configured to remove the carrier associated with the wide band communication channel and extract a baseband signal comprising the data stream transmitted by the transmitter. The baseband signal is then sent to correlator  2306  and demodulator  2308 . Correlator  2306  is configured to correlated with a synchronization code inserted in the data stream as described in section  3 . It is also preferably configured to perform SIR and multipath estimations as described in section  3 ( b ). Demodulator  2308  is configured to extract the parallel data streams from each sub-channel assigned to the device comprising receiver  2300  and to generate a single data stream therefrom.  
         [0148]      FIG. 24  illustrates an example embodiment of a demodulator  2400  in accordance with the systems and methods described herein. Demodulator  2402  comprises a frequency shifter  2402 , which is configured to apply a frequency offset to the baseband data stream so that parallel data streams comprising the baseband data stream can be independently processed in receiver  2400 . Thus, the output of frequency shifter  2402  is a plurality of parallel data streams, which are then preferably filtered by filters  2404 . Filters  2404  apply a filter to each parallel data stream that corresponds to the pulse shape applied in the transmitter, e.g., transmitter  1500 . Alternatively, an IFFT block can replace shifter  1614  and filtering can be done after the IFFT block. This type of implementation can be more efficient depending on the implementation.  
         [0149]     Next, receiver  2400  preferably includes decimators  2406  configured to decimate the data rate of the parallel bit streams. Sampling at higher rates helps to ensure accurate recreation of the data. But the higher the data rate, the larger and more complex equalizer  2408  becomes. Thus, the sampling rate, and therefore the number of samples, can be reduced by decimators  2406  to an adequate level that allows for a smaller and less costly equalizer  2408 .  
         [0150]     Equalizer  2408  is configured to reduce the effects of multipath in receiver  2300 . Its operation will be discussed more fully below. After equalizer  2408 , the parallel data streams are sent to de-scrambler, decoder, and de-interleaver  2410 , which perform the opposite operations of scrambler, encoder, and interleaver  1506  so as to reproduce the original data generated in the transmitter. The parallel data streams are then sent to parallel to serial converter  2412 , which generates a single serial data stream from the parallel data streams.  
         [0151]     Equalizer  2408  uses the multipath estimates provided by correlator  2306  to equalize the effects of multipath in receiver  2300 . In one embodiment, equalizer  2408  comprises Single-In Single-Out (SISO) equalizers operating on each parallel data stream in demodulator  2400 . In this case, each SISO equalizer comprising equalizer  2408  receives a single input and generates a single equalized output. Alternatively, each equalizer can be a Multiple-In Multiple-Out (MIMO) or a Multiple-In Single-Out (MISO) equalizer. Multiple inputs can be required for example, when a frequency encoder or rate controller, such as frequency encoder  1900 , is included in the transmitter. Because frequency encoder  1900  encodes information from more than one parallel data stream onto each sub-channel, each equalizers comprising equalizer  2408  need to equalize more than one sub-channel. Thus, for example, if a parallel data stream in demodulator  2400  comprises d( 1 )+d( 8 ), then equalizer  2408  will need to equalize both d( 1 ) and d( 8 ) together. Equalizer  2408  can then generate a single output corresponding to d( 1 ) or d( 8 ) (MISO) or it can generate both d( 1 ) and d( 8 ) (MIMO).  
         [0152]     Equalizer  2408  can also be a time domain equalizer (TDE) or a frequency domain equalizer (FDE) depending on the embodiment. Generally, equalizer  2408  is a TDE if the modulator in the transmitter performs TDM on the parallel data streams, and a FDE if the modulator performs FDM. But equalizer  2408  can be an FDE even if TDM is used in the transmitter. Therefore, the preferred equalizer type should be taken into consideration when deciding what type of block termination to use in the transmitter. Because of power requirements, it is often preferable to use FDM on the forward link and TDM on the reverse link in a wireless communication system.  
         [0153]     As with transmitter  1500 , the various components comprising demodulator  2400  are preferably programmable, so that a single device can operate in a plurality of different systems and still maintain superior performance, which is a primary advantage of the systems and methods described herein. Accordingly, the above discussion provides systems and methods for implementing a channel access protocol that allows the transmitter and receiver hardware to be reprogrammed slightly depending on the communication system.  
         [0154]     Thus, when a device moves from one system to another, it preferably reconfigures the hardware, i.e. transmitter and receiver, as required and switches to a protocol stack corresponding to the new system. An important part of reconfiguring the receiver is reconfiguring, or programming, the equalizer because multipath is a main problem for each type of system. The multipath, however, varies depending on the type of system, which previously has meant that a different equalizer is required for different types of communication systems. The channel access protocol described in the preceding sections, however, allows for equalizers to be used that need only be reconfigured slightly for operation in various systems.  
         [0155]     a. Sample Equalizer Embodiment  
         [0156]      FIG. 25  illustrates an example embodiment of a receiver  2500  illustrating one way to configure equalizers  2506  in accordance with the systems and methods described herein. Before discussing the configuration of receiver  2500 , it should be noted that one way to configure equalizers  2506  is to simply include one equalizer per channel (for the systems and methods described herein, a channel is the equivalent of a sub-channel as described above). A correlator, such as correlator  2306  ( FIG. 23 ), can then provide equalizers  2506  with an estimate of the number, amplitude, and phase of any multipaths present, up to some maximum number. This is also known as the Channel Impulse Response (CIR). The maximum number of multipaths is determined based on design criteria for a particular implementation. The more multipaths included in the CIR the more path diversity the receiver has and the more robust communication in the system will be. Path diversity is discussed a little more fully below.  
         [0157]     If there is one equalizer  2506  per channel, the CIR is preferably provided directly to equalizers  2506  from the correlator (not shown). If such a correlator configuration is used, then equalizers  2506  can be run at a slow rate, but the overall equalization process is relatively fast. For systems with a relatively small number of channels, such a configuration is therefore preferable. The problem, however, is that there is large variances in the number of channels used in different types of communication systems. For example, an outdoor system can have has many as 256 channels. This would require 256 equalizers  2506 , which would make the receiver design too complex and costly. Thus, for systems with a lot of channels, the configuration illustrated in  FIG. 25  is preferable. In receiver  2500 , multiple channels share each equalizer  2506 . For example, each equalizer can be shared by 4 channels, e.g., Ch1-Ch4, Ch5-Ch8, etc., as illustrated in  FIG. 25 . In which case, receiver  2500  preferably comprises a memory  2502  configured to store information arriving on each channel.  
         [0158]     Memory  2502  is preferably divided into sub-sections  2504 , which are each configured to store information for a particular subset of channels. Information for each channel in each subset is then alternately sent to the appropriate equalizer  2506 , which equalizes the information based on the CIR provided for that channel. In this case, each equalizer must run much faster than it would if there was simply one equalizer per channel. For example, equalizers  2506  would need to run 4 or more times as fast in order to effectively equalize 4 channels as opposed to 1. In addition, extra memory  2502  is required to buffer the channel information. But overall, the complexity of receiver  2500  is reduced, because there are fewer equalizers. This should also lower the overall cost to implement receiver  2500 .  
         [0159]     Preferably, memory  2502  and the number of channels that are sent to a particular equalizer is programmable. In this way, receiver  2500  can be reconfigured for the most optimum operation for a given system. Thus, if receiver  2500  were moved from an outdoor system to an indoor system with fewer channels, then receiver  2500  can preferably be reconfigured so that there are fewer, even as few as 1, channel per equalizer. The rate at which equalizers  2506  are run is also preferably programmable such that equalizers  2506  can be run at the optimum rate for the number of channels being equalized.  
         [0160]     In addition, if each equalizer  2506  is equalizing multiple channels, then the CIR for those multiple paths must alternately be provided to each equalizer  2506 . Preferably, therefore, a memory (not shown) is also included to buffer the CIR information for each channel. The appropriate CIR information is then sent to each equalizer from the CIR memory (not shown) when the corresponding channel information is being equalized. The CIR memory (not shown) is also preferably programmable to ensure optimum operation regardless of what type of system receiver  2500  is operating in.  
         [0161]     Returning to the issue of path diversity, the number of paths used by equalizers  2506  must account for the delay spread d s  in the system. For example, if the system is an outdoor system operating in the 5 Giga Hertz (GHz) range, the communication channel can comprise a bandwidth of 125 Mega Hertz (MHz), e.g., the channel can extend from 5.725 GHz to 5.85 GHz. If the channel is divided into 512 sub-channels with a roll-off factor r of 0.125, then each subchannel will have a bandwidth of approximately 215 kilohertz (KHz), which provides approximately a 4.6 microsecond symbol duration. Since the worst case delay spread d s  is 20 microseconds, the number of paths used by equalizers  2504  can be set to a maximum of 5. Thus, there would be a first path P 1  at zero microseconds, a second path P 2  at 4.6 microseconds, a third path P 3  at 9.2 microseconds, a fourth path P 4  at 13.8 microseconds, and fifth path P 5  at 18.4 microseconds, which is close to the delay spread d s . In another embodiment, a sixth path can be included so as to completely cover the delay spread d s ; however, 20 microseconds is the worst case. In fact, a delay spread d s  of 3 microseconds is a more typical value. In most instances, therefore, the delay spread d s  will actually be shorter and an extra path is not needed. Alternatively, fewer sub-channels can be used, thus providing a larger symbol duration, instead of using an extra path. But again, this would typically not be needed.  
         [0162]     As explained above, equalizers  2506  are preferably configurable so that they can be reconfigured for various communication systems. Thus, for example, the number of paths used must be sufficient regardless of the type of communication system. But this is also dependent on the number of sub-channels used. If, for example, receiver  2500  went from operating in the above described outdoor system to an indoor system, where the delay spread d s  is on the order of 1 microsecond, then receiver  2500  can preferably be reconfigured for 32 sub-channels and 5 paths. Assuming the same overall bandwidth of 125 MHz, the bandwidth of each sub-channel is approximately 4 MHz and the symbol duration is approximately 250 nanoseconds.  
         [0163]     Therefore, there will be a first path P 1  at zero microseconds and subsequent paths P 2  to P 5  at 250 ns, 500 ns, 750 ns, and 1 microsecond, respectively. Thus, the delay spread d s  should be covered for the indoor environment. Again, the 1 microsecond delay spread d s  is worst case so the 1 microsecond delay spread d s  provided in the above example will often be more than is actually required. This is preferable, however, for indoor systems, because it can allow operation to extend outside of the inside environment, e.g., just outside the building in which the inside environment operates. For campus style environments, where a user is likely to be traveling between buildings, this can be advantageous.  
         [0000]     7. Sample Embodiment of a Wireless Communication device  
         [0164]      FIG. 26  illustrates an example embodiment of a wireless communication device in accordance with the systems and methods described herein. Device  2600  is, for example, a portable communication device configured for operation in a plurality of indoor and outdoor communication systems. Thus, device  2600  comprises an antenna  2602  for transmitting and receiving wireless communication signals over a wireless communication channel  2618 . Duplexor  2604 , or switch, can be included so that transmitter  2606  and receiver  2608  can both use antenna  2602 , while being isolated from each other. Duplexors, or switches used for this purpose, are well known and will not be explained herein.  
         [0165]     Transmitter  2606  is a configurable transmitter configured to implement the channel access protocol described above. Thus, transmitter  2606  is capable of transmitting and encoding a wideband communication signal comprising a plurality of sub-channels. Moreover, transmitter  2606  is configured such that the various sub-components that comprise transmitter  2606  can be reconfigured, or programmed, as described in section  5 . Similarly, receiver  2608  is configured to implement the channel access protocol described above and is, therefore, also configured such that the various sub-components comprising receiver  2608  can be reconfigured, or reprogrammed, as described in section  6 .  
         [0166]     Transmitter  2606  and receiver  2608  are interfaced with processor  2610 , which can comprise various processing, controller, and/or Digital Signal Processing (DSP) circuits. Processor  2610  controls the operation of device  2600  including encoding signals to be transmitted by transmitter  2606  and decoding signals received by receiver  2608 . Device  2610  can also include memory  2612 , which can be configured to store operating instructions, e.g., firmware/software, used by processor  2610  to control the operation of device  2600 .  
         [0167]     Processor  2610  is also preferably configured to reprogram transmitter  2606  and receiver  2608  via control interfaces  2614  and  2616 , respectively, as required by the wireless communication system in which device  2600  is operating. Thus, for example, device  2600  can be configured to periodically ascertain the availability is a preferred communication system. If the system is detected, then processor  2610  can be configured to load the corresponding operating instruction from memory  2612  and reconfigure transmitter  2606  and receiver  2608  for operation in the preferred system.  
         [0168]     For example, it may preferable for device  2600  to switch to an indoor wireless LAN if it is available. So device  2600  may be operating in a wireless WAN where no wireless LAN is available, while periodically searching for the availability of an appropriate wireless LAN. Once the wireless LAN is detected, processor  2610  will load the operating instructions, e.g., the appropriate protocol stack, for the wireless LAN environment and will reprogram transmitter  2606  and receiver  2608  accordingly. In this manner, device  2600  can move from one type of communication system to another, while maintaining superior performance.  
         [0169]     It should be noted that a base station configured in accordance with the systems and methods herein will operate in a similar manner as device  2600 ; however, because the base station does not move from one type of system to another, there is generally no need to configure processor  2610  to reconfigure transmitter  2606  and receiver  2608  for operation in accordance with the operating instruction for a different type of system. But processor  2610  can still be configured to reconfigure, or reprogram the sub-components of transmitter  2606  and/or receiver  2608  as required by the operating conditions within the system as reported by communication devices in communication with the base station. Moreover, such a base station can be configured in accordance with the systems and methods described herein to implement more than one mode of operation. In which case, controller  2610  can be configured to reprogram transmitter  2606  and receiver  2608  to implement the appropriate mode of operation.  
         [0170]     While embodiments and implementations of the invention have been shown and described, it should be apparent that many more embodiments and implementations are within the scope of the invention. Accordingly, the invention is not to be restricted, except in light of the claims and their equivalents.

Technology Category: h