Patent Document

CROSS REFERENCE TO RELATED APPLICATIONS 
     This application is a divisional of U.S. patent application Ser. No. 09/816,590, filed Mar. 23, 2001, now U.S. Pat. No. 6,636,107, in the name of Brian R. Pelly and entitled ACTIVE FILTER FOR REDUCTION OF COMMON MODE CURRENT” which application is related to and claims priority to Provisional Application Serial No. 60/192,976, filed Mar. 28, 2000 and No. 60/211,999 filed Jun. 16, 2000. 
    
    
     FIELD OF THE INVENTION 
     This invention relates to filters for electrical circuits and more specifically relates to an active filter for reducing or redirecting the common mode current in switch mode power supplies and particularly for reducing the common mode current and EMI in a PWM motor drive circuit. 
     BACKGROUND OF THE INVENTION 
     High-speed switching devices such as bipolar transistors, MOSFETs and IGBT&#39;s enable increased carrier frequency for voltage-source PWM inverters, thus leading to much better operating characteristics. High-speed switching, however, causes the following serious problems, originating from a high rate-of-change in voltage and/or current: 
     a) ground current escaping to earth through stray capacitors inside motors and through long cables; 
     b) conducted and radiated EMI; 
     c) motor bearing current and shaft voltage; and 
     d) shortening of insulation life of motors and transformers. 
     The voltage and/or current change caused by high-speed switching produces high-frequency oscillatory common-mode and normal-mode currents when the switching device(s) change state because parasitic stray capacitance inevitably exists inside a load, for example, an ac motor, as well as inside the switching converter. Thus, each time an inverter switching event occurs, the potential of the corresponding inverter output terminal moves rapidly with respect to ground, and a pulse of common mode current flows in the d-c link to the inverter, via the capacitance of the heatsink motor cable and motor windings to ground. The amplitude of this pulse of current for a class B (residential) motor drive is typically several hundred millamps to several amps; and the pulse width is typically 250 to 500 ns. For a class A drive (Industrial), and depending on the size of the motor and length of the motor cable, the pulse current amplitude is typically several amperes with a pulse width of 250 ns to 500 ns, to 20 amperes or more with a pulse width of 1 to 2 micro seconds. 
     The common mode oscillatory currents may have a frequency spectrum range from the switching frequency of the converter to several tens of MHZ, which creates a magnetic field and will produce radiated electro-magnetic interference (EMI) throughout, thus adversely affecting electronic devices such as radio receivers, medical equipment, etc. 
     A number of Governmental restrictions apply to the degree of permissible line current EMI and permissible ground current in certain motor applications. Thus, in Class B residential (appliances), applications, ground current must be kept below from 1 to 20 mA over a frequency range from 0 to 30 kHz respectively (over a logarithmic curve); and conducted line current EMI must be kept below designated values (less than about 60 dBμV) over a frequency range of 150 kHz to 300 MHZ. For motor drive applications designated as class A Industrial applications, limitations on ground current are less stringent, but line current EMI is still limited over the 150 kHz to 30 MHZ range. 
     Generally, common-mode chokes and EMI filters, based on passive elements, may not completely solve these problems. Passive filters, consisting of a common mode inductor and “Y” capacitors in the input ac line have been used to filter the common mode current in such motor drive circuits. Passive common mode filters may place limits on the PWM frequency which can be used, are physically large (frequently a major fraction of the volume of the motor drive structure) and are expensive. Further, they are functionally imperfect in that they exhibit undesired resonance which runs counter to the desired filtering action. Further, in general purpose industrial drives, the drive circuit and motor are often connected by cables which are up to 100 meters or more long. The longer the cable, the greater the conducted common mode EMI in the motor cable, and the larger the required size of a conventional passive common mode input filter. 
     A common-mode transformer with an additional winding shorted by a resistor is known which can damp the oscillatory ground current. Unfortunately, a small amount of aperiodic ground current will still remain in this circuit. 
     Active filters for control of the common mode current in a pulse width modulated (PWM) controlled motor drive circuit are well known. Such devices are typically described in the paper an Active Circuit for Cancellation of Common-Mode Voltage Generated by a PWM Inverter, by Satoshi Ogasawara et al., IEES Transactions on Power Electronics, Vol. 13, No. 5, (September 1998 and in U.S. Pat. No. 5,831,842 in the names of Ogasawara et al. 
     FIG. 1 shows a typical prior art active filter circuit or EMI and noise canceller for an a-c motor. Thus, in FIG. 1, an a-c source comprising an input terminal L and a neutral terminal are connected to the a-c input terminals of a full wave bridge connected rectifier  40 . While a single phase supply is shown, the principles in this and in all Figures to be described can be carried out with a three-phase or multi-phase input. The positive and negative busses of rectifier  40  contain points A and D respectively and are connected to a three-phase bridge connected PWM controlled inverter  41 , at inverter terminals B and F. The output a-c terminals of the inverter are connected to a-c motor  42 . A filter capacitor  40   a  is also connected across terminals B and F. Motor  42  has a grounded housing connected to ground wire  43  with ground terminal  43   a.    
     The active filter consists of a pair of transistors Q 1  and Q 2 , connected across the d-c output lines of rectifier  40  with their emitters connected at node E. These define amplifiers which are controlled by output winding  44  of a differential transformer having input windings  45  and  46  connected in the positive and negative output busses of rectifier  40 . The winding polarities are designated by the conventional dot symbols. Winding  44  is connected between the control terminals of transistors Q 1  and Q 2  and the common emitter node E. A d-c isolating capacitor  47  is connected to ground line  43  at node C. 
     The active filter including capacitor  47  defines a path for diverting the majority of the common mode current which can otherwise flow in the path L or N, A, B, M (motor  42 ),  43 ,  43   a  and back to L or N; (or in the reverse path when polarity reverses) or in path L or N, D, F, M,  43 ,  43   a  (or in the reverse path when polarity reverses). Thus, most common mode current can be diverted, for currents from positive terminal A, in the path B, M, C, E, Q 2 , F, B, for “positive current”, and in the pattern B, M, C, E, Q 1 , B for “negative” current by the proper control of transistor Q 1  and Q 2 . The path for common mode current flowing into negative terminal D follows the path F, M, C, E, Q 2 , F for “positive” current and F, M, C, E, Q 1 , B for “negative” current. The degree of diversion depends on the current gain of winding  44  and the current gain of Q 2 , for “positive current”, and the current gain of winding  44  and current gain of Q 1 , for “negative” current. In order to obtain a sufficient degree of diversion of the common mode current, the overall current gain of winding  44  and transistors Q 1  and Q 2  must be high. 
     The sensing transformer  44 ,  45 ,  46  of FIG. 1 has been large and expensive in order to provide sufficiently high current gain. It would be very desirable to reduce the size and cost of this transformer without jeopardizing the operation of the circuit. A further problem is that because of the high gain required, this closed-loop circuit has a tendency to produce unwanted oscillation. 
     Further, it has been found that the transistors Q 1  and Q 2  may not be able to operate in their linear regions over a large enough range within the “headroom” defined by the circuit, thus defeating the active filtering action. The headroom, or the voltage between the collector and emitter of transistors Q 1  and Q 2  is best understood by considering the approximate equivalent circuit of FIG. 1, as shown in FIG. 2, in which the ground potential at C is the same as that of the neutral line in FIG.  1 . Transistors Q 1  and Q 2  are shown as resistors R 1  and R 2  respectively with respective parallel connected diodes. The d-c bridge  40  is shown as two d-c sources  50  and  51 , each producing an output voltage of V DC /2 where V DC  is the full output voltage between the positive and negative busses at terminals A and D, and an a-c source  52  having a peak a-c voltage of V DC /2. 
     It can be seen from FIG. 2 that headroom can disappear at different portions of the cycle of source  52 . Thus, consider a first situation in which the leakage impedances of transistors Q 1  and Q 2  are the same. In this case, the values of resistors R 1  and R 2  in FIG. 2 are about equal. Now, as the ground potential at terminal C swings between (+)V DC /2 and (−)V DC /2 with respect to the d-c midpoint at node  53  in FIG. 2, the potential at the emitters of transistors Q 1  and Q 2  also swings between (+)V DC /2 and (−) V DC 12, if it is assumed that the impedance of capacitor  47  is much smaller than R 1  and R 2 . Therefore, during the periods when the potential at node E is close or equal to the potential of the d-c bus (at points B or F), insufficient voltage headroom exists for the relevant transistors Q 1  or Q 2  to operate as linear amplifiers, and the active filtering action is lost. 
     Consider next a condition in which the ground potential at C is the same as that of the neutral input line N in FIG. 1, and the leakage current of transistor Q 2  is now much higher than that of transistor Q 1 , which, in FIG. 2 would be represented by the condition that R 2  is much less than R 1  This then biases the potential at E toward the negative d-c bus (at F). Therefore, the potential at E resides at the negative bus potential for a significant portion of each input cycle. During this period, transistor Q 2  cannot operate as a linear amplifier and the active filtering action is lost. 
     Consider next the condition where the ground is at N and the resistance of transistor Q 1  is much less than that of transistor Q 2 ; that is, R 1  is much less than R 2 . This would bias point E toward the positive d-c bus (point B) so that, for a significant portion of each input cycle, transistor Q 1  cannot operate as a linear amplifier. 
     Lastly, consider a condition in which a small a-c potential (dotted line a-c source  52   a  in FIG. 2) exists between the ground wire  43  and neutral N of FIG. 1, as would occur if the grounded neutral of the supply transformer is electrically remote from the ground connection of the motor drive itself. In that case, the ground potential will swing between +(dV+V DC /2) and −(dV+V DC /2) where dV is the peak of the voltage wave shape of source  52   a  in FIG.  2 . If the leakage characteristics of transistors Q 1  and Q 2  are about equal, the potential at E will attempt to swing by dV above the positive bus and by (−) dV below the negative d-c bus. During these periods, the voltage at E is clamped to the bus voltage by the low impedance reverse characteristics of transistors Q 1  and Q 2 . Thus, no voltage headroom exists for the transistors to operate during these periods. 
     It would be very desirable to provide a circuit which provides sufficient headroom for transistors Q 1  and Q 2  under the above described conditions and for clamping the headroom voltages to a prescribed minimum level; or for regulating the average voltage at point E to the d-c midpoint potential. 
     The active filter of the prior art, as shown in FIG. 1, is always connected across the full d-c bus voltage. This requires a high enough voltage rating for the transistors Q 1  and Q 2  and causes a relatively high power dissipation in the active filter components. It would therefore be desirable to operate the active filter at a lower voltage, if possible, without degrading the performance of the active filter. 
     SUMMARY OF THE INVENTION 
     In accordance with a first feature of the invention, an operational amplifier is used as a buffer/amplifier between the current sensing transformer and the transistors Q 1  and Q 2 . This permits a substantial reduction in the size of the common mode transformer, without affecting the operation of the circuit. 
     In accordance with a further feature of the invention, and to ensure sufficient headroom for the transistors Q 1  and Q 2  of FIG. 1, a pair of series connected balancing resistors are connected in parallel with respective ones of transistors Q 1  and Q 2 . The novel resistors will have a value which ensures that sufficient voltage headroom is always maintained for transistor Q 1  and Q 2 . This permits a current flow which is significantly higher than the maximum possible leakage current of transistors Q 1  and Q 2 . 
     Alternatively, a further novel circuit which has substantially less power dissipation while still maintaining sufficient headroom for transistors Q 1  and Q 2  is provided, using active clamps for the voltage headroom of the transistors Q 1  and Q 2 . In this circuit, the instantaneous voltage across each of transistors is sensed and compared to respective references. If the headroom voltage falls below the reference, a feedback error is fed back to the amplifier which drives the transistors Q 1  and Q 2  to maintain the required headroom. This novel circuit substantially reduces the power dissipation needed for voltage headroom maintenance and reduces the magnitude of ground current at line frequency through the d-c isolating capacitor  47 . 
     In accordance with a still further feature of the invention, the headroom voltage control is carried out employing a reference voltage which is equal to one-half of the d-c bus voltage. The voltage at the emitter node E is then compared to this reference and the transistors Q 1  and Q 2  are controlled by an active regulation scheme which regulates the average voltage at E to the d-c midpoint voltage. 
     In the circuits described above, the amplifiers used need a source of operating or control or biasing voltage. A novel floating power supply is provided which derives its power from the d-c bus voltage and permits all output control voltages to move dynamically within the positive and negative bus voltage. This is accomplished by providing respective current source circuits connected to the positive and negative busses, respectively, and connected to one another through zener reference diodes. The nodes between the current sources and diodes and the node between the diodes form outputs for two control voltages and a common voltage reference which all swing dynamically with the bus voltage. 
     In a still further embodiment of the invention, the operating voltage across the active filter is sourced from a separate “filter bus” voltage which is lower than the full bus voltage. The active filter acts otherwise identically to a circuit driven from the full bus voltage. This novel circuit reduces the power dissipation in the active filter components and lowers the voltage rating of the transistors Q 1  and Q 2 . 
     In accordance with a further important feature of the invention, selected active filter components are integrated into a single silicon chip, defining an active filter IC chip containing the principle active filter components and having suitable pin-outs for receiving the various input and bus connections. 
     A novel architecture is also provided, enabling the use of a very small toroidal current transformer as the common mode sensor. Unlike the architecture in FIG. 1, which requires a high gain, this novel architecture requires a gain of only unity, and thus avoids the above mentioned problem of unwanted oscillation. The architecture may also be implemented by MOSFET transistors instead of bipolars for improved linearization of the amplifier transfer characteristics wider bandwidth and improved ruggedness. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a circuit diagram of a prior art common mode active filter for a motor drive circuit. 
     FIG. 2 is an equivalent circuit of the circuit of FIG. 1 to illustrate the issue of headroom voltage. 
     FIG. 3 is a circuit diagram of one improvement of the invention which employs a buffer amplifier between the current sensing transformer and the transistors. 
     FIG. 4 shows modification of the circuit of FIG. 3 in which the current sense primary windings are placed in the a-c circuit. 
     FIG. 5 is a circuit diagram of a further modification of the circuit of FIG. 3 in which the primary winding of the current transformer is in the ground wire. 
     FIG. 6 (on the same sheet as FIG. 13) shows an improvement of the circuit of FIG. 3 which employs balancing resistors across the transistors to ensure sufficient headroom for their operation. 
     FIG. 7 shows a further embodiment of a circuit for insuring headroom using an active clamp circuit. 
     FIG. 8 is a still further embodiment of a headroom control circuit, in which the midpoint d-c bus potential is actively regulated. 
     FIG. 9 is an equivalent circuit of the circuit of FIG.  8 . 
     FIGS. 10,  11  and  12  show the ground voltage, the voltage across transistors Q 1  and Q 2 , the voltage across the filter capacitor and the ground current i GND  at line frequency for three different headroom conditions. 
     FIG. 13 (on the same sheet as FIG. 6) shows a novel power supply for the amplifier circuits of the prior figures which derives its power from the d-c bus. 
     FIG. 14 is a circuit diagram of an improvement of the prior described circuits in which the filter bus voltage is less than the main d-c bus voltage. 
     FIG. 15 is an alternative to the circuit of FIG. 14 in which the reduced filter bus voltage is derived from a dropping resistor and zener diode circuit. 
     FIG. 16 is another circuit for deriving a reduced filter voltage using series bus capacitors. 
     FIG. 17 is a circuit for deriving a reduced filter voltage of one-half of the full bus voltage and employing a voltage doubler. 
     FIG. 18 is a circuit similar to that of FIG. 17 in which the filter voltage is less than one-half of the full d-c bus voltage. 
     FIG. 19 is a diagram of an integrated circuit chip into which selected ones of the filter components of the preceding figures are integrated into a single chip of silicon. 
     FIG. 20 shows the architecture of a circuit having an active filter IC chip in a switching power supply. 
     FIG. 21 shows a feedback based architecture using an active filter IC chip in a motor drive circuit with the current sensor primary in the ground wire. 
     FIG. 22 shows a modification of the architecture of FIG. 21 employing a feed forward system which permits improved operation. 
     FIG. 23 shows an architecture like that of FIG. 21, but using a differential current transformer. 
     FIG. 24 shows the architecture of FIG. 22 with a differential transformer. 
     FIG. 25 shows the novel active filter of the present invention using power MOSFETs instead of bipolar transistors for devices Q 1  and Q 2 . 
     FIG. 26 is an improvement of FIG. 25, adding further control elements. 
     FIG. 27 shows a modification of the circuit of FIG. 26 for improved control of the quiescent bias current. 
     FIG. 28 shows a full circuit diagram of a preferred embodiment of the common mode filter of the invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     FIG. 3 shows an improvement of the circuit of FIG. 1 wherein like numerals designate similar components as is true throughout this specification. FIG. 3 further shows, in schematic fashion, a heat sink  49  which receives the IGBTs (or MOSFETs) of the PWM inverter  41 . Heat sink  49 , like the housing of motor  42 , is connected to ground line  43 . In FIG. 3, however, a novel improvement is added which permits a substantial reduction in the size of the transformer  44 ,  45 ,  46  without affecting the operation of the active filter function. Thus, in FIG. 3, an operational amplifier  70  is added to the circuit to act as a buffer/amplifier between the secondary of the common mode current sensing transformer  44 ,  45 ,  46  and the transistors Q 1  and Q 2 . This permits the size and cost of transformers  44 ,  45 ,  46  to be substantially reduced. 
     While the primary windings  45  and  46  of the common mode transformer are shown in FIG. 3 at the output of rectifier  40 , they may be placed in the a-c input lines as shown in FIG. 4; or, if desired, a single primary winding  71  can be connected in series with the ground terminal, as shown in FIG.  5 . In all cases, the secondary winding  44  is simply coupled to one or more primary windings which carry or otherwise produce a signal related to the common mode current. 
     Regardless of the connection used, the novel circuits of FIGS. 3,  4  and  5  will divert the majority of the common mode current flow into the isolator capacitor  47  and transistors Q 1  and Q 2  and away from the external ground wire connected to  43   a.    
     The circuits of FIGS. 1,  3 ,  4  and  5  will operate as desired provided transistors Q 1  and Q 2  always have sufficient voltage “headroom” to allow them to operate as linear amplifiers. However, as pointed out earlier with reference to FIG. 2, there are situations in which this headroom disappears during certain portions of the input voltage cycle. In accordance with another feature of the invention and as shown in FIG. 6 a novel circuit employing balancing resistors will ensure adequate headroom under these conditions. Thus, in FIG. 6, two resistors  85  and  86  are connected across transistors Q 1  and Q 2  respectively. Note that the full circuit is not shown in FIG. 6, it is being noted that such resistors are intended to be connected across the transistors Q 1  and Q 2  in FIGS. 1,  3 ,  4  and  5 . 
     Resistors  85  and  86  must satisfy the following conditions: 
     1. Their values must be of the same order of magnitude as the reactance of capacitor  47  to ensure that the peak value of the a-c component of voltage at line frequency and developed across transistors Q 1  and Q 2  is substantially less than one-half the peak d-c bus voltage. For example, capacitor  47  may have a typical value of 0.01 μF and, at line frequency, the resistors  85  and  86  will be 265 kohms each. 
     2. Resistors  85  and  86  must be sized to carry significantly higher current than the maximum possible leakage current of transistors Q 1  and Q 2 , in order to “swamp” differences in their leakage currents and ensure an approximately equal voltage balance across the transistors. In a specific example, in a circuit having a maximum d-c bus voltage of 400 volts with transistors Q 1  and Q 2  having a maximum leakage current of 0.5 mA each and capacitor  47  having a capacitance of 0.01 μF, the current through resistors  85  and  86  at half the bus voltage, i.e., 200 volts, should be about 2 mA. Thus, resistors  85  and  86  should be about 100 kohms. This is the value to be chosen since it is less than the 265 kohm value required by criteria 1 above. 
     The power dissipation in the 100 kohm resistors at the bus voltage of 400 volts is about 400 mw. A line frequency component of ground current of about 0.5 mA will be drawn through resistors  85  and  86  and the capacitor  47 . 
     As will be later discussed, it would be desirable to integrate the control circuitry for the active filter into an integrated circuit. Thus, the power dissipation should be reduced as much as possible. FIG. 7 shows a circuit for the control of headroom which does not require resistors and which reduces power dissipation to about 20 mw. The circuit of FIG. 7 also reduces the line frequency ground current drawn through capacitor  47  from 0.5 mA to 0.2 mA. 
     The principle of operation of the circuit of FIG. 7 is to actively clamp the voltage headroom for transistors Q 1  and Q 2 . This is done by sensing the instantaneous voltage across each of the transistors Q 1  and Q 2  and comparing it to a reference. If the headroom falls below the reference value, an error signal is produced and is fed back to amplifier  70  to drive the base of transistors Q 1 or Q   2  to maintain the desired headroom. Thus, transistors Q 1  and Q 2  will carry the common mode current and will also carry a small added component of current so as to maintain their necessary headroom. The power dissipation in each of transistors Q 1  and Q 2  is only about 20 mw at a bus voltage of 400 V, compared to 400 mV for each of resistors  85  and  86  in FIG.  6 . 
     Referring now to FIG. 7, summing circuits  90 ,  91  and  92  and amplifiers  93  and  94  are added as shown. FIG. 7 also shows power supply d-c sources  95  and  96  for providing to V DD  and V SS  inputs respectively to operational amplifier  70 . Each of summing circuits  90  and  91  have one input connected to (+)DC Bus and (−)DC Bus respectively, and another input (terminals  97  and  98  respectively) connected to reference voltages (−)ve and (+)ve respectively which are the headroom reference voltages for transistors Q 1  and Q 2  respectively. The summed voltages e in  and e 1   in  respectively of devices of  90  and  91  are applied to operational amplifiers  93  and  94  respectively which have outputs e out  and e 1   out  which are applied to inputs of summing circuit  92 . The linear shapes of the characteristic outputs e out  and e 1   out  are shown on the drawing in circled insets. The output of circuit  92  (an error signal output) is connected through high frequency filter  99  to the input of amplifier  70 . 
     Thus, amplifier  70  will amplify the error signal; and amplifiers  93  and  94  respectively prevent the headroom voltages for transistors Q 1  and Q 2  from falling below their respective reference voltages. 
     In more detail, +DC BUS to COM REF (−v e ) is a negative reference voltage, which sets the required value of the +DC BUS to COM voltage The value of this reference (−)ve is set to yield the desired headroom clamping level for transistor Q 1 , based on the worst case assumption that the emitters of transistors Q 1  and Q 2  could instantaneously be positive with respect to COM by VDD. Thus +DC BUS to COM REF is set to represent the required minimum headroom for Q 1 , plus VDD. 
     The difference between +DC BUS to COM REF(−ve) and the actual +DC BUS to COM voltage (+ve) is fed to the input of amplifier  93 . The output of amplifier  93  is zero for positive input, and positive for negative input voltage. When the +DC BUS to COM voltage is greater than the absolute value of +DC BUS to COM REF, e in  is positive, and the output of amplifier  93  is zero. When +DC BUS to COM attempts to become less than the absolute value of +DC BUS to COM REF which is set to represent the required minimum headroom for Q 1 , plus VDD, e in  becomes negative, and the output e out  of amplifier  93  becomes positive. This output is inverted at the summing junction  92 , then passed through the HF filter  99  to the amplifier  70 , biasing the output of this amplifier negatively. The result is that transistor Q 2  is biased on just sufficiently that the voltage headroom across transistor Q 1  is regulated to the set value. 
     The (−)DC BUS to COM REF is a set positive reference voltage at terminal  98 . The required value of this reference represents the required minimum headroom for transistor Q 2 , plus the absolute value of VSS. 
     The difference between (−)DC BUS to COM REF (+ve) and the −DC BUS to COM voltage (−ve) is fed to the input of amplifier  94 . The output of amplifier  94  is zero for negative input, and negative for positive input voltage. When the absolute value of the (−)DC BUS to COM voltage is greater than (−)DC BUS to COM REF, e 1   in  is negative, and the output of A3 is zero. When the absolute value of the (−)DC BUS to COM voltage attempts to become less than (−)DC BUS to COM REF, e 1   in  becomes positive, and the output e 1   out  of amplifier  94  becomes negative. This output is inverted at the summing junction  92 , then passed through the HF filter  99  to the input of amplifier  70 , biasing the output of this amplifier positively. The result is that transistor Q 1  is biased on just sufficiently that the voltage headroom across transistor Q 2  is regulated to the set value. 
     The HF filter  99  removes high frequency components from the error signal, caused by the common mode current of the inverter/motor that flows through CFILT  47 . Thus the closed loop regulator corrects against errors of potential at E that would otherwise occur at line frequency, but essentially does not attempt to make corrections for faster instantaneous deviations due to the switching events of the inverter. 
     FIG. 8 shows a further alternative to the circuit of FIG. 6 in which the voltage at node E of the common emitters of transistors Q 1  and Q 2  is sensed and the average value of this voltage is regulated to the d-c midpoint voltage. Thus, in FIG. 8, a reference voltage circuit defining the d-c mid-voltage consists of resistors  105  and  106  which are low power resistors which are of equal value and produce voltage at node  107  which is the midpoint of the d-c voltage between the (+)DC Bus and (−)DC Bus voltages. This d-c midpoint reference voltage, and the potential at node E are applied to, and compared by, a summing comparator  108 . The difference output is then applied to amplifier  109  and its output error signal is connected via HF filter  99  as an input to amplifier  70 . 
     In operation, if the voltage at E attempts to rise above the d-c midpoint reference potential at node  107 , amplifier  109  delivers a negative bias signal to the input of amplifier  70 . The output of amplifier  70  also assumes negative bias, turning on transistor Q 2  to the extent necessary to correct the voltage at emitter node E and back towards the d-c midpoint reference potential. 
     Conversely, if the voltage at E attempts to fall below the d-c midpoint potential, the output of amplifier  70  assumes a positive bias, turning on transistor Q 1  to the extent necessary to correct the voltage at E back towards the d-c midpoint potential. The HF filter  99  again removes high frequency components from the error signal. 
     Thus the regulator loop corrects against variations of potential at E that would otherwise occur at line frequency, but essentially does not attempt to make corrections for faster instantaneous deviations caused by the switching events of the inverter. 
     With the scheme of FIG. 8, the average voltage headroom for transistors Q 1  and Q 2  is maintained at approximately half the bus voltage. While this is more than sufficient headroom, a disadvantage, relative to the “headroom clamping” approach of FIG. 7, is that the line frequency component of voltage across capacitor  47  is increased and the corresponding line frequency component of ground current that flows through capacitor  47  is relatively high. 
     FIGS. 9 to  12  illustrate the design trade-offs between headroom voltage for transistors Q 1  and Q 2  and ground current for the circuits of FIGS. 1,  7  and  8  (at 60 Hz). 
     FIG. 9 is an equivalent circuit of a portion of the circuit of FIG.  8 . The a-c voltage source shown in dotted lines between nodes  107  and  43   a  is the a-c ground voltage with respect to the DC midpoint reference voltage. This voltage, as shown in FIG. 10 is one-half of the peak a-c voltage between a-c lines L and N (assuming a single phase input with N at ground potential) and this voltage drives ground current through capacitor  47 . When the circuit is designed to provide generous headroom for transistors Q 1  and Q 2  as shown in FIG. 10, as by regulating the voltage at E to the d-c midpoint potential, the ground current i GND  at 60 Hz increases as shown in the bottom graph of FIG.  10 . 
     If, however, the voltage at E is allowed to swing freely with respect to the bus potentials, as shown in FIG. 11, the line frequency voltage across capacitor  47  is zero and the corresponding line frequency ground current is also zero. The headroom voltage, however, (the center graph of FIG.  11 ), is insufficient for the desired filtering of the common mode motor current (as in FIG.  1 ). 
     Finally, if, as in FIG. 7, a headroom clamping circuit is employed, allowing the potential at node E to swing freely until the headroom falls below a set value, the curves appear as in FIG. 12 where headroom for the transistors is clamped to the set value while the 60 Hz ground current is reduced to a small value during the clamp intervals only. 
     An isolated floating power supply would normally be provided for the needed Vdd and Vss supply voltages for the amplifier circuits of FIGS. 3,  4 ,  5 ,  7  and  8 . An obvious way of deriving the required power supply voltages would be via a separate isolated winding on the power supply transformer that serves the control electronics and gate drive circuits. 
     FIG. 13, however, shows an alternative method for deriving the required power supply voltages. Thus, in FIG. 13, current sources  120  and  121  feed a fixed current from the d-c bus, to create the Vdd and Vss power supply voltages across the regulator zener diodes  122  and  123  respectively. The currents of sources  120  and  121  must be equal, and independent of the voltage at COM. The voltage then moves dynamically within the positive and negative bus voltages, and the headroom is controlled as described above. 
     Note that the current sources  120  and  121  cannot be replaced by simple dropping resistors, because the currents of sources  120  and  121  could not then be instantaneously equal, because the two resistors would have different voltages impressed across them as the potential at COM dynamically changes. The difference between such two resistor currents would be forced through capacitor  47 , interfering with the active filter function. 
     The use of the described bus-derived power supply circuit may be limited by the power dissipation associated with the circuit. This depends upon the required power supply current, and the bus voltage. For example, if the maximum required power supply current is 3 mA, and the bus voltage is 200V, the maximum total dissipation due to the power supply would be 600 mW, which is acceptable. 
     FIG. 14 is a circuit diagram of a novel active filter which has a reduced operating voltage. Thus, it has been recognized that it is not necessary for the active filter to be connected directly across the full d-c bus voltage. The active filter, complete with the previously described current source power supply, can be sourced from a separate “filter bus” voltage, that can be lower than the full d-c bus voltage. The headroom control methods already described will still operate in the same way to maintain the desired headroom for transistors Q 1  and Q 2 . It is necessary only that a low ac impedance path for the common mode current is provided from the collectors of both transistors Q 1  and Q 2  to one of the main d-c busses. A low ac impedance path to the other main bus is then already in place, via the main d-c bus capacitor  40   a.    
     FIG. 14 shows the arrangement with a separate positive filter bus  130  which provides the positive voltage for the active circuit; the negative bus of the active filter being common with the main negative d-c bus. (It is also possible to provide a separate negative filter bus voltage, with the positive bus of the active filter being common with the main positive d-c bus). The operational amplifier circuits are schematically shown as block  132 . 
     The capacitor  131  provides the required low ac impedance between the positive filter bus  130  and the main negative bus. Flow paths for the common mode current under various operating conditions are the same as those previously described. 
     The advantages of using a filter bus voltage that is lower than the main d-c bus voltage are: 
     (a) a lower power dissipation in the active filter components, including the current source power supply. 
     (b) a lower voltage rating is required for the transistors Q 1  and Q 2 , and for the current source transistors of the power supply. A lower voltage rating for transistors Q 1  and Q 2  is advantageous because this permits transistors with better Safe Operating Area and better high frequency performance, giving faster response and improved performance of the active filter. 
     The use of a filter bus voltage that is lower than the main d-c bus voltage will reduce the allowed swing of line-frequency voltage across transistors Q 1  and Q 2 , because their emitters E must now swing within narrower limits. Thus, in order to maintain the required operating headroom for transistors Q 1  and Q 2 , the line-frequency voltage swing across filter capacitor  47  may be forced to increase, resulting in an increase in the line-frequency component of ground current drawn through capacitor  47 . While this would be the case with single phase input with grounded neutral; it would not be the case for three phase input with grounded neutral, or for 1-phase input with midpoint ground. Thus, with a three phase input with neutral ground, the swing of ground voltage with respect to the d-c midpoint is less than 25% of the main d-c bus voltage. The filter bus voltage would therefore have to be reduced to less than about 25% of the main bus voltage before the line-frequency component of current drawn through capacitor  47  would increase. 
     FIGS. 15 to  18  show alternate circuits for deriving a lower filter bus voltage from the main d-c bus voltage. In FIG. 15, the filter bus voltage is derived via a dropping resistor  140  and a voltage regulator diode  141  from the d-c bus. While the resistor  140  dissipates power, it permits power savings in the active filter itself, and may allow improved filter performance, by virtue of using lower voltage and wider bandwidth active components. 
     FIG. 16 employs two series connected d-c bus capacitors  145  and  146 , each supporting approximately half the bus voltage. Series connected bus capacitors may be used where the input voltage is 380V or higher. The principle in FIG. 16 is to use the voltage across the lower bus capacitor  146 , i.e., approximately half the d-c bus voltage, as the filter bus voltage. This requires additional active means shown as active regulator block  147  for keeping the voltages across the two bus capacitors balanced. 
     FIG. 17 shows an arrangement with a single phase supply L, N input to a voltage doubler circuit. The voltage doubler replaces rectifier  40  of the preceding circuits and consists of diodes  160 ,  161  and capacitors  162  and  163 . The main d-c bus voltage is approximately twice the peak line voltage. The voltage across each capacitor is half the full d-c bus voltage. The filter bus  130  is connected to the lower bus capacitor  161  in accordance with the feature of the invention, and is half the full d-c bus voltage. No additional means of balancing the voltages across the doubler capacitors is needed, since balancing of these voltages is a natural result of the operation of the voltage doubler circuit. 
     FIG. 18 is similar to FIG. 17, but adds the resistor  140  and zener diode  141  as in FIG. 15 to derive a lower filter bus voltage, which is less than one-half of the main d-c bus voltage. 
     There is next described in FIG. 19 an arrangement for integrating the prior described circuits into an integrated circuit active filter chip product. 
     The integrated circuit of FIG. 19 may have numerous filter components integrated into a single silicon chip, as shown by the dotted line periphery  170 . The integrated circuit  171  contains transistors Q 1  and Q 2 ; the voltage divider  105 ,  106  of FIG. 8; the floating power supply  120 - 123  of FIG. 13; the amplifiers  70  and  109  of FIG. 8 (and a second amplifier  109   a ). The IC chip  170  then has plural pin-outs including the labeled pins for (+)Filter Bus; (−)Filter Bus; capacitor  47 ; CT winding  44 ; and capacitor C VDD , and C VSS . Added pins may be provided for decoupling capacitors and the like. 
     FIG. 20 shows how the integrated circuit  170  has a general application as an active filter in a switching power supply. Thus, the switching power supply may have an input rectifier  40  connected to a-c input lines L and N, and an inverter  41 . The output of inverters  41  is connected to a transformer  180  which is then connected to an output rectifier  181 . All of parts  40 ,  41 , and  181  are mounted on heat sink  49 . The heat sink is connected to a ground line which contains the primary of a sensing current transformer housing a secondary winding  44 , as in FIG.  5 . 
     The active common mode filter integrated circuit  170  is then connected to the secondary sense winding  44  and controls the common mode current as previously described. 
     The use of the current sensor  71 ,  44  in FIG. 20 is employed rather than the differential CT  44 ,  45 ,  46  because such a circuit is more universally applicable and advantageous for power supplies and because there is no external motor which could have an extraneous ground current which would not be picked up by the current transformer. 
     In the case of a power supply, all the grounded parts are contained within the power supply itself (the output transformer  180  can have a ground screen), and, therefore, there may be no stray external ground current. 
     In the ground connection shown in FIGS. 5 and 20, the CT  71 ,  44  will be smaller because it has only one primary winding, sized for the ground current, (the ground wire actually must be same size as the “live” wires) versus two primaries, each sized for the full load current. Also, since the ground current is sensed directly, there is no possibility for differential errors. 
     One basic system architecture for the active common mode filter of the invention is shown in FIG. 21 showing a toroidal current sensing transformer  200 . A basic design objective is that the toroidal current sensing transformer  200  should be as small as possible. Preferably the primary should be just a single wire  201  that passes directly through the center of the toroidal core, as illustrated in FIG.  21 . 
     The architecture of FIG. 21 is a closed loop feedback based system. It is a feedback system in that the signal on the primary of the sensor  200  is an attenuated signal, so the amplifier consisting of winding  44 , buffer  70  and transistors Q 1  and Q 2  must be a high gain amplifier. Consequently, the system is subject to possible oscillation and requires a moderately large CT. That is, the architecture in FIG. 21 basically requires a high current gain, G, between the output current of the amplifier, i o , and the current input, i GND , to the primary  201  of the current transformer. The reasons for this are:              a   )               i   GND       i     COM                 MOT         =     1     1   +   G                                 where                 G     =       i   o       i   GND                                    
     Therefore, in order to minimize i GND , G must have a high value. 
     b) the propagation delay times of the bipolar transistors, Q 1  and Q 2 , result in a lag between the output current i o , and the input current, i GND , of the CT  200 . This lag causes spikes on the ground current waveform at the crossover points. This effect becomes more pronounced as the half-period of the oscillatory current, i COM MOT , decreases. A typical half-period of i COM MOT  is 200 to 250 nanoseconds, and the crossover spikes are significant. 
     The crossover spikes can be reduced by increasing the gain. The higher the gain, the more overdrive current is fed to the bases of transistors Q 1  and Q 2  during the crossover periods, which reduces the lag. The gain can be increased by increasing the number of primary turns of the current transformer. Unfortunately, this is not desirable because it is contrary to the basic design objective of minimizing the size of the current transformer and using a primary wire that passes just once through the center of the CT. 
     In principle, the gain could be increased by increasing the gain of the operational amplifier circuit  70 . In practice, this tends to cause closed-loop oscillation, which has been found to be difficult to suppress. 
     An improved architecture is shown in FIG.  22 . The architecture of FIG. 22 is a feed forward architecture in which the full forward current i com mot  flows through the primary winding of CT  200  and the full current is always sensed even as the ground current is being reduced. Consequently, the amplifier needs a gain equal to 1.0. This system, with unity gain, has good stability, and a small CT size. Thus, in contrast to the architecture in FIG. 21, this architecture requires unity current gain, G, between the current i com mot  at the input of the current transformer, and the output current, i o , of the amplifier. 
     The ground current, i GND , is:          i   GND     =         i     COM                 MOT       -     i   o            
                =       i     COM                 MOT       -     G   ·     i     COM                 MOT                                    
     For perfect cancellation of the ground current, the gain G must therefore be exactly 1.0. 
     The fundamental advantages of the architecture of FIG. 22 are: 
     a) Because the required current gain, G, is only unity, the current transformer  200  can be physically much smaller, with just a single primary wire passing through the center of the toroidal core. 
     b) The circuitry has reduced susceptibility to instability, because the overall current gain, from input of the current transformer  200  to the output of the amplifier  170 , is only 1.0. 
     c) A further advantage of the basic architecture of FIG. 22 over that of FIG. 21 arises where differential primary windings must be used on the current transformer  200 , rather than a single ground current sensing primary. 
     FIG. 23 shows the same basic feedback system architecture as in FIG. 21 for the case of a CT  210  with two differential secondary windings  211  and  212  carrying only a small fraction of the common mode current, but carrying the full normal mode current. Small unbalances between the differential windings  211  and  212  will result in imperfect cancellation of the normal mode current, and will yield a relatively large unwanted normal mode signal at the secondary, in addition to the “desired” common mode signal. The unwanted normal mode signal distorts the desired feedback signal, and upsets the operation of the active filter. 
     FIG. 24 shows the same basic proposed feedforward system architecture as in FIG. 22, and the differential primary windings  211  and  212  of the CT  210  carry the full common mode current, as well as the full normal mode current. Thus the ratio of unwanted normal mode signal at the secondary  44  of the transformer (caused by imperfect cancellation between the two primary windings), to the wanted common mode signal, is now much lower, (because the common mode signal is much higher). Any residual normal mode signal thus has a relatively much smaller distorting effect on the output of the amplifier. 
     A potential design problem with the architecture of FIG. 22 is that to obtain good cancellation of the ground current, the input/output current characteristic of the amplifier  170  must be linear, and it must have as close to unity gain and as small a phase lag as possible. MOSFETs are potentially better candidates than bipolar transistors for devices Q 1  and Q 2 , because of their minor propagation delays, as well as their better SOA. 
     Linearization of the transfer characteristic can be largely achieved through the use of MOSFET source followers, in conjunction with a standing d-c bias that offsets the gate threshold voltage. Additional “linearizing” feedback around the amplifier can also be added to correct against residual non-linearity and propagation delays. This added feedback does not require high gain, therefore it does not invite closed-loop instability. 
     A basic amplifier circuit implementation employing N channel MOSFET transistor Q 1  and P channel MOSFET transistor Q 2  is shown in FIG.  25 . In FIG. 25, components similar to those of the prior Figures have the same identifying numeral. A floating power supply  300  is provided to supply biases V DD  and V SS . The circuit components are easily integrated into a chip  170  as shown in dotted line outline. Amplifiers  301  and  302  drive N channel MOSFET Q 1  and P-channel MOSFET Q 2  respectively, and provide respective positive and negative d-c bias voltages that offset the gate threshold voltages of the MOSFETs. 
     Consider amplifier  301 , and assume that resistor RFB is open. The output voltage of amplifier  301 , shown as e o , is:                e                 o     =         i   S            R   S          (     1   +   B     )         +         R   4               V   SS              R   2                 (   1   )                                
     where i S  is the output of winding  44 , and Rs, R 1 , R 2 , R 4 , and V SS  are shown in FIG. 25, and        B   =     (     1   +       R   4       R   1       +       R   4       R   2         )                            
     R 2  is selected so that:                R   4               V   SS            R2     =   Vgth     ,   where                          
     Vgth is the gate threshold voltage of MOSFET Q 1  Thus the output voltage of amplifier  301  has a standing d-c bias that substantially cancels the threshold voltage of MOSFET Q 1 . 
     The output current, i o , of the source follower circuit comprising Q 1 , R SOURCE , R SENSE  is:                i   o     =           i   S            R   S          (     1   +   B     )             R   SOURCE     +     R   SENSE         ×     (       1   +     gfs        (       R   SOURCE     +     R   SENSE       )           gfs        (       R   SOURCE     +     R   SENSE       )         )               (   2   )                                
     If gfs.(R SOURCE +R SENSE )&gt;&gt;1, then equation (2) approximates to:                i   o     =         i   S            R   S          (     1   +   B     )             R   SOURCE     +     R   SENSE                 (   3   )                                
     Equation (3) shows that i o  is proportional to i S . This of course is based on the assumptions that (a) Vgth is exactly canceled by            R   4               V   SS            R2                          
     and (b) gfs(R SOURCE +R SENSE )&gt;&gt;1. 
     In practice, some degree of non-linearity between i o  and i S  will occur. This non-linearity is reduced by the feedback circuit comprising R SENSE  and R FB . 
     R SENSE  is chosen so that e FB  (=i o .R SENSE ) is nominally equal to e in  (=i S .R S ). i.e.e FB  nominally tracks e in . If e FB  exactly tracks e in , then the voltage across R FB  will be zero, no current will flow in R FB , and the feedback circuit has no modifying effect. 
     If (i o .R SENSE ) becomes greater than (i S .R S ), feedback current flows through R FB , towards the negative input terminal of amplifier  301 ; this has the effect of decreasing e o , thus reducing the error. 
     Conversely, if (i o .R SENSE ) becomes less than (i S .R S ), feedback current flows through R FB , away from the negative input terminal of amplifier  301 ; this has the effect of increasing e o , thus reducing the error. 
     Consider next the CT  200  and designate its numbers of primary and secondary turns as N p  and N S  respectively. 
     Returning to equation (3), and substituting          N   p       N   S                            
     i COM MOT  for i S :                i   o     =       (         N   P       N   S       -         R   S          (     1   +   B     )           R   SOURCE     +     R   SENSE           )          i     COM                 MOT                 (   4   )                                
     By appropriate choice of resistor values and N S  (secondary turns of the CT  200 ):              N   P       N   S                R   S          (     1   +   B     )           R   ROURCE     +     R   SENSE           =   1.0                          
     and equation (4) becomes: 
     
       
         i o =i COM MOT    
       
     
     This is the required design condition for i GND =0. 
     The design principles for the amplifier circuit  302 , which drives MOSFET Q 2 , are similar. 
     It is desirable that MOSFETs Q 1  and Q 2  are automatically biased, so that a small standing d-c bias current flows through these transistors, from the positive d-c bus to the negative d-c bus. This will ensure that these transistors are biased just to the point of conduction, thereby minimizing crossover distortion of the output current i o . 
     Some form of closed-loop control of the d-c bias point of MOSFETs Q 1  and Q 2  is thus required, (a) to maintain a small standing bias current, which is sufficient to avoid crossover distortion, but small enough to avoid significant d-c dissipation in MOSFETs Q 1  and Q 2 , and (b) to maintain the potential of the common point of the floating supply  300  that powers amplifiers  301  and  302  at approximately the midpoint potential between the positive and negative d-c busses, thus ensuring operating headroom as well as substantially equal dissipation and voltage sharing for MOSFETs Q 1  and Q 2 . 
     FIG. 26 shows the added circuits which perform these functions. In FIG. 26, amplifiers  311 ,  312  and  313  are added to the circuit of FIG.  25  and may be included within the IC chip  170 . Thus, referring to FIG. 26, Amplifier  311  senses and amplifies the voltage across R SOURCE +R SENSE . During “passive” periods, when i o  is zero, the voltage across R SENSE  is zero, and the voltage across R SOURCE  is due to standing d-c bias current, i BIAS , that flows from the positive d-c bus to the negative d-c bus, via MOSFETs Q 1  and Q 2 . 
     During the passive periods, the output voltage of amplifier  311  thus represents an amplified inverted version of the voltage across R SOURCE , which itself represents the standing d-c bias current through MOSFET Q 1 . The output voltage of amplifier  311  due to the bias current is less than, but close to, the forward threshold voltage of diode D 1 . This diode therefore does not conduct during the passive periods. 
     During the active periods, when output current i o  flows through MOSFET Q 1 , the voltage at the output of amplifier  311  quickly tries to exceed the forward threshold voltage diode of D 1 , but is clamped to this voltage. The duty cycle of the active periods is very low relative to the passive periods, and thus the average output voltage of amplifier  311  essentially just represents the standing d-c bias voltage across R SOURCE , inverted and multiplied by the gain of amplifier  311 :            e   0          (   311   )       =       -     R9   R8              i   BIAS     ·     R   SOURCE                                
     The output voltage of amplifier  312  then becomes:                  e   0          (   312   )       =         -     V   DD            R12   R11       -       e     0        (   311   )              R12   R10                     =       -     (           e   0          (   311   )          REF     +     e     0        (   312   )           )            R12   R10                     =       -     (           e   0          (   311   )          REF     -       R9   R8            i   BIAS     ·     R   SOURCE           )       ·     R12   R10              
              where                     e   0          (   311   )          REF     =       V   DD          R10   R11         ,   and                                  
     e o ( 311 )REF represents the desired fixed reference value for the standing d-c bias current, i BIAS , and; 
     e o(312) , thus represents the amplified error between the desired and actual d-c bias current. 
     This error voltage is fed to the input of amplifier  301 , via R 13 , such that, in addition to its other functions, amplifier  301  regulates i BIAS  to essentially the set reference level. Note that i BIAS  must of necessity also flow in MOSFET Q 2 , since the capacitor  47  blocks the flow of any d-c current through R SENSE . It remains now to add a means for regulating the common point of the floating power supply substantially to the midpoint potential between the positive and negative d-c bus voltages. This function is carried out by amplifier  313  which amplifies the difference between the positive and negative bus voltages, sensed by resistors R 15  and R 16  respectively. The amplified difference voltage is fed as an input to amplifier  302 , via resistor R 14 , such that, in addition to its other functions, amplifier  302  regulates the common point of the floating power supply substantially to the midpoint potential between the positive and negative bus voltages. 
     The novel invention as described above enables the production of a motor drive circuit which can reduce conductive emission noise and meet the requirements of Class A and Class B motor drives. A further advantage of the novel invention is that ground leak current is reduced to eliminate fault trip of the circuit due to excessive ground current. The reduction of ground leak current is of great importance for the drive of compressor motors using high dielectric constant cooling materials such as R410A which will have an increased capacitance from motor frame to ground. 
     FIG. 27 shows modifications in the circuit of FIG. 26 in which the feedback circuit for quiescent bias current is revised; and FIG. 28 is a detailed overall circuit diagram of the common mode filter of the invention. Referring to FIGS. 27 and 28, the main feedback loop around the amplifiers  301  and  302  is now via R 47  and R 27  to the sources of transistors Q 1  and Q 2  respectively. This feedback, directly to the sources of the MOSFETs, essentially forces the voltages e o301  and e o302  to follow the input voltages e in    301  and e in    302  respectively, overcoming the nonlinear transfer characteristics of the MOSFETs Q 1  and Q 2 , capacitors C 1  and C 13  prevent unwanted oscillation of amplifiers  301  and  302  respectively. Overall linearity between e o  and e in  for each amplifier is significantly improved, versus that obtained with the circuit of FIG.  26 . 
     With the above arrangement of feedback resistor connected directly to the source of the MOSFET, there is no feedback resistor directly around the amplifier itself. Thus the output of amplifier  301  will saturate if e in    301  goes negative, because transistor Q 1  cannot replicate negative input current, and the output of amplifier  302  would saturate if e in    302  goes positive, because transistor Q 2  cannot replicate positive input current. Separate “allowed polarity only” inputs are derived via diode D 4  and R 22  for amplifier  301 , and diode D 5  and R 47  for amplifier  302 . e in    301  is positive during the positive period of i com mot , and zero during negative periods. e in    302  is negative during negative periods of i com mot , and zero during positive periods. 
     Thus the output voltage of amplifier  301  does not saturate when i com mot  is negative, since it receives no input during this period; instead, it stays essentially at the threshold voltage of Q 1 . Likewise, the output voltage of amplifier  302  stays essentially at the threshold voltage of Q 2  when i com mot  is positive. Thus the outputs of each of these amplifiers remain at the desired MOSFET gate threshold level during their idle periods, ready to drive current in their respective MOSFETs at the crossover points of i com mot , with minimal crossover distortion. 
     Note that the secondary winding  44  of the current-sensing transformer is essentially a current source. The voltage across R 22  is therefore directly proportional to i com mot , when this is positive. The voltage across R 47  is directly proportional to i com mot , when this is negative. The voltage drops across diodes D 4  and D 5  do not distort the current waveform, not introduce any significant distortion of the current-dependent signals across R 22  and R 47 . 
     Amplifiers  311  and  312  in FIG. 26 regulate the quiesent bias current via resistor R 13 , amplifier  301  and MOSFET Q 1  The amplifier  313  in FIG. 26 regulates the potential at the common point of the source resistors via resistor R 14 , amplifier  302  and MOSFET Q 2 . These functions are reversed in the circuit of FIG.  28 . Amplifiers U 2  and U 3  (equivalent to amplifiers  311  and  312 ) regulate the quiescent current via R 43  and amplifier U 1 B (equivalent to  302 ) and Q 2 . Amplifier U 4  (equivalent to  313 ) regulates the potential at the common point of the source resistors via resistor R 7 , amplifier U 1 A (equivalent to  301 ) and Q 1 . The swapping of these functions between Q 1  and Q 2  does not change the basic principle but the arrangement of FIG. 28 has been found to be better in practice. 
     With reference to FIG. 26 the clamping diode D 1  across the amplifier  311  is one way of mitigating unwanted output voltage of amplifier  311  when output current flows through Q 1  to ground via  47 . Ideally, however, the output of amplifier  311  should always represent just the quiescent bias current, without any superimposed component of the output current, and should not change during the pulses of output current. In practice, this method of FIG. 26 of “clamping out” the output current component from the quiescent current feedback signal at the output of amplifier  311  generally does not allow sufficiently accurate regulation of the quiescent bias current, because the quiescent bias current feedback signal is corrupted to some degree during each pulse of output current. 
     In the circuits of FIGS. 27 and 28, components R 50 , R 51 , R 52 , D 10  and d 11  are added to solve this problem. Thus, voltage proportional to the output current carried by transistor Q 2  is developed across R 50 . This voltage is applied as a second input to amplifier  311 /U 2 , via R 52 , so that it exactly cancels the voltage developed across R SOURCE (Q2)  (R 30  etc), caused by the output current component, which is applied to the input of amplifier  311  (U 2 ) via R 12 . The net input to amplifier  311  (U 2 ), via the combination of R 12  and R 52 , is thus due only to the quiesent bias current that flows through R SOURCE (Q2)  and Q 2 . It does not flow at the output into R 50  or R 51 . The output of  311  (U 2 ) now stays constant at a level that represents just the quiescent bias current, whether or not output current is flowing. With this arrangement, the clamping diode D 1  in FIG. 26 is no longer needed. 
     Although the present invention has been described in relation to particular embodiments thereof, many other variations and modifications and other uses will become apparent to those skilled in the art. It is preferred, therefore, that the present invention be limited not by the specific disclosure herein.

Technology Category: h