Patent Document

CROSS REFERENCE TO RELATED APPLICATIONS 
   This application a continuation of application Ser. No. 10/357,869, filed on Feb. 3, 2003, now U.S. Pat. No. 6,707,845, which is a continuation of application Ser. No. 10/043,850, filed on Jan. 11, 2002, now issued U.S. Pat. No. 6,516,022, issued on Feb. 4, 2003, which is a continuation of application Ser. No. 09/078,417, filed on May 14, 1998, now issued U.S. Pat. No. 6,366,607, issued on Apr. 2, 2002. 

   BACKGROUND 
   1. Field of the Invention 
   The present invention relates generally to digital communications. More specifically, the invention relates to a system for and method of using a code division multiple access air interface which greatly reduces the signal power required for the global and assigned-pilots while improving performance by using the quadrature phase shift keyed (QPSK) traffic signal for a particular channel to perform channel estimation and carrier recovery. 
   2. Description of the Prior Art 
   Most advanced communication technology today makes use of digital spread spectrum modulation or code divisional multiple access (CDMA). Digital spread spectrum is a communication technique in which data is transmitted with a broadened band (spread spectrum) by modulating the data to be transmitted with a pseudo-noise signal. CDMA can transmit data without being affected by signal distortion or an interfering frequency in the transmission path. 
   Shown in  FIG. 1  is a simplified CDMA communication system that involves a single communication channel of a given bandwidth which is mixed by a spreading code which repeats a predetermined pattern generated by a pseudo-noise (pn) sequence generator. A data signal is modulated with the pn sequence producing a digital spread spectrum signal. A carrier signal is then modulated with the digital spread spectrum signal establishing a forward link, and transmitted. A receiver demodulates the transmission extracting the digital spread spectrum signal. The transmitted data is reproduced after correlation with the matching pn sequence. The same process is repeated to establish a reverse link. 
   During terrestrial communication, a transmitted signal is disturbed by reflection due to varying terrain and environmental conditions and man-made obstructions. This produces a plurality of received signals with differing time delays at the receiver. This effect is commonly known as multipath propagation. Moreover, each path arrives delayed at the receiver with a unique amplitude and carrier phase. 
   To identify the multiple components in the multipath propagation, the relative delays and amplitudes and phases must be determined. This determination can be performed with a modulated data signal, but typically, a more precise rendering is obtained when compared to an unmodulated signal. In most digital spread spectrum systems, it is more effective to use an unmodulated pilot signal discrete from the transmitted modulated data by assigning the pilot an individual pn sequence. A global-pilot signal is most valuable on systems where many signals are transmitted from a base station to multiple users. 
   In the case of a base station which is transmitting many channels, the global-pilot signal provides the same pilot sequence to the plurality of users serviced by that particular base station and is used for the initial acquisition of an individual user and for the user to obtain channel-estimates for coherent reception and for the combining of the multipath components. However, at the required signal strength, the global-pilot signal may use up to 10 percent of the forward direction air capacity. 
   Similar multipath distortion affects a user&#39;s reverse link transmission to the base station. Inserting in each individual user&#39;s return signal an assigned-pilot may consume up to 20 percent of the total reverse channels air capacity. 
   Without phase and amplitude estimation, noncoherent or differentially coherent reception techniques must be performed. Accordingly, there exists a need for a coherent demodulation system that reduces the air capacity of the global-pilot and assigned-pilot signals while maintaining the desired air-interface performance. 
   SUMMARY 
   The present invention is a user equipment (UE), including a receiver and method for receiving one of a plurality of channels in a communication signal. An adaptive matched filter produces a filtered signal by using a weighting signal. A rake receiver produces a filter weighting signal using a pseudo-noise signal generator. A channel despreader despreads the filtered signal using the pseudo-noise signal generated to produce a despread channel signal of the selected channel. A pilot channel despreader despreads the filtered signal using a pseudo-noise signal generator to produce a despread pilot signal of the pilot channel. A hard decision processor receives the despread channel signal of the selected channel and produces a correction signal. A phase-locked loop utilizes at least the despread pilot signal and produces a phase correction signal which is applied to produce phase-corrected channel signals. 
   Accordingly, it is an object of the present invention to provide a code division multiple access communication system which reduces the required global and assigned-pilot signal strength. 
   It is a further object of the invention to reduce the transmitted levels of the global and assigned-pilots such that they consume negligible overhead in the air interface while providing information necessary for coherent demodulation. 
   Other objects and advantages of the system and method will become apparent to those skilled in the art after reading the detailed description of the preferred embodiment. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a simplified block diagram of a typical, prior art, CDMA communication system. 
       FIG. 2  is a detailed block diagram of a B-CDMAJ communication system. 
       FIG. 3A  is a plot of an in-phase bit stream. 
       FIG. 3B  is a plot of a quadrature bit stream. 
       FIG. 3C  is a plot of a pseudo-noise (pn) bit sequence. 
       FIG. 4  is a detailed block diagram of the present invention using one pseudo-pilot signal, with carrier-offset correction implemented at the chip level. 
       FIG. 5  is a block diagram of a rake receiver. 
       FIG. 6  is a diagram of a received symbol p o  on the QPSK constellation showing a hard decision. 
       FIG. 7  is a diagram of the angle of correction corresponding to the assigned symbol. 
       FIG. 8  is a diagram of the resultant symbol error after applying the correction corresponding to the assigned symbol. 
       FIG. 9  is a block diagram of a conventional phase-locked loop. 
       FIG. 10  is a detailed block diagram of the present invention using a pseudo-pilot signal with carrier-offset correction implemented at the symbol level. 
       FIG. 11  is a detailed block diagram of the present invention using a pseudo-pilot signal and the MIPLL, with carrier-offset correction implemented at the chip level. 
       FIG. 12  is a block diagram of the multiple input phase-locked loop (MIPLL). 
       FIG. 13  is a detailed block diagram of the present invention using a pseudo-pilot signal and the MIPLL, with carrier-offset correction implemented at the symbol level. 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   The preferred embodiment will be described with reference to the drawing figures where like numerals represent like elements throughout. 
   A B-CDMAJ communication system  25  as shown in  FIG. 2  includes a transmitter  27  and a receiver  29 , which may reside in either a base station or a mobile user receiver. The transmitter  27  includes a signal processor  31  which encodes voice and nonvoice signals  33  into data at various rates, e.g. data rates of 8 kbps, 16 kbps, 32 kbps, or 64 kbps. The signal processor  31  selects a rate in dependence upon the type of signal, or in response to a set data rate. 
   By way of background, two steps are involved in the generation of a transmitted signal in a multiple access environment. First, the input data  33  which can be considered a bi-phase modulated signal is encoded using forward error-correcting coding (FEC)  35 . For example, if a R=2 convolution code is used, the single bi-phase modulated data signal becomes bivariate or two bi-phase modulated signals. One signal is designated the in-phase channel I  41   a . The other signal is designated the quadrature channel Q  41   b . A complex number is in the form a+bj, where a and b are real numbers and j 2 =−1. Bi-phase modulated I and Q signals are usually referred to as quadrature phase shift keying (QPSK). In the preferred embodiment, the tap generator polynomials for a constraint length of K=7 and a convolutional code rate of R=2 are G 1 =171 8  37 and G 2 =133 8  39. 
   In the second step, the two bi-phase modulated data or symbols  41   a ,  41   b  are spread with a complex pseudo-noise (pn) sequence. The resulting I  45   a  and Q  45   b  spread signals are combined  53  with other spread signals (channels) having different spreading codes, multiplied (mixed) with a carrier signal  51 , and transmitted  55 . The transmission  55  may contain a plurality of individual channels having different data rates. 
   The receiver  29  includes a demodulator  57   a ,  57   b  which mixes down the transmitted broadband signal  55  into an intermediate carrier frequency  59   a ,  59   b . A second down conversion reduces the signal to baseband. The QPSK signal is then filtered  61  and mixed  63   a ,  63   b  with the locally generated complex pn sequence  43   a ,  43   b  which matches the conjugate of the transmitted complex code. Only the original waveforms which were spread by the same code at the transmitter  27  will be effectively despread. Others will appear as noise to the receiver  29 . The data  65   a ,  65   b  is then passed onto a signal processor  59  where FEC decoding is performed on the convolutionally encoded data. 
   As shown in  FIGS. 3A and 3B , a QPSK symbol consists of one bit each from both the in-phase (I) and quadrature (Q) signals. The bits may represent a quantized version of an analog sample or digital data. It can be seen that symbol duration t s  is equal to bit duration. 
   The transmitted symbols are spread by multiplying the QPSK symbol stream by a unique complex pn sequence. Both the I and Q pn sequences are comprised of a bit stream generated at a much higher rate, typically 100 to 200 times the symbol rate. One such pn sequence is shown in FIG.  3 C. The complex pn sequence is mixed with the complex-symbol bit stream producing the digital spread signal. The components of the spread signal are known as chips having a much smaller duration t c . 
   When the signal is received and demodulated, the baseband signal is at the chip level. Both the I and Q components of the signal are despread using the conjugate of the pn sequence used during spreading, returning the signal to the symbol level. However, due to carrier-offset, phase corruption experienced during transmission manifests itself by distorting the individual chip waveforms. If carrier-offset correction is performed at the chip level, it can be seen that overall accuracy increases due to the inherent resolution of the chip-level signal. Carrier-offset correction may also be performed at the symbol level, but with less overall accuracy. However, since the symbol rate is much less than the chip rate, less overall processing speed is required when the correction is done at the symbol level. 
   System architectures for receivers taught in accordance with the system and method of the present invention that do not require large magnitude pilot signals follow. The following systems replace the filtering, despreading and signal processing shown in FIG.  2 . The systems are implemented with carrier-offset correction at both the chip and symbol levels. 
   As shown in  FIG. 4 , a receiver using the system  75  and method of the present invention is shown. A complex baseband digital spread spectrum signal  77  comprised of in-phase and quadrature phase components is input and filtered using an adaptive matched filter (AMF)  79  or other adaptive filtering means. The AMF  79  is a transversal filter (finite impulse response) which uses filter coefficients  81  to overlay delayed replicas of the received signal  77  onto each other to provide a filtered signal  83  having an increased signal-to-noise ratio (SNR). The output  83  of the AMF  79  is coupled to a plurality of channel despreaders  85   1 ,  85   2 ,  85   n  and a pilot despreader  87 . In the preferred embodiment, n=3. The pilot signal  89  is despread with a separate despreader  87  and pn sequence  91  contemporaneous with the transmitted data  77  assigned to channels which are despread  85   1 ,  85   2 ,  85   n  with pn sequences  93   1 ,  93   2 ,  93   n  of their own. After the data channels are despread  85   1 ,  85   2 ,  85   n , the data bit streams  95   1 ,  95   2 ,  95   n  are coupled to Viterbi decoders  97   1 ,  97   2 ,  97   n  and output  99   1 ,  99   2 ,  99   n . 
   The filter coefficients  81 , or weights, used in adjusting the AMF  79  are obtained by the demodulation of the individual multipath propagation paths. This operation is performed by a rake receiver  101 . The use of a rake receiver  101  to compensate for multipath distortion is well known to those skilled in the communication arts. 
   As shown in  FIG. 5 , the rake receiver  101  consists of a parallel combination of path demodulators (Afingers@)  103   0 ,  103   1 ,  103   2 ,  103   n  which demodulate a particular multipath component. The pilot sequence tracking loop of a particular demodulator is initiated by the timing estimation of a given path as determined by a pn sequence  105 . In the prior art, a pilot signal is used for despreading the individual signals of the rake. In this embodiment of the present invention, the pn sequence  105  may belong to any channel  93   1  of the communication system. The channel with the largest received signal is typically used. 
   Each path demodulator includes a complex mixer  107   0 ,  107   1 ,  107   2 ,  107   n , and summer and latch  109   0 ,  109   1 ,  109   2 ,  109   n . For each rake element, the pn sequence  105  is delayed τ  111   1 ,  111   2 ,  111   n  by one chip and mixed  107   1 ,  107   2 ,  107   n  with the baseband spread spectrum signal  113  thereby despreading each signal. Each multiplication product is input into an accumulator  109   0 ,  109   1 ,  109   2 ,  109   n  where it is added to the previous product and latched out after the next symbol-clock cycle. The rake receiver  101  provides relative path values for each multipath component. The plurality of n-dimension outputs  115   0 ,  115   1 ,  115   2 ,  115   n  provide estimates of the sampled channel impulse response that contain a relative phase error of either 0□, 90□, 180□, or 270□. 
   Referring back to  FIG. 4 , the plurality of outputs from the rake receiver are coupled to an n-dimensional complex mixer  117 . Mixed with each rake receiver  101  output  115  is a correction to remove the relative phase error contained in the rake output. 
   A pilot signal is also a complex QPSK signal, but with the quadrature component set at zero. The error correction  119  signal of the present invention is derived from the despread channel  95   1  by first performing a hard decision  121  on each of the symbols of the despread signal  95   1 . A hard decision processor  121  determines the QPSK constellation position that is closest to the despread symbol value. 
   As shown in  FIG. 6 , the Euclidean distance processor compares a received symbol p o  of channel  1  to the four QPSK constellation points x 1, 1 , x −1, 1 , x −1, −1 , x 1, −1 . It is necessary to examine each received symbol p o  due to corruption during transmission  55  by noise and distortion, whether multipath or radio frequency. The hard decision processor  121  computes the four distances d 1 , d 2 , d 3 , d 4  to each quadrant from the received symbol p o  and chooses the shortest distance d 2  and assigns that symbol location x −1, 1 . The original symbol coordinates p o  are discarded. 
   Referring back to  FIG. 4 , after undergoing each hard symbol decision  121 , the complex conjugates  123  for each symbol output  125  are determined. A complex conjugate is one of a pair of complex numbers with identical real parts and with imaginary parts differing only in sign. 
   As shown in  FIG. 7 , a symbol is demodulated or derotated by first determining the complex conjugate of the assigned symbol coordinates x −1, −1 , forming the correction signal  119  which is used to remove the relative phase error contained in the rake output. Thus, the rake output is effectively derotated by the angle associated with the hard decision, removing the relative phase error. This operation effectively provides a rake that is driven by a pilot signal, but without an absolute phase reference. 
   Referring back to  FIG. 4 , the output  119  from the complex conjugate  123  is coupled to a complex n-dimensional mixer  117  where each output of the rake receiver  101  is mixed with the correction signal  119 . The resulting products  127  are noisy estimates of the channel impulse response p 1  as shown in FIG.  8 . The error shown in  FIG. 8  is indicated by a radian distance of π/6 from the in-phase axis. 
   Referring back to  FIG. 4 , the outputs  129  of the complex n-dimensional mixer  117  are coupled to an n-dimensional channel estimator  131 . The channel estimator  131  is a plurality of low-pass filters filtering each multipath component. The outputs of the n-dimensional mixer  117  are coupled to the AMF  79 . These signals act as the AMF  79  filter weights. The AMF  79  filters the baseband signal to compensate for channel distortion due to multipath without requiring a large magnitude pilot signal. 
   Rake receivers  101  are used in conjunction with phase-locked loop (PLL)  133  circuits to remove carrier-offset. Carrier-offset occurs as a result of transmitter/receiver component mismatches and other RF distortion. The present invention  75  requires that a low level pilot signal  135  be produced by despreading  87  the pilot from the baseband signal  77  with a pilot pn sequence  91 . The pilot signal is coupled to a single input PLL  133 . The PLL  133  measures the phase difference between the pilot signal  135  and a reference phase of 0. The despread pilot signal  135  is the actual error signal coupled to the PLL  133 . 
   A conventional PLL  133  is shown in FIG.  9 . The PLL  133  includes an arctangent analyzer  136 , complex filter  137 , an integrator  139  and a phase-to-complex-number converter  141 . The pilot signal  135  is the error signal input to the PLL  133  and is coupled to the complex filter  137 . The complex filter  137  includes two gain stages, an integrator  145  and a summer  147 . The output from the complex filter is coupled to the integrator  139 . The integral of frequency is phase, which is output  140  to the converter  141 . The phase output  140  is coupled to a converter  141  which converts the phase signal into a complex signal for mixing  151  with the baseband signal  77 . Since the upstream operations are commutative, the output  149  of the PLL  133  is also the feedback loop into the system  75 . 
   By implementing the hard decision  121  and derotation  123  of the data modulation, the process provides channel estimation without the use of a large pilot signal. If an error occurs during the hard decision process and the quadrant of the received data symbol is not assigned correctly, the process suffers a phase error. The probability of phase error is reduced, however, due to the increased signal-to-noise ratio of the traffic channel. The errors that occur are filtered out during the channel-estimation and carrier-recovery processes. The traffic channel is approximately 6 dB stronger (2×) than the level of the despread pilot. 
   As described earlier, the present invention can also be performed with carrier-offset correction at the symbol level. An alternative embodiment  150  implemented at the symbol level is shown in FIG.  10 . The difference between the chip and symbol level processes occur where the output of the conventional PLL  133  is combined. At the symbol level, the PLL output  140  does not undergo chip conversion  141  and is introduced into the AMF  79  weights after the rake receiver  101  by another n-dimensional mixer  153 . The phase correction  140  feedback must also be mixed  154   1 ,  154   2 ,  154   n  with the outputs  95   1 ,  95   2 ,  95   n  of each of the plurality of channel despreaders  85   1 ,  85   2 ,  85   n  and mixed  156  with the output  135  of the pilot despreader  87 . 
   As shown in  FIG. 11 , another alternative embodiment  193  uses a variation of the earlier embodiments whereby a hard decision is rendered on each received symbol after despreading and derotated by a radian amount equal to the complex conjugate. The alternate approach  193  uses a plurality of channel despreaders  85   1 ,  85   2 ,  85   n  and the pilot despreader  87  as inputs to a multiple input phase-locked loop (MIPLL)  157  shown in FIG.  12 . Since each of the despread channels  95   1 ,  95   2 ,  95   n  contains an ambiguous representation of the pilot signal, a small signal pilot  135  is required to serve as an absolute reference. The despread symbols from all channels in conjunction with the despread small signal pilot signal are input to the MIPLL  157 . 
   Referring to  FIG. 12 , the output from each channel  95   1 ,  95   2 ,  95   n  is coupled to a hard decision/complex conjugate operation  159   1 ,  159   2 ,  159   n . The derotated pseudo-pilots  161   1 ,  161   2 ,  161   n  are then mixed with the delayed symbols producing a complex voltage error  163   1 ,  163   2 ,  163   n . The error  165   1 ,  165   2 ,  165   n  is input into a converter  167   1 ,  167   2 ,  167   n ,  167   n+1  which takes an inverse tangent converting the complex number into a phase error  169   1 ,  169   2 ,  169   n ,  169   n+1 . Each phase error  169   1 ,  169   2 ,  169   n ,  169   n+1  is input into a maximum likelihood combiner  171  which assigns various weights to the plurality of inputs and produces a sum output. Also included in the sum is the small signal pilot  135  phase  169   n+1  which is despread  135  and converted  167   n+1 . The weighting of the small pilot signal may be emphasized since its phase is unambiguous. 
   The output of the combiner  173  is the estimate of the carrier-offset and is coupled to a complex filter  175  and coupled to an integrator  177 . All channels contribute to the estimate of the carrier-offset frequency with the absolute phase error removed by the unambiguous pilot signal. The integrator accumulates the history of the summed signal over many samples. After integration, the estimate of the phase error is output  179  converted to a complex voltage and output  183 . 
   Referring back to  FIG. 11 , the output  183  of the MIPLL  157  is coupled to a complex mixer  185  upstream of the rake receiver. This completes the error feedback for the MIPLL  157 . Even though this embodiment requires additional resources and complexity, the MIPLL  157  architecture can be efficiently implemented and executed in a digital signal processor (DSP). 
   Referring now to the alternative embodiment 195 shown in  FIG. 13 , this embodiment  195  mixes the output of the MIPLL  157  at the symbol level. The MIPLL  157  is mixed  197  with the output of the rake receiver  101 . As described above, the output of the rake receiver  101  is at the symbol level. The symbol-to-chip conversion  181  in the MIPLL  157  architecture is disabled. Since the output  183  of the MIPLL  157  is mixed with the outputs of the rake  101  which are used only for the AMF  79  weights, the phase correction for carrier-offset must be added to the portion of the receiver that processes traffic data. A plurality of mixers  199   1 ,  199   2 ,  199   n  downstream of each channel despreader  85   1 ,  85   2 ,  85   n  and a mixer  193  downstream of the pilot despreader  87  are therefore required to mix the phase-corrected output  183  (at the symbol level) as feedback into the system. 
   The present invention maintains the transmitted pilot signal at a low level to provide an absolute phase reference while reducing pilot interference and increasing air capacity. The net effect is the virtual elimination of the pilot overhead. 
   While specific embodiments of the present invention have been shown and described, many modifications and variations could be made by one skilled in the art without departing from the spirit and scope of the invention. The above description serves to illustrate and not limit the particular form in any way.

Technology Category: h