Patent Document

BACKGROUND 
   The present invention relates generally to filters and filtering methods, and more particularly, to a parallel decimator adaptive filter and filtering method such as may be used with all-rate gigabit-per-second modems, and the like. 
   Adaptive filtering is necessary in high-rate modems to suppress inter-symbol interference, remove analog filter dispersion, remove adjacent channel interference, perform pulse-shaping and interpolation, and mitigate co-channel interference. Nearly all low-cost digital integrated circuits are built from CMOS technology. Use of conventional technology limits the practical processing rate to about 100 MHz, the maximum clock rate of CMOS technology. There is no known conventional high-speed digital adaptive filter implementation that can operate on input data above 100 MHz. 
   It would be an improvement to have a parallel decimator adaptive filter and filtering method such as may be used with all-rate gigabit-per-second modems, and the like, that improves upon conventional filtering schemes. 
   SUMMARY OF THE INVENTION 
   The present invention provides for a parallel implementation of an adaptive filter and filtering method. The parallel adaptive filter and filtering method enables processing of the signal in a technology (e.g. CMOS) which has an inherent clocking speed many times slower than the filter input rate. The present invention thus provides for low-cost digital implementations of very high-speed modems, for example, that require adaptive or fixed filtering. 
   The present invention is based on an extension of a previously known polyphase decimator structure that has always been applied to a totally different problem. The polyphase decimator structure optimally processes a data stream requiring a low pass filtered bandlimited (i.e., low-rate) output. In the present invention this low-rate output generated by the polyphase filtering structure is used for high-rate output structures. 
   The basic polyphase decimator structure breaks an input stream into parallel paths that efficiently produce a bandlimited (i.e., decimated, low-rate) filtered output. Each of the parallel paths is also processed at a decimated rate. Operating multiple polyphase decimating filter structures allows a non-decimated filtered output to be generated by lower-speed integrated circuits. If partially decimated outputs are desired, the unused polyphase structures may be combined into the structure to extend the effective filter length to provide a sub-set of the filtered outputs. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The various features and advantages of the present invention may be more readily understood with reference to the following detailed description taken in conjunction with the accompanying drawing figures, wherein like reference numerals designate like structural elements, and in which: 
       FIG. 1  illustrates an exemplary parallel decimator adaptive filter in accordance with the principles of the present invention; 
       FIG. 2  illustrates an exemplary adaptive filtering method in accordance with the principles of the present invention; 
       FIG. 3  illustrates an exemplary decimate-by-four polyhase filter structure; and 
       FIG. 4  illustrates an exemplary decimate-by-four polyhase filter structure. 
   

   DETAILED DESCRIPTION 
   Referring to the drawing figures,  FIG. 1  illustrates an exemplary parallel decimator adaptive filter  10  in accordance with the principles of the present invention. The exemplary parallel decimator adaptive filter  10  is adapted to filter input signal that is input at a high clock rate on the order of many gigasamples per second, for example, using relatively low speed circuits that are clocked at a rate on the order of 100 megasamples per second, for example, to produce a filtered output signal at the high clock rate. The exemplary parallel decimator adaptive filter  10  comprises a plurality of polyphase filter integrated circuits  11   a ,  11   b ,  11   c  each having a plurality of inputs  12 . Each polyphase filter integrated circuits  11   a ,  11   b ,  11   c  implements a filter such as is described below. 
   In order to fully understand polyphase filter integrated circuits  11   a ,  11   b ,  11   c , a standard finite impulse response (FIR) filter will be explained. Then, an efficient method of implementing the FIR filter, known as a polyphase filter, will then be explained.  FIG. 3  illustrates an exemplary finite impulse response filter, while  FIG. 4  illustrates an exemplary decimate-by-four polyphase filter structure. and 
   A description of the finite impulse response (FIR) filter is as follows, taken with reference to FIG.  3 . An 8-tap FIR filter structure is shown in  FIG. 3. A  FIR filter has two parts, including a delay section and a multiplier-adder section. The delay section stores a finite number of past input samples. For example, let the number of storage elements be denoted by N. This filter is called an N-tap FIR filter. 
   The input data enters the delay line on the left side of FIG.  3 . On each clock cycle all the data shifts over one delay to the right and the new input enters on the left, the data are shown as s 0 -s 7  in  FIG. 2  (“s” is an abbreviation for sample). The content of each delay element is multiplied by an associated fixed coefficient, labeled c 0 -c 7  in  FIG. 2  (“c” is an abbreviation for coefficient). All the multiplier outputs are summed to form the filtered result. 
   Consider the specific example shown in FIG.  3 . The contents of the shift register are s 7 , s 6 , s 5 , s 4 , s 3 , s 2 , s 1 , s 0  (i.e., samples zero through seven are stored in the shift register). Each of these samples are multiplied by its associated multiplier coefficient within the filter (c 0 , c 1 , c 2 , c 3 , c 4 , c 5 , c 6 , c 7 ) to form the final filter output at time zero, F( 0 ). The final filter output at time zero is:
 
 F ( 0 )=s 0 *c 0 +s 1 *c 1 +s 2 *c 2 +s 3 *c 3 +s 4 *c 4 +s 5 *c 5 +s 6 *c 6 +s 7 *c 7 .
 
   The F( 0 ) filtered result is output. On the next clock cycle the contents of the shift register are right shifted and the next data sample is added on the left side of FIG.  2 . The contents of the register are: 
   s 8 , s 7 , s 6 , s 5 , s 4 ,s 3 , s 2 , s 1 . 
   The filtered output at time  1  is F( 1 ) and is:
 
 F ( 1 )=s 1 *c 0 +s 2 *c 1 +s 3 *c 2 +s 4 *c 3 +s 5 *c 4 +s 6 *c 5 +s 7 *c 6 +s 8 *c 7 .
 
   The filtered outputs at time  2 ,  3 , and  4  are as follows:
 
 F ( 2 )=s 2 *c 0 +s 3 *c 1 +s 4 *c 2 +s 5 *c 3 +s 6 *c 4 +s 7 *c 5 +s 8 *c 6 +s 9 *c 7 ,
 
 F ( 3 )=s 3 *c 0 +s 4 *c 1 +s 5 *c 2 +s 6 *c 3 +s 7 *c 4 +s 8 *c 5 +s 9 *c 6 +s 10 *c 7 ,
 
and
 
 F ( 4 )=s 4 *c 0 +s 5 *c 1 +s 6 *c 2 +s 7 *c 3 +s 8 *c 4 +s 9 *c 5 +s 10 *c 6 +s 11 *c 7 
 
   A description of the polyphase filter is as follows, taken with reference to FIG.  3 . Suppose that a decimated output is required from an FIR filter, say only every fourth output of the FIR filter, for example. A straightforward implementation described with reference to the above-described FIR filter would be very inefficient because it computes F( 0 ), F( 1 ), F( 2 ), F( 3 ), F( 4 ), and so forth. However, only F( 0 ), F( 4 ), F( 8 ), . . . are desired (i.e., a decimated by 4 output). A polyphase filter provides an efficient way to compute a decimated output of a prototype FIR filter. The original filter and its coefficients are referred to as the “prototype” filter. The polyphase filter implementation rearranges the prototype filter so that it efficiently implements only the decimated outputs of the prototype filter. 
   Consider the polyphase structure shown in FIG.  4 . This decimate by 4 polyphase structure includes four length-2 FIR filters (called polyphase filters). Note that the total number of coefficients in all of the four polyphase filters matches the total number of coefficient in the prototype filter (i.e., 2*4=8). The polyphase filter coefficients are simply decimated versions of the prototype filter. Decimating the filter coefficients of the prototype filter means taking every fourth filter coefficient. The first polyphase filter takes coefficients 0 and 4 from the prototype filter. The second polyphase filter takes coefficients 1 and 5 from the prototype filter. The third polyphase filter takes the 2 and 6 th  coefficient from the prototype filter. The fourth polyphase filter takes coefficients 3 and 7 from the prototype filter. 
   The left side of  FIG. 4  is the input commutator. The first sample from the commutator feeds the lower 2-tap polyphase filter. The next sample from the commutator feeds the third from top polyphase filter. The third sample from the commutator feeds the second from the top polyphase filter. The fourth sample feeds the top polyphase filter. After all four commutator input samples are collected, a single output sample is generated (i.e., every fourth input clock a single decimated output is formed). The single output is the sum of the output of all four length-2 polyphase filters. 
   Consider the output this filter structure produces. The lower polyphase filter produces the output s 0 *c 0 +s 4 *c 4 . The third from the top polyphase filter produces the output s 1 *c 1 +s 5 *c 5 . The second from the top polyphase filter produces the output s 2 *c 2 +s 6 *c 6 . The top polyphase filter produces the output s 3 *c 3 +s 7 *c 7 . The sum of all these outputs is:
 
 P ( 0 )=s 0 *c 0 +s 1 *c 1 +s 2 *c 2 +s 3 *c 3 +s 4 *c 4 +s 5 *c 5 +s 6 *c 6 +s 7 *c 7 = F ( 0 ).
 
   P( 0 ) exactly matches the output of the prototype filter at time  0 , F( 0 ). Thus, the structure is equivalent to the output of the prototype filter at time  0 . Further analysis will show that the filter structure matches the output of the prototype filter at every fourth output sample. The second output of the polyphase filter is:
 
 P ( 1 )=s 4 *c 0 +s 5 *c 1 +s 6 *c 2 +s 7 *c 3 +s 8 *c 4 +s 9 *c 5 +s 10 *c 6 +s 11 *c 7 = F ( 4 ).
 
   The advantage of the polyphase filter is now clear. At each new input sample to the polyphase filter only a 2-tap FIR filter must be update, versus a 8-tap FIR filter for the direct implementation of the prototype FIR filter. Also, each polyphase filter must only be updated every 4-clock cycles since this is the revisit rate of the input commutator. After all four inputs from the commutator are processed, and then the outputs of all four 2-tap polyphase filters are summed to form the decimated output. 
   The reason people use polyphase filters for decimator filters is that it is D times more efficient then the direct implementation of the prototype filter, where D is the decimation ratio. Only 1 polyphase filter with 2-taps must be evaluated every input clock cycle, versus the 8-taps that must be evaluated if the direct prototype filter is implemented. 
   Alternatively, four pairs of multipliers can be run in parallel to achieve a high-speed filter where each of the four parallel polyphase filters only runs at one-fourth of the input data rate. This allows high-rate input data to be processed by relatively low-rate filter circuits behind the commutator. 
   If a full-rate output is desired, the additional outputs can be generated (i.e., sample 0 plus samples 1 or 2 or 3) by adding additional polyphase filter structures in parallel with the one shown in  FIG. 3 , with input delays of 1, 2, and 3 respectively. Thus, if four filters are run in parallel then the output of the aggregate is a full-rate filter output. However, each of the four parallel circuits is running at one-fourth of the input data rate. Thus, it is possible to build a high-speed modem with parallel processing in lower-rate back-end polyphase filters. 
   Referring again to  FIG. 1 , each of the plurality of inputs of the polyphase filter integrated circuits  11   a ,  11   b ,  11   c  comprise a filter bank  11 . Each of the plurality of inputs of the polyphase filter integrated circuits  11   a ,  11   b ,  11   c  are fed from a commutator  13  having a single input  14  and a plurality of commutated outputs  14   a.    
   Each of the polyphase filter integrated circuits  11   a ,  11   b ,  11   c  or filter banks  11  comprise a plurality of short-length polyphase finite impulse response (FIR) filters  15 . Outputs of the plurality of short-length polyphase FIR filters  15  of each of the polyphase filter integrated circuits  11   a ,  11   b ,  11   c  or filter banks  11  are combined (added) to produce a single output  16  of each respective polyphase filter integrated circuit  11   a ,  11   b ,  11   c.    
   Signals supplied to the first polyphase filter integrated circuit  11   a  is not delayed. Signals supplied to the second and third polyphase filter integrated circuits  11   b ,  11   c  are coupled through and delayed by first and second pluralities of delay elements  17   a ,  17   b  respectively. For example, in the exemplary filter  10 , the input data from the commutator  13  is delayed by two samples by the first delay elements  17   a  prior to being input to the second filter bank  11  comprising the second polyphase filter integrated circuit  11   b . The input data from the commutator  13  is delayed by four samples by the second delay elements  17   b  prior to being input to the third filter bank  11  comprising the third polyphase filter integrated circuit  11   c.    
   In operation, an input signal arrives at the input  14  of the commutator  13  at a predetermined input rate R. For example, an input rate of 400 Megasamples per second (Msps) may be used. The input signal is commutated into N parallel paths  14   a  (six paths are shown in the exemplary filter  10 ) by the commutator  13 . The commutated input data is fed to three (N/2) sets of filter banks  11  or polyphase filter integrated circuits  11   a ,  11   b ,  11   c  that each comprise the plurality of short-length polyphase FIR filters  15 . Each filter bank  11  operates in parallel with the other filter banks  11 , generating one of N/2 filtered output signals  16  required every N input samples. 
   The input data from the commutator  13  is not delayed prior to being input to the first filter bank  11 . The input data from the commutation  13  is delay by 2 samples prior to being input to the second filter bank  11 . The input data from the commutator is delayed by 4 samples prior to being input to the third filter bank  11 . Each output of the respective filter banks  11  is formed by adding the outputs of the six (N) short-length polyphase FIR filters  15 . Each polyphase filter  15  operates at a rate equal to R/N. 
   The present invention also allows the filter length to increase as the bandwidth of the signal decreases (all-rate operation). That is, when only a sub-set of the outputs are required, the computing resources of the unused outputs are directed to assist the desired outputs. For example, if one of the N/2 filter outputs  16  is not needed in a particular application, its polyphase filters  15  may be appended to one of the other filter banks  11  to extend the effective filter length of the filter banks  11  that remain in operation. 
   The present invention thus provides for operation of multiple decimators (the filter banks  11 ) in parallel to enable non-decimated filtering beyond the limit of the technology that is used to produce the filter  10 . For example, the present invention allows filtering of giga-bit-per-second signals with low-cost CMOS integrated circuits as the respective filter banks  11  that have a maximum clock rate of 100 MHz. Thus, the filter  10  has a systolic, parallel architecture structure that is amenable to ASIC implementation and allows a vast reduction in the number of integrated circuit chips required to process high-rate signals. 
   In addition, the use of filter coefficient double buffering allows filter coefficients to be changed “on-the-fly”, enabling adaptive filter coefficient updates. Also, high-rate transmitter pulse shaping and interpolation by two may be implemented using the polyphase decimator structure of the present filter  10 . 
     FIG. 4  illustrates an exemplary adaptive filtering method  30  in accordance with the principles of the present invention. The exemplary adaptive filtering method  30  comprises the following steps. 
   An input signal to be filtered is commutated  31  to provide a plurality of commutated signals. A plurality of parallel filter banks  11  are provided  32  that each comprise a plurality of polyphase filters  15 . The plurality of commutated signals are filtered  33  using the plurality of polyphase filters of a first filter bank to generate a first plurality of filtered signals. Each of the plurality of commutated signals are delayed  34  by a first delay value and each of the plurality of delayed commutated signals is adaptively filtered  35  using the plurality of polyphase filters of a second filter bank to generate a second plurality of filtered signals. Each of the plurality of commutated signals is delayed  36  by a second delay value and each of the plurality of delayed commutated signals is adaptively filtered  37  using the plurality of polyphase filters of a third filter bank to generate a third plurality of filtered signals. The first, second and third pluralities of filtered signals are respectively combined  38  to produce first, second and third filtered output signals that comprise a filtered version of the input signal. 
   Thus, a parallel decimator adaptive filter for use with all-rate gigabit-per-second modems, and the like, has been disclosed. It is to be understood that the described embodiment is merely illustrative of some of the many specific embodiments that represent applications of the principles of the present invention. Clearly, numerous and other arrangements can be readily devised by those skilled in the art without departing from the scope of the invention.

Technology Category: h