Patent Document

CROSS-REFERENCE TO RELATED APPLICATION(S) 
     This application is a divisional application of U.S. patent application Ser. No. 13/298,254, filed Nov. 16, 2011, which claims benefit of U.S. Provisional Patent Application No. 61/429,034 filed Dec. 31, 2010, incorporated herein by reference. 
    
    
     BACKGROUND 
     1. Technical Field 
     Embodiments relate generally to communications systems and, more specifically, to demodulation circuits having interference cancellation and improved signal-to-noise ratio performance. 
     2. Description of the Related Art 
     In both narrowband and wideband communications systems interference cancellation is the process of removing or “cancelling” the degrading effects of a high power interfering signal that is spectrally adjacent to the input signal of interest (i.e., the desired input signal). This degradation is measured in terms of a reduced bit error rate (BER), reduced distance over which input signals may be communicated over the corresponding communications channel, and increased guard band protection resulting in lower efficiency utilization of the frequency or band of interest, as will be appreciated by those skilled in the art. The use of interference cancellation is prevalent in wideband systems such as cellular telephone systems, terrestrial wireless systems employing IEEE802.16 protocols, and in systems where a wideband, high power transmitter is co-located with a receiver bank, as will also be appreciated by those skilled in the art. In many situations the type or characteristics of the interference is known, or in other situations the interference cancellation system attempts to adaptively characterize or “learn” the type of interference prior to implementing approaches to remove or cancel this interference. There is a need for improved methods, circuits, and systems for interference cancellation in communications systems. 
     BRIEF SUMMARY 
     Embodiments are directed to circuits, systems, and methods of interference cancellation for wideband and narrowband communications systems without a priori knowledge of statistical information about an interfering signal. According to one embodiment, a demodulator circuit can operate in an environment where a “no lock” situation would normally occur to remove the interference and acquire signals in low signal-to-noise ratio (SNR) conditions and high signal-to-interference ratio (SIR) conditions. In other embodiments, performance is improved by introducing statistics of the interfering signal, and these statistics regarding the communications channel and interference properties (i.e., characteristics of the interfering signal) can be adaptive or “learned.” 
     According to one embodiment, a demodulation circuit includes an interference estimation circuit adapted to receive an input signal and generate an interference estimation signal from the input signal and an adaptive filtering circuit coupled to the interference estimation circuit and operable to perform spectral inversion on the estimation signal to obtain an inverted signal that is then applied to the input signal to substantially cancel interference contained in the input signal. 
    
    
     
       BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS 
         FIG. 1  is a functional block diagram of a demodulation circuit including an interference estimation circuit and adaptive filtering circuit according to one embodiment. 
         FIG. 2  is a graph illustrating characteristics of the input signal received by the interference estimation circuit of  FIG. 1 . 
         FIG. 3  is a graph illustrating the operation of the complex multiplier, adaptive low pass filter, and I-Q phase order filter in the interference estimation circuit of  FIG. 1  in isolating a portion of the interfering signal and the desired signal of the input signal of  FIG. 2 . 
         FIG. 4  is a graph showing the operation of the complex multiplier, adaptive low pass filter, and I-Q phase order filter of the adaptive filtering circuit of  FIG. 1  in isolating the portion of the interfering signal of  FIG. 3 . 
         FIG. 5  is a graph showing the spectral inversion of the portion of the interfering signal of  FIG. 4  that is performed by the I-Q phase order filter contained in the adaptive filtering circuit of  FIG. 1 . 
         FIG. 6  is a graph showing the operation of the second complex multiplier, sin c(x) compensation circuit, and digital-to-analog (DAC) converter contained in the interference estimation circuit of  FIG. 1  in re-modulation of the input signal. 
         FIG. 7  is a graph showing the spectrally inverted estimate of the portion of the interfering signal of  FIG. 5  after digital-to-analog conversion that is output from the adaptive filtering circuit of  FIG. 1 . 
         FIG. 8  is a graph showing the compensated input signal obtained by summing the signals of the graphs of  FIGS. 6 and 7  to thereby remove or significantly reduce the interfering signal and on which on which analog-to-digital conversion and demodulation is ultimately performed by the demodulation circuit of  FIG. 1 . 
         FIG. 9  is a functional block diagram illustrating an all-digital implementation of the interference estimation circuit and adaptive filtering circuits of  FIG. 1  according to another embodiment. 
         FIG. 10  is a functional block diagram of an electronic system such as a communications system including the demodulation circuit of  FIG. 1  according to another embodiment. 
         FIG. 11  is an alternative graph illustrating the desired signal S and interfering signal I present on the input signal of  FIG. 1 . 
         FIG. 12  is an alternative graph illustrating the input signal of  FIG. 11  after the estimate of the interfering signal generated by the interference estimation circuit of  FIG. 1  has been summed with the input signal to cancel or greatly reduce the interfering signal. 
         FIG. 13  is a polar graph illustrating the vector representation of the estimate of the interfering signal and the vector representation of the interfering signal of  FIGS. 11 and 12 . 
         FIG. 14  is a graph illustrating a QPSK example for the concept of  FIG. 13  and the interference estimation circuit of  FIG. 1  in providing an estimate of the interfering signal to cancel the interfering signal present on the input signal of  FIG. 1 . 
         FIG. 15  is a graph illustrating the improved signal-to-interference ratio (SIR) of the input signal after processing of the input signal by the interference estimation circuit and adaptive filtering circuit of  FIG. 1 . 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 1  is a functional block diagram of a demodulation circuit  100  including an interference estimation circuit  102  and an adaptive filtering circuit  104  contained in the demodulation circuit according to one embodiment. In operation, the interference estimation circuit  102  and an adaptive filtering circuit  104  cancel an interfering signal I present on an input signal  106  applied to the demodulation circuit  100 , as will be explained in more detail below. The interference estimation circuit  102  and adaptive filtering circuit  104  detect and remove the interfering signal I on the input signal  106  without prior knowledge of the characteristics of the interfering signal. The adaptive filtering circuit  104  also utilizes “coherent” “knowledge” or information as derived from the demodulation circuit  100  to further remove degradation effects due to the interfering signal I, as will be described in more detail below. 
     In the present description, certain details are set forth in conjunction with the described embodiments to provide a sufficient understanding of the invention. One skilled in the art will appreciate, however, that the invention may be practiced without these particular details. Furthermore, one skilled in the art will appreciate that the example embodiments described below do not limit the scope of the present disclosure, and will also understand that various modifications, equivalents, and combinations of the disclosed embodiments and components of such embodiments are within the scope of the present disclosure. Embodiments including fewer than all the components or steps of any of the respective described embodiments may also be within the scope of the present disclosure although not expressly described in detail below. Finally, the operation of well-known components and/or processes has not been shown or described in detail below to avoid unnecessarily obscuring the present disclosure. 
     The demodulation circuit  100  receives the input signal  106  which includes a known signal of interest or desired signal S and the interfering signal I. This concept is illustrated in the graph of  FIG. 2  showing the input signal  106 . The interference estimation circuit  102  executes a “non-coherent” process while the adaptive filtering circuit  104  executes a coherent process in removing the unwanted interfering signal I from the input signal  106 . In the example of  FIG. 2 , the input signal  106  is assumed to include the desired signal S and the interfering signal I to include adjacent interfering signals as illustrated. As will be discussed in more detail below with regard to  FIG. 11 , the signal-to-interference ratio (SIR) is the ratio of the average received modulated carrier power (i.e., power of desired signal S) to the average received co-channel interference power (i.e., power of interfering signal I). The co-channel interference is crosstalk from two different radio transmitters using the same frequency as a primary receiver containing the demodulation circuit  100  of  FIG. 1 , as will be understood by those skilled in the art. Also note that the desired signal S corresponds to the modulated signal that is transmitted by a primary transmitter and that is intended to be received by the primary receiver containing the demodulation circuit  100 . The desired signal S portion of the input signal  106  contains data to be received by the primary receiver containing the demodulation circuit  100 . The unwanted interfering signal I corresponds to a signal or signals from a secondary transmitter or transmitters that may also be received by the primary receiver, and which accordingly can degrade the performance of the primary receiver. Note that the term “data” that is being communicated through the desired signal S is used broadly herein to include any type of data, such as audio data, video data, programming instructions, communications protocol related information, and so on. 
     In  FIG. 1 , the input signal  106  may be represented by the signal structure shown in  FIG. 2 . The structure of the input signal  106 , as shown in  FIG. 2 , includes the desired signal S and the unwanted interfering signal I. In the example of  FIG. 2  the unwanted interfering signal I is assumed to include adjacent interfering signals as illustrated. Initially, the input signal  106  is filtered by a first varactor variable filter  108  that performs some initial “tuning” or filtering of the input signal  106  in order to coarsely “tune” the demodulation circuit  100  such that the input signal contains only the desired signal S and the interfering signal I that is to be removed from the input signal. This is so that subsequent processing by the remaining components of the interference estimation circuit  102  can remove the interfering signal I, as will be described in more detail below. Thus, the varactor variable filter  108  can be viewed as filtering the input signal  106  to isolate the interfering signal I that is to be removed from the input signal. It should be noted, however, the input signal  106  may of course include other interfering signals or noise not shown in  FIG. 2 , and in this situation the varactor variable filter  108  filters out such other interfering signals and noise such that the output of the varactor variable filter includes both the interfering signal I to be removed and the desired signal S as seen in  FIG. 2 . 
     An analog-to-digital converter (ADC)  110  samples the filtered input signal from the varactor variable filter  108 , with this sampling adhering to the requirements of the Nyquist sampling theorem, and outputs digital values corresponding to these samples. An I-Q clock generator  112  generates a plurality of clock signals that are applied to appropriately clock the ADC  110  and other components in the interference estimation circuit  102 . As seen in  FIG. 2 , the desired signal S has a bandwidth BW_ 1  and the clock signals from the clock generator  112  have a frequency that enables the entire spectrum of interest to be sampled by the ADC  110 . In one embodiment, the clock generator  112  applies clock signals to the ADC  110  having a frequency corresponding to at least four times the bandwidth BW_ 1  of the desired signal S. 
     A complex multiplier  114  receives these digital values from the ADC  110  and performs complex multiplication on these digital values to thereby effectively multiply this digital signal into baseband and form an equivalent I-Q sample set, as will be discussed in more detail below. An adaptive low pass filter  116  receives the I-Q sample set from the complex multiplier  114  and this filter in combination with an I-Q phase order filter  118  operate to filter this sample set to thereby isolate the multiplication images generated by the complex multiplication, and provide passband shaping in the form of adaptive filtering. In this way, the I-Q phase order filter  118  outputs an estimate of the interfering signal I to be removed as seen in  FIG. 3 . As seen in  FIG. 3 , the low pass filtering by the adaptive low pass filter  116  and operation of the I-Q phase order filter  118  results in an in-band portion (at center frequency fc-F 1 ) of the interfering signal I being retained while an out-of-band portion (at center frequency fc-F 2 ) of the interfering signal is rejected. The frequency fc is the center frequency of the desired signal S having bandwidth BW_ 1  as shown in  FIG. 3 . The in-band portion of the interfering signal I corresponds to that portion below a cutoff frequency as indicated by the dotted line in  FIG. 3 , and the out-of-band portion of the interfering signal corresponds to the portion above this cutoff frequency. 
     The output of the I-Q phase order filter  118  is supplied to a second complex multiplier  120  that is clocked by a clock generated by a second I-Q clock generator  122 . This second I-Q clock generator  122  generates a clock that is corrected based upon corrections being applied by a demodulator circuit  124  on a symbol-by-symbol basis. More specifically, the demodulator circuit  124  demodulator provides symbol clock information to a numerically controlled oscillator (NCO)  126  which operates at a multiple of the symbol clock frequency and operates in combination with a fine phase adjustment circuit  128  to fine tune phase correction of the clock generated by the second I-Q clock generator  122 . This coupling of the demodulator circuit  124  and the coherent symbol clock via the NCO  126  and fine phase adjustment circuit  128  functions to perform the coherent removal of interference. The demodulator circuit  124  is capable of locking and synchronizing due to the first stage non-coherent interference reduction performed by the interference estimation circuit  102 , as will be described in more detail below. 
     The second complex multiplier  120  receives the output from the I-Q phase order filter  118  an complex multiplies responsive to the clock signals from the second I-Q clock generator  122 . These clock signals from the second I-Q clock generator have a frequency of approximately (fc-F 1 ), where recall as discussed above the frequency F 1  is the frequency of the in-band portion of the interfering signal I that is being removed. An adaptive low pass filter  130  then filters the output from the complex multiplier  120  to thereby isolate the in-band portion of the interfering signal I as shown in  FIG. 4  by the dotted line. Thus, the output of the adaptive low pass filter  130  corresponds to the in-band portion of the interfering signal I that is being removed. A second I-Q phase order filter  132  receives the in-band interfering signal I from the adaptive low pass filter  130  (see  FIG. 4 ) and functions to perform spectral and amplitude inversion of the in-band interfering signal I about the center frequency (fc-F 1 ) of this signal. This is illustrated in  FIG. 5 , with the arrows illustrating the spectral inversion performed by the second I-Q phase order filter  132 . 
     At this point, the spectrally inverted in-band interference signal I output by the I-Q phase order filter  132  as illustrated in  FIG. 5  is an estimate of the in-band portion of the interfering signal I to be removed. This estimate from the I-Q phase order filter  132  is then re-modulated so that it can then be subtracted from the input signal  106  prior to being demodulated by the demodulator  124 , as will now be described in more detail. In order to do so, as seen in  FIG. 1  the spectrally inverted in-band interference signal I output by the I-Q phase order filter  132  is supplied to a third complex multiplier  134  that performs complex multiplication on the signal estimate from the I-Q phase order filter  132  to return the signal estimate to its original spectral center frequency. A sin(x)/x compensation circuit  136  then receives the signal estimate from the complex multiplier  134  and filters that estimate, with the filtered estimate being supplied to a first digital-to-analog (DAC) converter  138 . The compensation circuit  136  filtering ensures that the resultant estimate of the interfering signal I output by the digital-to-analog converter (DAC)  138  does not spectrally spill into the frequency band containing the signal of interest or desired signal S. The estimate of the interfering signal I output from the DAC  138  may be referred to as the “non-coherent estimate of the in-band interfering signal I” in the discussion below. 
     In a similar way, a complex multiplier  140 , sin(x)/x compensation circuit  142 , and digital-to-analog converter (DAC)  144  operate in combination to receive the output signal from the I-Q phase order filter  118  and to re-modulate this signal to the passband. Thus, the output of the DAC  144  essentially represents the original input signal  106  supplied to the ADC  110 .  FIG. 6  illustrates the signal output from the DAC  144 , and by comparing  FIG. 6  to  FIG. 2  this is seen to be the case.  FIG. 7  illustrates the non-coherent estimate of the in-band interfering signal I output from the DAC  138  resulting from the operation of the operation of the complex multiplier  134 , compensation circuit  136 , and DAC  138 . A summation circuit  146  sums the outputs from the DAC  144  and the DAC  138  and outputs this sum as an interference-corrected signal as illustrated in  FIG. 8 . A negative sign at the DAC  138  output being supplied to the summation circuit  146  indicates that the output from the DAC  138  is the non-coherent estimate of the in-band interfering signal I, which is the re-modulated spectrally inverted estimate of the in-band interference signal I. 
       FIG. 8  illustrates that the interference-corrected signal output from the summation circuit  146  has a greatly reduced in-band portion of the interfering signal I. Note that the out-of-band portion of the interfering signal I, namely that portion at center frequency fc-F 2 , remains in the interference-corrected signal output from the summation circuit  146  as shown in  FIG. 8 . If the removal of this out-of-band portion of the interfering signal I, or of other portions (not shown in the figures) is desired, the series-connected components  120 - 138  contained in the adaptive filtering circuit  104  are simply duplicated for each such portion to be removed. The sampling frequency and thus the frequency of the clock signals applied by the I-Q clock generator  122  to clock each such group of series-connected components is adjusted accordingly to thereby remove the desired portion of the interfering signal I. For example, if the output-of-band portion having bandwidth BW_ 3  shown in  FIG. 8  is desired to be removed, the series-connected components  120 - 138  are duplicated and the sampling frequency adjusted accordingly, with the output of the DAC being another input to the summation circuit  146  to thereby remove the out-of-band portion of the interfering signal I shown in  FIG. 8 . 
     The interference-corrected signal output from the summation circuit  146  is input to an analog-to-digital converter (ADC)  148  that samples and digitizes this signal and provides corresponding digital values to the demodulator circuit  124  which, in turn, demodulates these digital values to obtain the original unmodulated I-Q encoded data. 
     In another embodiment, the interference cancellation circuit  102  of  FIG. 1  further includes a delay circuit  150  including a second varactor variable filter  152 , analog delay line  154 , and summation circuit  156  coupled in series as shown. The varactor variable filter  152  is tuned to have a center frequency of the desired signal S and thus provides a delayed version of this signal to the summation circuit  156 . Also in this embodiment, the variable varactor filter  108  is tuned such that the it passes the interfering signal I. In this embodiment the input signal  106  is filtered through the alternative path via the second programmable varactor tuned filter  152  and the analog delay line  154 . This filter  152  isolates the desired signal in frequency and is delayed through the analog delay line  154  to compensate for the computation time required to compute the non-coherent estimate of the interfering vector by components  120 - 138 . It must be noted, that the delay introduced by the analog delay line  154  is to be evaluated on an application by application basis, and if clocking constraints are not encountered then this delay circuit  150  is not required. In the first described embodiment, the required delay in incorporated into the adaptive low pass filter  116  and I-Q phase order filter  118 . 
       FIG. 9  is a functional block diagram of a demodulation circuit  900  illustrating an all-digital implementation of the interference estimation circuit  902  and adaptive filtering circuit  904  of  FIG. 1 . The theory of operation of this embodiment is the same as previously described for the demodulation circuit  100  of  FIG. 1 . The components  900 - 932  operate in a similar manner to the corresponding components  100 - 132  of the demodulation circuit  100 , and the demodulation circuit additionally includes a summation circuit  933 , sample alignment circuit  935 , and summation circuit  937  that operate on the corresponding digital values. Also, the demodulation circuit  900  includes components  920   a - 932   a  and  920   b - 932   b  that are coupled in parallel with outputs summed by the summation circuit  933  to thereby remove both the lower and upper side bands, which correspond to both the in-band and out-of-band portion of the interfering signal illustrated and described with reference to  FIGS. 1-8 . 
       FIG. 10  is a functional block diagram of an electronic system  100  such as a communications system including the demodulation circuit  100  of  FIG. 1  according to another embodiment. An information source provides information to a source coding component  1002 , which suitable encodes the information and provides the encoded information to a channel coding component  1004 . The channel coding component likewise performs suitable channel coding on the received information and provides this encoded information to a modulator  1006 . The modulator  1006  modulates the encoded information from the channel coding component  1004  and communications the suitably modulated information over a communications channel  1008 , such as a wireless communications channel. A demodulator  1010  including the demodulation circuit  100  or  900  demodulates the received input signal from the communication channel  108  and provides this demodulated information to a channel decoding component  1012  and source decoding component  1014  which function to reverse the operations of the components  1004  and  1002 . The source decoding component  1014  outputs received information which ideally corresponds to the information source supplied to the source coding component  1002 . 
     The demodulation circuit  100  described in  FIG. 1  can be significantly simplified if all the signal processing is accomplished in the digital domain. Once the ADC  110  has sampled the input signal  106 , processing can be accomplished entirely digitally on two parallel paths.  FIG. 10  shows the simplified digital cancellation circuit while the theory of operation is the same. The ADC  110  samples at 4 or 8 times the symbol rate of the input signal  106  or signal of interest. Removal of expensive additional ADCs and DACs is thus possible. Adjacent carriers, meaning the in-band and out-of-band portions of the interfering signal I need not be of the same modulation or format and the sampling process can operate as a coherent or non-coherent process. 
       FIG. 11  is an alternative representation of the input signal  106 .  FIG. 12  shows the interfering signal I canceled output. This block will take a signal to interference ratio of 30 to 40 dB and reduce the interference level to a range 5 to 15 dB.  FIG. 13  shows the vector relationship of the input interference vector and the estimated cancellation vector (output from DAC  138 ). The jitter reflects the phase and amplitude estimate of the non-coherent computation and the fact that the clocking is non-coherent. 
     The adaptive filtering circuit of  FIG. 1  performs the addition of “coherent cancellation and SNR enhancement”. The non-coherent stage is a coarse method to remove interference. It allows for removal of energy, in the order of 10-30 dB a reduction sufficient to allow a demodulator or related receiver to acquire synchronization and lock. When the interfering signal and the desired signal are within 0 to 10 dB of each other in terms of relative power, the non-coherent method is not significantly effective. The coherent method offers significant other improvement towards enhancing the SNR and removing residual interference capable of degrading the receiver/demodulator performance. 
       FIG. 14  shows the interference cancelling concept within a QPSK constellation. In the top right hand corner of  FIG. 5  the vector corresponding to a synchronized sample. The ideal reference constellation points are shown in this figure. Several samples representing noise and channel degradations are shown clustered about the reference constellation point. Also shown in this figure is the interfering vector in the bottom right quadrant. The interference cancellation vector, corresponding to the output form the DAC  138 , is also shown. This vector is rotating in the opposite direction of the interfering vector, with opposite amplitude and phase. Note that it is only required to cancel the interference at the sampling instance of time.  FIG. 14  shows the output of the summation circuit  146  ( FIG. 1 ). This embodiment of the demodulation circuit  100 / 900  provides interference and SNR enhancement as well as allowing the ADC  148  having a lower number of bits to be utilized. 
     Analysis and removal of in band interference is possible with the demodulation circuits  100 / 900  of  FIGS. 1 and 9 . The adaptive filter  130  can be structured to remove both in band and out of band interference. 
     The following equations provide a mathematical foundation for the operation of the previously described embodiments: 
     Input (0,  FIG. 1 ): Signal at ω 1 , represents interferer, Signal at ω 2 , represents desired signal
 
 A   1  cos(ω 1   t+φ   1 )+ A   2  cos(ω 2   t+φ   2 )  (1)
 
     ADC  110 : 
     
       
         
           
             
               
                 
                   
                     
                       
                         A 
                         1 
                       
                       ⁢ 
                       
                         cos 
                         ⁡ 
                         
                           ( 
                           
                             
                               
                                 ω 
                                 1 
                               
                               ⁢ 
                               t 
                             
                             + 
                             
                               φ 
                               1 
                             
                           
                           ) 
                         
                       
                     
                     + 
                     
                       
                         A 
                         k 
                       
                       ⁢ 
                       
                         cos 
                         ⁡ 
                         
                           ( 
                           
                             
                               
                                 ω 
                                 2 
                               
                               ⁢ 
                               t 
                             
                             + 
                             
                               φ 
                               2 
                             
                           
                           ) 
                         
                       
                     
                   
                   ⁢ 
                   
                     
 
                   
                   ⁢ 
                   
                     
                       A 
                       k 
                     
                     = 
                     
                       
                         A 
                         2 
                       
                       k 
                     
                   
                 
               
               
                 
                   ( 
                   2 
                   ) 
                 
               
             
           
         
       
     
     Complex Multiplier  114 :
 
[ A   1  cos(ω 1   t+φ   1 )+ A   k  cos(ω 2   t+φ   2 )][cos(ω s   t+φ   s )+sin(ω s   t+φ   x )]  (3)
 
     I-Q Phase Order Filter  118 : 
     
       
         
           
             
               
                 
                   
                     
                       A 
                       1 
                     
                     2 
                   
                   ⁡ 
                   
                     [ 
                     
                       
                         cos 
                         ⁡ 
                         
                           ( 
                           
                             
                               
                                 ( 
                                 
                                   
                                     ω 
                                     s 
                                   
                                   - 
                                   
                                     ω 
                                     1 
                                   
                                 
                                 ) 
                               
                               ⁢ 
                               t 
                             
                             + 
                             ϕ 
                           
                           ) 
                         
                       
                       - 
                       
                         j 
                         ⁢ 
                         
                             
                         
                         ⁢ 
                         
                           sin 
                           ⁡ 
                           
                             ( 
                             
                               
                                 
                                   ( 
                                   
                                     
                                       ω 
                                       s 
                                     
                                     - 
                                     
                                       ω 
                                       1 
                                     
                                   
                                   ) 
                                 
                                 ⁢ 
                                 t 
                               
                               + 
                               ϕ 
                             
                             ) 
                           
                         
                       
                     
                     ] 
                   
                 
               
               
                 
                   ( 
                   4 
                   ) 
                 
               
             
           
         
       
     
     Complex Multiplier  140 : Spectrally inverted signal with non-coherent sampling errors embedded 
     
       
         
           
             
               
                 
                   
                     
                       A 
                       1 
                     
                     ⁡ 
                     
                       ( 
                       
                         ± 
                         Δ 
                       
                       ) 
                     
                   
                   ⁢ 
                   
                     cos 
                     ⁡ 
                     
                       ( 
                       
                         
                           - 
                           
                             ω 
                             1 
                           
                         
                         ± 
                         
                           
                             m 
                             n 
                           
                           ⁢ 
                           
                             ω 
                             s 
                           
                         
                       
                       ) 
                     
                   
                 
               
               
                 
                   ( 
                   5 
                   ) 
                 
               
             
           
         
       
     
     Where m/n is the sampling error term representing the non-coherent representation of the input interfering signal. 
     Coherent Cancellation: 
     Identical output to equation (5) above except that the ratio of (m/n) approaches 0 and Δ approaches “1”. 
     Input to ADC  148 : 
     Represents the required signal plus interfering signal attenuated by a factor of K, where K represents the attenuation due to the interference canceller non coherent and coherent final inversion of the interfering signal: 
     
       
         
           
             
               
                 
                   
                     
                       A 
                       2 
                     
                     ⁢ 
                     
                       cos 
                       ⁡ 
                       
                         ( 
                         
                           
                             
                               ω 
                               2 
                             
                             ⁢ 
                             t 
                           
                           + 
                           
                             φ 
                             2 
                           
                         
                         ) 
                       
                     
                   
                   + 
                   
                     
                       
                         A 
                         1 
                       
                       K 
                     
                     ⁢ 
                     
                       cos 
                       ⁡ 
                       
                         ( 
                         
                           
                             
                               ω 
                               1 
                             
                             ⁢ 
                             t 
                           
                           + 
                           
                             φ 
                             1 
                           
                         
                         ) 
                       
                     
                   
                 
               
               
                 
                   ( 
                   6 
                   ) 
                 
               
             
           
         
       
     
     One skilled in the art will understand that even though various embodiments and advantages have been set forth in the foregoing description, the above disclosure is illustrative only, and changes may be made in detail, and yet remain within the broad principles of the invention. Moreover, the functions performed by various components described above may be implemented through circuitry or components other than those disclosed for the various embodiments described above. Moreover, the described functions of the various components may be combined to be performed by fewer elements or performed by more elements, depending upon design considerations for the device or system being implemented, as will appreciated by those skilled in the art. Therefore, the present invention is to be limited only by the appended claims.

Technology Category: h