Patent Document

RELATED APPLICATION 
       [0001]    The present invention claims priority of provisional patent application No. 61/261,481 filed Nov. 16, 2009, the contents of which are incorporated herein in their entirety. 
     
    
     BACKGROUND 
       [0002]    1. Technical Field 
         [0003]    The present teaching relates to method and system for analog circuits. More specifically, the present teaching relates to method and system for limiting amplifiers and systems incorporating the same. 
         [0004]    2. Discussion of Technical Background 
         [0005]    Modern high-performance communication systems increasingly rely on precision clocks for data conversion. Clocks used to drive analog-to-digital converters (ADC&#39;s) sample the analog input signal, where any imperfections in the sampling timing will result in imperfections in the output data. Such imperfections can be characterized by degradation of the ADC signal-to-noise ratio (SNR), where it is known that the SNR degradation due to clock imperfections is proportional to both clock jitter and the analog input frequency. It is commonly known that this SNR degradation is indistinguishable from SNR degradation caused by quantization noise or other noise present on the analog input signal. 
         [0006]    As digital processing capability continues to improve, it has become popular to digitize signals earlier and earlier in the signal chain, hence at ever higher frequencies. For example, both direct-IF and even direct-RF receivers have become common. In these applications, an ADC is actually under-sampled. Under-sampling takes advantage of the fact that as long as the sampling occurs at the Nyquist frequency of the bandwidth of interest or above, the modulated signal can be recovered. Therefore, an analog input signal can actually be sampled at a frequency much lower than its carrier frequency, and the sampled data still retains the modulated content of interest. 
         [0007]    However, under-sampling places stringent requirements on the clock used for the sampling. As noted above, the SNR degradation due to jitter is dependent on both clock jitter and analog input frequency. In an under-sampled ADC, the clock will still be sampling a signal with the high slew-rates of the analog carrier input. So, as the analog input frequency increases, often a significant limitation to the SNR performance is clock jitter. 
         [0008]    Both clock jitter and phase noise are measures of clock variation. Phase noise is defined as the power spectral density of the phase variation of a signal. Thus, it is a frequency domain quantity. Integration of the phase noise over a specified offset frequency bandwidth converts it to a time-domain quantity, or aperture jitter, which is considered to be the variations of the clock zero-crossings relative to the zero-crossings of an ideal clock. Although there are other jitter metrics that reflect period variations and adjacent cycle variations of a clock, the most descriptive measure of clock variation is phase noise, because no offset frequency specific information is lost. 
         [0009]    A complication in designing low phase noise clocking systems is the issue of frequency aliasing. In sampled data systems such as ADC&#39;s, aliasing of the clock spectrum will occur for frequency content beyond the sampling frequency. As noted by W. Egan in the paper “Modeling Phase Noise in Frequency Dividers,” IEEE Transactions on Ultrasonics, Ferroelectrics and Frequency Control, Volume 37, Issue 4, July 1990, pages 310-311, a similar issue occurs for frequency dividers used in synthesizer applications. In both situations, large bandwidths in a clock path can result in degraded phase noise due to noise folding into the baseband. For this reason, as is noted in the U.S. Pat. No. 7,345,528, “Method and Apparatus for Improved Clock Preamplifier with Low Jitter”, columns 2-3, it is a common practice to design high performance communications systems with a bandpass filter following a low-noise sine-wave signal. Such a bandpass filter aims at reducing the out of band noise to minimize degradation in overall phase noise caused by spectral folding. 
         [0010]    In most situations, the output of such a bandpass filter is followed by a limiting amplifier to “square up” the signal before it is used to sample data (in ADC applications) or clock a frequency divider (in frequency synthesizer applications). The “squaring-up” of the clock is done to minimize the region where sampler or divider noise adversely affects the output signal noise. As noted by C. Xu, et al. in “Analysis of Clock Buffer Phase Noise”, 9 th  International Conference on Electronics, Circuits and Systems, 2002, Volume 2, September 2002, pages 425-428, the output phase noise scaling factor for a clock amplifier is improved by ensuring fast, symmetrical rising and falling output signals. A similar observation can be made that conversion of voltage noise to timing jitter is proportional to the inverse of slew-rate. Therefore, having a high slew-rate clock signal is highly desirable. 
         [0011]    It is often assumed that the output slew rate of an amplifier stage is determined by the charging of parasitic capacitors. While this is the case for a limiting amplifier driven by a high slew-rate signal, it is not the case when the limiting amplifier is driven by a slow sine wave or other low slew-rate input. Thus, it is a challenge for a limiting amplifier to achieve low phase noise when it is presented with a low slew-rate input, especially because a limiting amplifier is usually designed to operate over a wide range of input slew-rates. 
         [0012]    A typical limiting amplifier includes a series of cascaded gain stages  120 ,  130 , . . . ,  140 ,  150  that take an input signal  110  and produce an output signal  160 , as shown in  FIG. 1  (Prior Art), which not only limit the signal amplitude, but also increase the output slew rate. A common configuration for a gain stage of a prior art limiting amplifier is a circuit  200  shown in  FIG. 2  (Prior Art). In this circuit  200 , there is a differential pair Q 0   215  and Q 1   225  as well as corresponding emitter followers  240  and  235 , respectively. In such an implementation, bipolar devices are chosen based on their superior 1/f noise characteristics. However, large MOSFET devices may also be used. When a differential signal applied to the base inputs of the differential pair circuit, or nodes  205  IN+ and  210  IN−, has a high slew-rate, the output slew-rate is limited by the charging of parasitic capacitances, such as ITAIL charging parasitic capacitors at the outputs of the differential pair circuit, or nodes  225  DP+ and  215  DP−. 
         [0013]    In this configuration, any increase in parasitic capacitance at critical nodes, e.g., nodes  225  DP+ and  215  DP−, will degrade the output slew rate. But in case of a low slew-rate input signal, the parasitic capacitances play less of a role in output slew-rate, which is primarily determined by the amplifier gain multiplied by the input slew-rate. 
         [0014]    An important observation is that any circuit bandwidth beyond that which improves or maintains the output slew-rate will degrade the output phase noise because of the increasing effect of noise folding, as described earlier. Given such observations, it is desirable to limit the bandwidth when a low slew-rate input is applied to reduce the degradation in phase noise caused by excess bandwidth. This may be achieved by adding a filter to a limiting amplifier stage so that broadband noise can be reduced and output phase noise can be improved without affecting the output slew-rate. In fact, an optimal bandwidth reduction may actually result in a slight reduction of output slew rate. This is because, up to a point determined by circuit characteristics and operating conditions, the output noise reduction may outweigh the slew-rate reduction and actually result in superior phase noise. 
         [0015]    U.S. Pat. No. 4,591,805, describes a method of implementing an adaptive bandwidth amplifier. An example adaptive bandwidth amplifier gain stage  300  is illustrated in  FIG. 3  (Prior Art). Similar to the circuit shown in  FIG. 2 , circuit  300  includes a pair of differential bipolar transistors Q 0   320  and Q 1   330 . The differential bipolar transistors receive differential input IN+  305  and IN−  310  and produce, via their followers output OUT+  370  and OUT−  375 . Otherwise, in circuit  300 , a capacitor  315  is introduced across the base-collector nodes of the bipolar transistor Q 0   320  and another capacitor  325  is introduced across the base-collector nodes of the bipolar transistor Q 1   330 . With this construction, circuit  300  takes advantage of the Miller multiplication property, which results in an increased effective capacitance proportional to the gain. In addition to allowing small capacitors  315  and  325  (or C CB ) to achieve a specific bandwidth reduction, this prior art circuit  300  also allows adaptive bandwidth control as the input signal swing varies due to the gain dependence of the effective input capacitance. 
         [0016]    One drawback of circuit  300  for filtering broadband noise is that, in addition to creating a dominant pole, this configuration also results in a zero at g m /C CB . Because of this zero, as frequency increases, the gain flattens until other poles take effect. Therefore, a single-pole filter at the same frequency as a dominant pole of circuit  300  is actually more effective for filtering broadband noise. Another observation is that inputs IN+  305  and IN−  310  connecting to the bases of transistors Q 0   320  and Q 1   330 , as shown in  FIG. 3 , likely are driven by a low impedance source, particularly in the context of a limiting amplifier as described earlier. In this case, the Miller multiplied effective capacitance at the base nodes of Q 0   320  and Q 1   330  likely will have less effect on circuit bandwidth than the effective capacitance at the collectors of Q 0   320  and Q 1   330 , since the collector nodes will usually be higher impedance. Therefore, an improved approach for effective reduction of bandwidth is needed. 
       SUMMARY 
       [0017]    The teachings disclosed herein related to methods and systems for improving limiting amplifier phase noise for low slew-rate input signals. 
         [0018]    In one example, a limiting amplifier circuit with improved phase noise comprises an input port, an output port, and one or more cascaded gain stages, having an input of a first gain stage connected to the input port, an output of a last gain stage connected to the output port, and an output of each gain stage connected to an input of an adjacent gain stage. In this example, each gain stage i, 1&lt;i&lt;n−1, is configured so that it is capable of selecting at least one lowpass filter corner frequency ω p     i   , and thereby reducing the phase noise of the gain stage through the broadband noise reduction for frequencies greater than ω p     i   , and ω p     i    is selected from a plurality of values associated with the gain stage to optimize the phase noise of the limiting amplifier by trading off reducing the broadband noise of the gain stage versus maintaining a sufficient output slew-rate of the gain stage. 
         [0019]    In another example, a limiting amplifier circuit with improved phase noise comprises an input port, an output port, and one or more cascaded gain stages, having an input of a first gain stage connected to the input port, an output of a last gain stage connected to the output port, and an output of each gain stage connected to an input of an adjacent gain stage. In this example, each gain stage i, 1&lt;i&lt;n−1, is configured so that it is capable of selecting at least one lowpass filter corner frequency ω p     i   , and thereby reducing the phase noise of the gain stage through the broadband noise reduction for frequencies greater than ω p     i   , and ω p     i    is selected from a plurality of values associated with the gain stage to optimize the phase noise of the limiting amplifier by trading off reducing the broadband noise of the gain stage versus maintaining a sufficient output slew-rate of the gain stage. Each gain stage comprises a differential input pair circuit having first and second transistors with their bases connected to differential positive and negative inputs, respectively, emitters connected to a first current source, and collectors coupled to a first power source via corresponding first and second resistors, an output circuit having third and fourth transistors with their emitters connected to second and third current sources, collectors connected to the first power source, and bases to the respective collectors of the first and second transistors, a first capacitor coupled in parallel to the first resistor, a second capacitor coupled in parallel to the second resistor, and a first set of switches to couple the first and second capacitors to the first and second resistors, respectively, when the bandwidth reduction sub-circuitry is enabled, and to decouple the first and second capacitors otherwise. 
         [0020]    In a different example, a limiting amplifier circuit with improved phase noise comprises an input port, an output port, and one or more cascaded gain stages, having an input of a first gain stage connected to the input port, an output of a last gain stage connected to the output port, and an output of each gain stage connected to an input of an adjacent gain stage. In this example, each gain stage i, 1&lt;i&lt;n−1, is configured so that it is capable of selecting at least one lowpass filter corner frequency ω p     i   , and thereby reducing the phase noise of the gain stage through the broadband noise reduction for frequencies greater than ω p     i   , and ω p     i    is selected from a plurality of values associated with the gain stage to optimize the phase noise of the limiting amplifier by trading off reducing the broadband noise of the gain stage versus maintaining a sufficient output slew-rate of the gain stage. Each gain stage comprises a differential input pair circuit having first and second transistors with their bases connected to differential positive and negative inputs, respectively, emitters connected to a first current source, and collectors coupled to a first power source via corresponding first and second resistors, an output circuit having third and fourth transistors with their emitters connected to second and third current sources, collectors connected to the first power source, and bases to the respective collectors of the first and second transistors, a first capacitor connecting the positive input and the second power source, a second capacitor connecting the negative input and the second power source, and a first set of switches to couple the first and second capacitors to the positive and negative inputs, respectively, when the bandwidth reduction sub-circuitry is enabled, and to decouple the first and second capacitors otherwise. 
         [0021]    In another different example, a limiting amplifier circuit with improved phase noise comprises an input port, an output port, and one or more cascaded gain stages, having an input of a first gain stage connected to the input port, an output of a last gain stage connected to the output port, and an output of each gain stage connected to an input of an adjacent gain stage, with each gain stage i, 1&lt;i&lt;n−1, is configured so that it is capable of selecting at least one lowpass filter corner frequency ω p     i   , and thereby reducing the phase noise of the gain stage through the broadband noise reduction for frequencies greater than ω p     i   , and ω p     i    is selected from a plurality of values associated with the gain stage to optimize the phase noise of the limiting amplifier by trading off reducing the broadband noise of the gain stage versus maintaining a sufficient output slew-rate of the gain stage. Each gain stage comprises a differential input pair circuit having first and second transistors with their bases connected to differential positive and negative inputs, respectively, emitters connected to a first current source, and collectors coupled to a first power source via corresponding first and second resistors, an output circuit having third and fourth transistors with their emitters connected to second and third current sources, collectors connected to the first power source, and bases to the respective collectors of the first and second transistors, a first capacitor connecting the base of the first transistor and the collector of the second transistor, a second capacitor connecting the base and collector of the first transistor, a third capacitor connecting the base of the second transistor and the collector of the first transistor, a fourth capacitor connecting the base and collector of the second transistor, and a first set of switches to couple the first, second, third and fourth capacitors to the appropriate base and collector junctions of the first and second transistors when the bandwidth reduction sub-circuitry is enabled, and to decouple them otherwise. 
         [0022]    In yet another different example, a device incorporating a limiting amplifier with improved phase noise comprises one or more circuitries configured for performing corresponding one or more functions, at least one limiting amplifier, each of which is coupled with at least one of the circuitries, taking an input and producing an output. The limiting amplifier comprises an input port, an output port, and one or more cascaded gain stages, having an input of a first gain stage connected to the input port, an output of a last gain stage connected to the output port, and an output of each gain stage connected to an input of an adjacent gain stage. Each gain stage i, 1&lt;i&lt;n−1, is configured so that it is capable of selecting at least one lowpass filter corner frequency ω p     i   , and thereby reducing the phase noise of the gain stage through the broadband noise reduction for frequencies greater than ω p     i   , and ω p     i    is selected from a plurality of values associated with the gain stage to optimize the phase noise of the limiting amplifier by trading off reducing the broadband noise of the gain stage versus maintaining a sufficient output slew-rate of the gain stage. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0023]    The inventions claimed and/or described herein are further described in terms of exemplary embodiments. These exemplary embodiments are described in detail with reference to the drawings. These embodiments are non-limiting exemplary embodiments, in which like reference numerals represent similar structures throughout the several views of the drawings, and wherein: 
           [0024]      FIG. 1  (Prior Art) shows a limiting amplifier with cascaded gain stages; 
           [0025]      FIG. 2  (Prior Art) shows a circuit of a gain stage of a limiting amplifier; 
           [0026]      FIG. 3  (Prior Art) shows a gain stage circuit in a limiting amplifier with filter capacitors to reduce bandwidth; 
           [0027]      FIGS. 4-6  depict exemplary gain stage circuits in a limiting amplifier with filter capacitors to provide improved phase noise, according to an embodiment of the present teaching; 
           [0028]      FIG. 7  depicts an exemplary gain stage circuit in a limiting amplifier with phase noise reduction sub-circuitry and bandwidth selection sub-circuitry, according to an embodiment of the present teaching; 
           [0029]      FIG. 8  depicts a limiting amplifier with cascaded gain stages each of which has the ability of bandwidth selection for phase noise reduction, according to an embodiment of the present teaching; and 
           [0030]      FIG. 9  is an exemplary device incorporating a limiting amplifier capable of bandwidth selection and phase noise reduction in each gain stage, according to an embodiment of the present teaching. 
       
    
    
     DETAILED DESCRIPTION 
       [0031]    Apparatus and method for a limiting amplifier which has the ability to reduce the bandwidth at each different gain stage to improve the phase noise performance of the limiting amplifier when it is presented with a low slew-rate input. In addition, the present teaching incorporates selectable noise filtering at each gain stage of the limiting amplifier so that a specific bandwidth can be selected for that gain stage suitable for reducing broadband noise and hence output phase noise when inputs to the limiting amplifier are known to have a low slew-rate input. 
         [0032]      FIGS. 4-6  show different exemplary implementations of a gain stage of a limiting amplifier capable of reducing bandwidth, according to an embodiment of the present teaching.  FIG. 4  depicts a circuit  400  that is an improved circuit over circuit  300  as shown in  FIG. 3 . The circuit  400  comprises a pair of differential bipolar transistors  415  and  420 , and a corresponding pair of followers  450 ,  455 , respectively. The differential bipolar transistors receive differential input IN+  405  and IN−  410  and produce, via their followers output OUT+  470  and OUT−  475 . The emitters of the differential bipolar transistor pair are tied together to a current source  425 . The emitters of followers  450  and  455  are connected to separate current sources  460  and  465 , respectively. The collector of transistor  415  is connected to a voltage VCC via a resistor  435  and the collector of transistor  420  is connected to the voltage VCC via a resistor  440 . 
         [0033]    In this improved circuit, two filter capacitors,  430  and  445  are introduced to reduce bandwidth. As illustrated, capacitor  430  connects between the voltage VCC and the collector of transistor  415 . In parallel, capacitor  445  connects between the voltage VCC and the collector of transistor  420 . The prior art circuit  300 , as shown in  FIG. 3 , has capacitors for reducing bandwidth across the collector and base terminals of each differential transistor. In circuit  400 , such filter capacitors are between the voltage VCC and the collector of the respective transistors. The capacitance of capacitors  430  and  445  is chosen to be C L . 
         [0034]      FIG. 5  depicts another exemplary gain stage circuit  500  of a limiting amplifier with an improved ability for bandwidth reduction, according to a different embodiment of the present teaching. As can be seen, circuit  500  is constructed in a similar manner as circuit  400  except that two bandwidth reducing filter capacitor,  525  and  530 , are now connecting between the base terminals of the respective differential bipolar transistors  515  and  520  and a voltage VEE. Specifically, capacitor  525  is coupled between the base terminal of transistor  515  and VEE and capacitor  530  is coupled between the base terminal of transistor  520  and VEE. Here, VEE can be the ground. Due to the fact that the filter capacitors are added to the base of the differential bipolar transistors, the capacitance of both  525  and  530  may likely need to be very high in order to achieve the same level of bandwidth reduction as what can be achieved using circuit  400 . In addition, the configuration shown in  FIG. 5  filters only input noise, while circuit  400  can filter both input noise and noise generated by the differential pair circuit. 
         [0035]      FIG. 6  depicts another gain stage circuit  600  of a limiting amplifier with an improved ability of bandwidth reduction, according to another embodiment of the present teaching. In circuit  600 , while other components (differential bipolar transistors and followers) are similarly connected, the filter capacitors used are different. Specifically, in addition to a capacitor C 2   630  coupled at the base-collector junction of transistor  615 , another capacitor C 0   635  is added connecting from the base of Q 0   615  to the collector of Q 1   620 . Symmetrically, in addition to a capacitor C 1   645  coupled at the base-collector junction of transistor  620 , another capacitor C 3   650  is added connecting from the base of Q 1   620  to the collector of Q 0   615 . 
         [0036]    The additional capacitors, C 0   635  and C 3   650 , effectively provide a Miller multiplication, but across a positive gain. When the capacitance of this additional capacitor ( 635  or  650 ) is chosen to be the same, say C CB , as that of capacitors ( 630  or  645 ) across the base-collector nodes of Q 0   615  and Q 1   620 , the combined Miller multiplications cancel each other leaving an approximate effective capacitance of 2×C CB  at each of the base nodes of Q 0   615  and Q 1   620 , and at each of their collector nodes. 
         [0037]    The circuit  600  as shown in  FIG. 6  achieves performance similar to that of the circuit  400  shown in  FIG. 4 , when the capacitance of each capacitor, or C CB , is chosen to be one-half of the value of C L  (the capacitance of filter capacitors  430  or  445 ) as used in  FIG. 4 . The bandwidth limiting performance that is achieved by circuit  600  is slightly superior to that of circuit  400  because circuit  600  also provides an effective capacitance on the bases of Q 0   615  and Q 1   620 . 
         [0038]    There is an additional consideration. A limiting amplifier usually needs to operate over a wide range of input slew-rates. When a high slew-rate input is provided, it may be needed to minimize the bandwidth reduction effect intended for the case of a low slew-rate input. In a BiCMOS process, MOSFET devices can be used to switch the filter capacitors in and out. In such applications, however, the MOSFET devices used need to be large to reduce parasitic on-resistance. It is often desired to minimize the number of switches in order to reduce the area needed and the circuit complexity. In light of such considerations, the topology depicted in  FIG. 4  may be more suitable when compared with that of circuit  600 . 
         [0039]      FIG. 7  depicts a gain stage circuit  700  of a limiting amplifier with a capability of switching filter capacitors on and off, according to an embodiment of the present teaching. The circuit  700  is constructed based on circuit  400  with additional switches that are used to switch the filter capacitors on and off in accordance with a control signal (FILTB) that dictates the filter bandwidth for that gain stage. As shown, circuit  700  comprises all the components as are in circuit  400  such as differential bipolar transistors  715  and  720 , their respective followers  765  and  780 , current sources  725 ,  792 , and  795 , as well as resistors  750  and  755  connecting the collectors of the transistors  715  and  720  to voltage VCC. Although there are filter capacitors  745  and  770 , which couple the respective collectors to voltage VCC, their connections to collectors of transistors  715  and  720  can be turned on and off via switches. 
         [0040]    As shown, devices M 0   730  and M 2   740  serve as a switch with respect to capacitor  745  and devices M 1   760  and M 3   775  serve as a switch with respect to capacitor  770 . Those devices ( 730 ,  740 ,  760 , and  775 ) are controlled by a signal filter bar (FILTB) via an inverter  735 . Whenever signal FILTB is high, it turns off devices M 0   730  and M 1   760 . At the same time, the output signal of the inverter  735  provides a low signal which turns on both M 2   740  and M 3   775 . Together, those switches effectively decouple the filter capacitors  745  and  770  from the transistors  715  and  720 . Conversely, whenever signal FILTB is low, it turns on devices M 0   730  and M 1   760  and turns off M 2   740  and M 3   775 , which together connect the filter capacitors  745  and  770  with transistors  715  and  720  so that they serve to reduce the bandwidth. 
         [0041]    While devices M 0   730  and M 1   760  add some parasitic capacitance to the collectors of Q 0   715  and Q 1   720 , even when switched off, this capacitance is relatively small, just C GD . A tradeoff can be made to determine the optimal sizing of M 0   730  and M 1   760  based on the particular application needs and circuit specifics. This is because making these devices very large minimizes on-resistance and making them very small minimizes off-state parasitic capacitance. Note that M 2   740  and M 3   775  are optional small devices and they may be added so the source nodes of M 0   730  and M 1   760  do not float and are forced to be VCC. 
         [0042]    As discussed herein, circuit  700  is capable of selectively reducing bandwidth based on the state of the controlling signal FILTB. For example, FILTB can be controlled to have a low state when the input slew-rate is below a certain threshold, and have a high state otherwise. In accordance with the present teaching, each gain stage of a limiting amplifier can incorporate such selectable noise filtering capability so that each gain stage of the limiting amplifier has the ability to select a specific bandwidth suitable for that gain stage for reducing broadband noise and hence output phase noise when inputs to the limiting amplifier are known to have a low slew-rate. 
         [0043]    In some embodiments, further improvement may be made by modifying circuit  700  to allow selection of more than one circuit bandwidth. This may be achieved by replicating (not shown) the sub-circuit comprising devices M 0   730 , M 1   760 , M 2   740 , M 3   775  and filter capacitors  745  and  770  so that each sub-circuit can be independently selected, thereby allowing configuration of different amplifier bandwidths based on the specific input slew rate. In addition, in a limiting amplifier with cascaded stages, different gain stages may need to be optimized to handle different ranges of low slew-rate input signals. Each stage may use the topology as shown in  FIG. 7 , with or without replicated sub-circuits, to select one or more bandwidths, but each stage is configured to select different bandwidth ranges and corresponding filtering capabilities. For instance, each stage may select different bandwidths and employ filter capacitors of different capacitance values determined, e.g., based on the expected input slew-rate and noise characteristics of that specific stage. 
         [0044]    The block diagram in  FIG. 8  shows an exemplary implementation of a limiting amplifier having multiple cascaded gain stages, each of which has the ability of bandwidth reduction based on a selected bandwidth, according to an embodiment of the present teaching. In this general form of a limiting amplifier, there are a plurality of cascaded gain stages,  820 ,  830 , . . . ,  840 , and  850 , which takes an input IN  810  and produces an output OUT  860 . In this configuration, each of the cascaded gain stages is implemented with a single-pole filter whose cutoff frequency may be selected from a range of values. For example, the first gain stage  820  has the ability of selecting one of K 1  cutoff frequencies, the second gain stage  830  has the ability of selecting one of K 2  cutoff frequencies, . . . , and the second to last stage can select one of K n−1  cutoff frequencies. 
         [0045]    In some embodiments, the selection of cutoff frequencies can be achieved by means of bandwidth selectors, shown as b 1  through b n−1  in  FIG. 8 . Since each individual gain stage has its own selector, each stage may therefore be customized to reduce the bandwidth based on some expected input slew-rate of that stage. It is worth mentioning that each consecutive stage may limit the bandwidth at a higher frequency, since each consecutive gain stage is to be presented with a higher slew-rate input than its predecessor. 
         [0046]    The difference between the exemplary block diagram of a limiting amplifier with multiple gain stages shown in  FIG. 8  as compared with a prior art limiting amplifier with multiple gain stages shown in  FIG. 1  is that the bandwidth at each gain stage can not be adjusted based on an input slew-rate. As a consequence, the prior art limiting amplifier usually has a much greater bandwidth than required or desired when presented with a low slew-rate input signal, while the limiting amplifier as shown in  FIG. 8  has the ability to optimize the performance of each gain stage by selecting appropriate bandwidth at each stage. 
         [0047]      FIG. 9  depicts a device  900  incorporating a limiting amplifier  910  capable of bandwidth selection and phase noise reduction in each gain stage, according to an embodiment of the present teaching. Device  900  includes one or more circuitries, circuit  1   950 , circuit  2   960 , . . . , circuit K  970 . In addition, device  900  also includes a limiting amplifier  910  constructed and functioning in accordance with what is disclosed herein. The limiting amplifier  910  may connect to or be used in any of the circuits  950 , . . . ,  970  (connection is not shown). The limiting amplifier  910  comprises a series of cascaded gain stages  920 ,  930 , . . . ,  940 , each of which has the ability of bandwidth reduction based on a selected bandwidth b 1 , b 2 , b n−1 , respectively. The limiting amplifier  910  provides improved performance as to phase noise in a manner as described in the present teaching disclosed herein. In many instances, the limiting amplifier  910  may use MOSFET rather than bipolar transistors. In such cases, the present teaching would be modified so the MOSFET gate terminal would be used instead of the bipolar base terminal, the MOSFET source in place of the bipolar emitter, and the MOSFET drain in place of the bipolar collector. In addition, although an exemplary limiting amplifier  910  is included in device  900  in  FIG. 9 , more than one such limiting amplifiers may be incorporated in a single device. It is generally understood that any appropriate technologies, whether currently existing or developed in the future, may be employed to implement the teachings disclosed herein. 
         [0048]    While the inventions have been described with reference to the certain illustrated embodiments, the words that have been used herein are words of description, rather than words of limitation. Changes may be made, within the purview of the appended claims, without departing from the scope and spirit of the invention in its aspects. Although the inventions have been described herein with reference to particular structures, acts, and materials, the invention is not to be limited to the particulars disclosed, but rather can be embodied in a wide variety of forms, some of which may be quite different from those of the disclosed embodiments, and extends to all equivalent structures, acts, and, materials, such as are within the scope of the appended claims.

Technology Category: h