Patent Document

CROSS-REFERENCE TO RELATED APPLICATIONS 
       [0001]    This application is based upon and claims the benefit of priority from the prior Japanese Patent Applications No. 2006-251262 filed on Sep. 15, 2006, the entire contents of which are incorporated herein by reference. 
       BACKGROUND OF THE INVENTION 
       [0002]    1. Field of the Invention 
         [0003]    The present invention relates to a filter circuit, for example, a band limiting filter circuit provided in a posterior stage of a power amplifier in a transmission unit of a radio communication apparatus. 
         [0004]    2. Related Art 
         [0005]    Conventionally, a filter circuit is constructed by cascade-connecting resonators (resonance circuits) whose conductor part is made of, for example, superconductor. Superconductor has a limit value in a current per unit area that can flow in a superconducting state. An equivalent circuit of a resonator is made up of an inductor and a capacitor and is also provided with a resistor when an effect of loss is considered. A resonance frequency of a resonator when there is no resistor is given by the following expression. “L” and “C” denote inductance and capacitance of the resonator respectively. 
         [0000]        f 0=( L×C ) 1/2    
         [0006]    In this filter circuit, a pass frequency range and an amount of attenuation of a filtering region can be determined by appropriately determining an inter-resonator coupling coefficient which indicates the amount of coupling between resonators and a value of external Q which indicates an amount of excitation for resonators on the input side and the output side. 
         [0007]    In such a filter circuit made up of cascade-connected resonators, a current flows through each resonator, that is, a current of all frequency components flows through each resonator, and therefore power handling capability of each resonator needs to be increased. This results in a problem of increasing the size of the circuit. The specification of U.S. Pat. No. 6,633,208 describes that a highest current passes through a first resonator in a cascade connection type filter circuit, and a multi-wavelength structure is adopted for the first resonator (i.e. line length is set to half wavelength×n (n is an integer equal to or greater than 2)) to disperse the current in the resonator. 
         [0008]    On the other hand, as another filter circuit, there is a parallel connection type filter circuit made up of resonators connected in parallel whose conductor part is made of superconductor (see, for example, JP-A 2001-345601 (Kokai) and JP-A 2004-96399 (Kokai)). This parallel connection type filter circuit combines signals which pass through resonators having neighboring resonance frequency so as to have phases opposite to each other and thereby realizes a filter characteristic. This filter circuit distributes input power to the respective resonators, and can thereby increase the power handling capability as a whole, yet reduce power handling capability of each resonator compared to a cascade connection type filter circuit and thereby also reduce the circuit scale. However, there is a demand for a further reduction in the circuit scale. 
       SUMMARY OF THE INVENTION 
       [0009]    According to an aspect of the present invention, there is provided with a filter circuit comprising: 
         [0010]    an input terminal configured to input an input signal; 
         [0011]    first to ith blocks which have first to ith resonators as transmission lines having first to ith resonance frequencies (first resonance frequency&lt;second resonance frequency&lt; . . . &lt;ith resonance frequency); 
         [0012]    a power divider configured to distribute the input signal to the first to ith blocks; 
         [0013]    a power combiner configured to combine signals which have passed through the first to ith blocks to obtain a combined signal; and 
         [0014]    an output terminal configured to output the combined signal, 
         [0015]    wherein a jth block (j is an integer between 1 and i−1) includes a phase adjustment unit which provides a signal of the jth block with a phase difference within a range of {(180±30)+(360×n)} degrees (n is an integer equal to or greater than 0) from a signal of a (j+1)th block, and 
         [0016]    a resonator having a large amount of group delay has a greater line width than a resonator having a small amount of group delay. 
         [0017]    According to an aspect of the present invention, there is provided with a filter circuit comprising: 
         [0018]    an input terminal configured to input an input signal; 
         [0019]    first to ith blocks which have first to ith resonators as transmission lines having first to ith resonance frequencies (first resonance frequency&lt;second resonance frequency&lt; . . . &lt;ith frequency); 
         [0020]    a power divider configured to distribute the input signal to the first to ith blocks; 
         [0021]    a power combiner configured to combine signals which have passed through the first to ith blocks to obtain a combined signal; and 
         [0022]    an output terminal configured to output the combined signal, 
         [0023]    wherein a jth block (j is an integer between 1 and i−1) includes a phase adjustment unit which provides a signal of the jth block with a phase difference within a range of {(180±30)+(360×n)} degrees (n is an integer equal to or greater than 0) from a signal of a (j+1)th block, and 
         [0024]    when the line length of the resonator having a large amount of group delay is Nd 1  times a half wavelength at the resonance frequency and the line length of the resonator having a small amount of group delay is Nd 2  times a half wavelength at the resonance frequency, Nd 1  and Nd 2  have a relationship of Nd 1 &gt;Nd 2  (Nd 1  is an integer equal to or greater than 2, Nd 2  is an integer equal to or greater than 1). 
         [0025]    According to an aspect of the present invention, there is provided with a filter circuit comprising: 
         [0026]    an input terminal configured to input an input signal; 
         [0027]    first to ith blocks which have first to ith resonators as transmission lines having first to ith resonance frequencies (first resonance frequency&lt;second resonance frequency&lt; . . . &lt;ith resonance frequency); 
         [0028]    a power divider configured to distribute the input signal to the first to ith blocks; 
         [0029]    a power combiner configured to combine signals which have passed through the first to ith blocks to obtain a combined signal; and 
         [0030]    an output terminal configured to output the combined signal, 
         [0031]    wherein a jth block (j is an integer between 1 and i−1) includes a phase adjustment unit which provides a signal of the jth block with a phase difference within a range of {(180±30)+(360×n)} degrees (n is an integer equal to or greater than 0) from a signal of a (j+1)th block, 
         [0032]    a resonator having a large amount of group delay has a greater line width than a resonator having a small amount of group delay, and 
         [0033]    when the line length of the resonator having a large amount of group delay is Nd 1  times a half wavelength at the resonance frequency and the line length of the resonator having a small amount of group delay is Nd 2  times a half wavelength at the resonance frequency, Nd 1  and Nd 2  have a relationship of Nd 1 &gt;Nd 2  (Nd 1  is an integer equal to or greater than 2, Nd 2  is an integer equal to or greater than 1). 
     
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0034]      FIG. 1  is a plan layout diagram showing a first embodiment of a filter circuit of the present invention; 
           [0035]      FIG. 2  is an equivalent circuit diagram of the embodiment shown in  FIG. 1 ; 
           [0036]      FIG. 3  shows a frequency response characteristic of the embodiment shown in  FIG. 1 ; 
           [0037]      FIG. 4  shows a group delay characteristic of the embodiment shown in  FIG. 1 ; 
           [0038]      FIG. 5  is a configuration diagram of a filter circuit illustrating the principle of the present invention; 
           [0039]      FIG. 6  shows a frequency response characteristic when coupling M 2  of the circuit shown in  FIG. 5  is negative; 
           [0040]      FIG. 7  shows a frequency response characteristic when coupling M 2  of the circuit shown in  FIG. 5  is positive; 
           [0041]      FIG. 8  shows a general cascade connection type filter circuit; 
           [0042]      FIG. 9  shows a current distribution of each resonator of the filter circuit in  FIG. 8 ; 
           [0043]      FIG. 10  shows a general parallel connection type filter circuit; 
           [0044]      FIG. 11  shows a current distribution of each resonator of the filter circuit in  FIG. 10 ; 
           [0045]      FIG. 12  is a plan layout diagram showing a specific numerical value example of each element shown in  FIG. 1 ; 
           [0046]      FIG. 13  is a plan layout diagram showing a second embodiment of the filter circuit of the present invention; 
           [0047]      FIG. 14  is a plan layout diagram showing a first modification example of the first embodiment; 
           [0048]      FIG. 15  is a plan layout diagram showing a second modification example of the first embodiment; 
           [0049]      FIG. 16  is a plan layout diagram showing an example combining the first and second embodiments; 
           [0050]      FIG. 17  is a plan layout diagram showing a third modification example of the first embodiment; 
           [0051]      FIG. 18  is a plan layout diagram showing a fourth modification example of the first embodiment; and 
           [0052]      FIG. 19  is a configuration diagram schematically showing an example of a radio communication apparatus. 
       
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       [0053]      FIG. 1  is a plan layout diagram showing a first embodiment of the filter circuit according to the present invention. 
         [0054]      FIG. 1  shows a microstrip line type filter circuit. A conductor strip is formed in a pattern as shown in the figure on the surface of a dielectric substrate  110  (e.g., sapphire substrate, MGO substrate) and a ground conductor is formed over the entire back surface of the dielectric substrate  110 . The conductor and the ground conductor (conductor part of the microstrip line type filter circuit) are made of a material having a limit value in a current per unit area that can flow in a superconducting state, for example, superconductor. This filter circuit is incorporated in, for example, a freezer. An example of the microstrip line type filter circuit is shown here, but it is also possible to apply the present invention to other type filter circuits such as a coplanar line type. 
         [0055]    The power of a signal inputted from an input line  101  is distributed to a first signal and a second signal by a power distributor  103 . The first signal is transmitted to resonators  105  and  107  configured as transmission lines (microstrip lines) via a line  121   a . The second signal is transmitted to resonators  106  and  108  configured as transmission lines (microstrip lines) via a line  121   b . The joint between the input line  101  and the lines  121   a  and  121   b  corresponds to the power distributor  103 . The resonators  105 ,  106 ,  107  and  108  have resonance frequencies of f 1 , f 2 , f 3  and f 4 . Suppose these resonance frequencies have a relationship of f 1 &lt;f 2 &lt;f 3 &lt;f 4 . That is, the resonators  105 ,  106 ,  107  and  108  resonate at resonance frequencies different from each other. External Q of the resonators  105  and  108  at both ends of the filter band (pass band) (suppose the external Q is the same on the input side and on the output side of the resonator here for simplicity of explanation, but the present invention also naturally includes a case where they are different) is set to be greater than that of the resonators  106  and  107  on the center side (the amount of group delay of the resonators  105  and  108  is greater than that of the resonators  106  and  107 ), and for this reason, the line widths of the resonators  105  and  108  are set to be greater than those of the resonators  106  and  107  to increase the power handling capability of the resonators  105  and  108 . The resonance frequency of a resonator can be measured by placing a probe for detecting radio wave close to the upper part of the resonator and measuring the return loss characteristic of a network analyzer. This makes it possible to arrange a resonator using a wide line to an end of the filter band. The amount of group delay of resonators can also be measured through measurement using a network analyzer likewise. 
         [0056]    The signal which has passed through the resonators  106  and  108  having resonance frequencies f 2  and f 4  is given to a power combiner  104  via a line  131 . The signal which has passed through the resonators  105  and  107  having resonance frequencies f 1  and f 3  is given to the power combiner  104  via a delay circuit (line)  109  which has an electric length of approximately 180 degrees at a center frequency of the filter circuit. This delay circuit  109  realizes a phase difference of 180 degrees at a point of combination between the signal which has passed through the resonators  105  and  107  having resonance frequencies f 1  and f 3  and the signal which has passed through the resonators  106  and  108  having resonance frequencies f 2  and f 4 . That is, the delay circuit  109  realizes a phase difference of 180 degrees (opposite phases) between the signals which have passed through the resonators of neighboring resonance frequencies. As will be described later, the neighboring signals may have substantially opposite phases, if not completely opposite phases, that is, a phase difference within a range of (180±30)+360×n degrees (n is an integer equal to or greater than 0). The amount of delay by the delay circuit  109  can be determined by adjusting the arrangement relationship between the resonators  105 ,  107  and delay circuit  109  (for example, length of the parts parallel to each other or distance from each other). 
         [0057]    The power combiner  104  combines power of the signals given from the resonators  105  to  108 , acquires a combined signal and outputs the combined signal from an output line  102 . The joint between the output line  102  and lines  131 ,  109  corresponds to the power combiner  104 . 
         [0058]    Impedance matching when performing signal distribution at the power distributor  103  and signal combination at the power combiner  104  can be realized by making up a matching circuit using an impedance conversion circuit with a changed line width and elements of L and C. That is, impedance matching is realized in the case of distribution by adjusting the width of the input line  101  and the widths of the two lines  121   a  and  121   b  which branch from the input line  101 . On the other hand, impedance matching is realized in the case of combination by adjusting the width of the output line  102  and the widths of the two lines  109  and  131  leading to the output line  102 . 
         [0059]    An equivalent circuit of the filter circuit in  FIG. 1  is shown in  FIG. 2 . The elements in  FIG. 2  corresponding to the elements shown in  FIG. 1  are assigned the same reference numerals. 
         [0060]    An input terminal  11  corresponds to the part of the input line  101  which combines the lines  121   a  and  121   b  in  FIG. 1 . An output terminal  12  corresponds to the part of the output line  102  which combines the lines  131  and  109  in  FIG. 1 . 
         [0061]    A power divider  103  is combined with resonators  105  to  108  and the resonators  105  to  108  are cascade-connected with phase adjustment units  109 ( 1 ) to  109 ( 4 ). 
         [0062]    The cascade-connected resonator  105  and phase adjustment unit  109 ( 1 ) are referred to as a block BL( 1 ). Likewise, the cascade-connected resonator  106  and phase adjustment unit  109 ( 2 ) are referred to as a block BL( 2 ). The cascade-connected resonator  107  and the phase adjustment unit  109 ( 3 ) are referred to as a block BL( 3 ). The cascade-connected resonator  108  and the phase adjustment unit  109 ( 4 ) are referred to as a block BL( 4 ). 
         [0063]    The phase adjustment unit  109 ( 1 ) is set so as to cause the signal passing through the block BL( 1 ) to have a phase substantially opposite to the phase of the signal passing through the BL( 2 ). The phase adjustment unit  109 ( 2 ) is set so as to cause the signal passing through the block BL( 2 ) to have a phase substantially opposite to the phase of the signal passing through the BL( 3 ). The phase adjustment unit  109 ( 3 ) is set so as to cause the signal passing through the block BL( 3 ) to have a phase substantially opposite to the phase of the signal passing through the BL( 4 ). The configuration of  FIG. 1  is, for example, equivalent to that in the case where the phase adjustment units  109 ( 2 ) and  109 ( 4 ) are set to 0 degrees and the phase adjustment units  109 ( 1 ) and  109 ( 3 ) are set to −(180±30) degrees. The phase adjustment units  109 ( 1 ) and  109 ( 3 ) correspond to the delay circuit  109  of  FIG. 1 . 
         [0064]    In the filter circuit shown in  FIG. 1  and  FIG. 2 , the aspect that signals passing through resonators of neighboring resonance frequencies are provided with a phase difference between substantially opposite phases and the aspect that the line widths of the resonators  105  and  108  are set to be greater than the line widths of the resonators  106  and  107  will be explained in detail respectively. 
         [0065]    First, the aspect that signals passing through resonators of neighboring resonance frequencies are provided with a phase difference between substantially opposite phases will be explained. 
         [0066]      FIG. 5  shows an example of a filter circuit which includes two general resonators. This filter circuit is provided with an input terminal  301 , a power divider  303 , two resonators (resonance circuits)  305  and  306 , a power combiner  304  and an output terminal  302 . The resonator  305  has a resonance frequency f 1  and the resonator  306  has a resonance frequency f 2 . Coupling M 2  denotes coupling between the resonator  306  and the power combiner  304 , m 1 ( 1 ) denotes coupling between the resonator  305  and the power divider  303 , m 1 ( 2 ) denotes coupling between the resonator  305  and the power combiner  304  and m 2  denotes coupling between the resonator  306  and the power divider  303 . Though inductive coupling is shown here, coupling may be any one or both of capacitative coupling and inductive coupling. 
         [0067]      FIG. 6  shows a frequency response of the filter circuit in  FIG. 5  when it is assumed that coupling M 2  is opposite-phase coupling (the phase is reversed by 180 degrees) and m 1 ( 1 ), m 1 ( 2 ) and m 2  are in-phase coupling (the phase does not change). Reference numeral  205   a  denotes a frequency response of the resonator  305 ,  205   b  denotes a frequency response of the resonator  306 ,  204  denotes a frequency response (combined signal) of the output terminal  302 . The frequency response  204  is a frequency response when the output signals of the two resonators  305  and  306  are combined as opposite-phase coupling, which is expressed as the sum of the single frequency responses  205   a  and  205   b  of the two resonators  305  and  306 . In this way, a desired frequency response (combined signal) can be obtained by combining the signals which have passed through the two resonators  305  and  306  so as to have phases opposite to each other. An amount of ripple  207  between the resonance frequencies f 1  and f 2  seen in the frequency response  204  can be adjusted to a desired value by setting the interval between the resonance frequencies f 1  and f 2 , coupling m 1 ( 1 ), m 1 ( 2 ), m 2  and M 2  of the respective resonators  305  and  306  to appropriate values. Furthermore, when coupling m 1 ( 1 ), m 1 ( 2 ) and m 2  are assumed to be opposite-phase coupling, making coupling M 2  in-phase coupling causes the signals which have passed through the resonators  305  and  306  to be combined so as to have phases opposite to each other making it possible to realize a combination of sum likewise. 
         [0068]      FIG. 7  shows a frequency response when coupling m 1 ( 1 ), m 1 ( 2 ) and m 2  are assumed to be in-phase coupling and coupling M 2  is also assumed to be in-phase coupling. 
         [0069]    Reference numeral  205   a  denotes a frequency response of the resonator  305 ,  205   b  denotes a frequency response of the resonator  306 ,  206  denotes a frequency response (combined signal) of the output terminal  302 . The frequency response  206  is a frequency response when the output signals of the two resonators  305  and  306  are combined so as to have the same phase, which is expressed as a difference between single frequency responses  205   a  and  205   b  of the two resonators  305  and  306 . It is understandable that signal intensity in the vicinity of the center frequency in a target band decreases and it is no longer possible to obtain a desired signal. Thus, a combination of difference results because the phases of signals before and after the respective resonance frequencies of the resonators  305  and  306  are inverted. Even when all coupling m 1 ( 1 ), m 1 ( 2 ), m 2  and M 2  are assumed to be opposite-phase coupling, a combination of difference results likewise. 
         [0070]    In the case of  FIG. 6 , since the two signals which have passed through the resonators  305  and  306  have phases opposite to each other before being combined, the phase inversion produced in the resonators  305  and  306  is canceled out and a desired signal can be obtained. As described above, when the two signals to be combined have phases substantially opposite to each other if not completely opposite phases, that is, a phase difference within a range of (180±30)+360×n degrees (n is an integer equal to or greater than 0), it is possible to obtain a desired signal. 
         [0071]    Based on the above described principle, the filter circuit shown in  FIG. 1  is provided with the delay circuit  109  to obtain a desired output signal so that the signals that have passed through the resonators having neighboring resonance frequencies have phases substantially opposite to each other. 
         [0072]    Next, the aspect that the line widths of the resonators  105  and  108  in the filter circuit in  FIG. 1  are set to be greater than the resonators  106  and  107  will be explained. 
         [0073]      FIG. 3  and  FIG. 4  show frequency characteristics of the filter circuit in  FIG. 1 .  FIG. 3  shows a graph  201  indicating a transmission characteristic (S 21  characteristic) and a graph  202  indicating a return loss characteristic (S 11  characteristic) and  FIG. 4  shows a graph  203  indicating a group delay characteristic. When a combination is performed as the filter characteristic as in  FIG. 2 , the resonance frequency of each resonator does not match the peak position of the return loss characteristic  202  in the strict sense of the word. This is because resonance frequencies are subject to perturbation under the influences of other resonators as a result of the combination of waveforms. However, their order never changes. 
         [0074]    To realize a steep skirt characteristic, as described above, the filter circuit in  FIG. 1  has greater external Q at both ends of the filter band (suppose the external Q is the same on the input side and on the output side of the resonator here for simplicity of explanation, but the present invention also naturally includes a case where they are different) than the external Q of other resonators. That is, the total of the coupling amount of the resonators with the circuit placed on the input side of the resonator at both ends of the filter band (the coupling amount is defined as the reciprocal of the external Q) and the coupling amount of the resonators with the circuit placed on the output side is smaller than the total of the coupling amount of the other resonators with the circuit placed on the input side of the other resonator and the coupling amount of the other resonators with the circuit placed on the output side. In this way, a higher current is obtained from the resonators at both ends of the filter band (see parts  201   a  and  201   b  in the graph  201  in  FIG. 3 ). That is, when the external Q of the resonator at both ends of the filter band is increased (when the coupling amount is decreased), the amount of group delay at both ends of the filter band increases as shown in  FIG. 4  and the value of current that can be extracted also increases in proportion thereto. More specifically, the greater the amount of group delay, the longer the signal stays in the resonator, and therefore the superimposition of waves produces a high current value. 
         [0075]    In this way, as a result of the increase in the amount of group delay of the resonators  105  and  108 , a high current stays in the resonators  105  and  108 , and therefore the resonators  105  and  108  are required to have greater power handling capability than the other resonators  106  and  107 . To put it the other way around, the resonators  106  and  107  are required to have not so large power handling capability as the resonators  105  and  108 . That is, it is not necessary to increase power handling capability of all the resonators and it is possible to obtain sufficient power handling capability for the filter circuit by increasing power handling capability of only resonators having a large amount of group delay. Focusing on this point, the inventor has implemented a filter circuit with the smallest possible circuit area while maintaining high power handling capability by increasing only the line widths of the resonators  105  and  108  having a large amount of group delay more than the line widths of the other resonators  106  and  107 . That is, a filter circuit with a small circuit area having a steep skirt characteristic has been realized. 
         [0076]    Hereinafter, the process through which the inventor has come up with the present invention will be explained in detail. 
         [0077]      FIG. 8  shows the configuration of a general cascade connection type filter circuit. In this filter circuit, six resonators  401  to  406  are cascade-connected. The conductor parts of resonators  401  to  406  are made of superconductor.  FIG. 9  shows current values of the respective resonators  401  to  406  in this filter circuit. The current values of the respective resonators  401  to  406  are shown in graphs G 401  to G 406 . The graph in  FIG. 9  is obtained through a simulation whereby a signal is inputted to the filter circuit while sequentially changing the frequency of the input signal within the frequency range on the horizontal axis in the figure and the current value of each resonator at a time of each frequency is measured. 
         [0078]    As is understandable from  FIG. 9 , a current in a whole frequency band passes through the respective resonators  401  to  406 , and therefore a high current (integral value in the graph) flows through the resonators  401  to  406 . The graph G 403  shows that the current value of the third resonator  403  becomes a maximum. In order for a high current to flow through the resonators, it is possible to effectively decrease the peak current value by distributing the current over a wider range using a large resonator. However, using a large resonator increases the size of the filter circuit. 
         [0079]      FIG. 10  shows the configuration of a general parallel connection type filter circuit. In this filter circuit, six resonators  411  to  416  are connected in parallel. The conductor parts of the resonators  411  to  416  are made of superconductor. The resonators  411  to  416  have the same power handling capability. The resonators  415  and  416  correspond to both ends of the filter band.  FIG. 11  shows current values of the respective resonators  401  to  406  of this filter circuit. Current values of the respective resonators  411  to  416  are shown in graphs G 411  to G 416 . The graph in  FIG. 11  is obtained through a simulation similar to that in  FIG. 9 . 
         [0080]    Since an input signal is distributed to the resonators  411  to  416 , a current (integral value in the graph) which flows through one resonator is smaller than that of the resonator in the cascade connection type filter circuit. Therefore, the power handling capability of each resonator can be made smaller than that of the filter circuit in  FIG. 8 , and it is thereby possible to reduce the circuit area more in the parallel connection type filter circuit than the cascade connection type filter circuit. 
         [0081]    Here, as is understandable from  FIG. 11 , the current values (G 415 , G 416 ) of the resonators  415  and  416  at both ends of the filter band in the parallel connection type filter circuit are greater than those of the other resonators  411  to  414 . Furthermore, in the parallel connection type filter circuit using superconductor, it is possible to use resonators having different power handling capabilities according to the current valued of the respective resonators. Focusing on this point, the inventor has realized both high power handling capability and downsizing of the filter circuit by increasing the power handling capability using a line of a greater line width for only resonators through which a high current flows. 
         [0082]    Here, specific numerical examples of the layout shown in  FIG. 1  are shown in  FIG. 12 . 
         [0083]    A dielectric constant ∈r of the dielectric substrate  110  is 24. The line length of the resonator  105  is 20.26 mm and the width is 0.8 mm. The line length of the resonator  106  is 20.18 mm and the width is 0.2 mm. The line length of the resonator  107  is 20.10 mm and the width is 0.2 mm. The line length of the resonator  108  is 20.02 mm and the width is 0.8 mm. Therefore, the widths of the resonators  105  and  108  are 4 times those of the resonators  106  and  107 . The line length of the delay circuit  109  is 40 mm. The line length of the line  131  is 20 mm. 
         [0084]      FIG. 13  shows a second embodiment of the filter circuit according to the present invention. 
         [0085]    This filter circuit is equipped with resonators  105   a ,  106   a ,  107   a  and  108   a  having resonance frequencies f 1 , f 2 , f 3  and f 4 . These frequencies have a relationship of f 1 &lt;f 2 &lt;f 3 &lt;f 4 . The line lengths of the resonators  105   a  and  108   a  having f 1  and f 4  at the ends of the filter band are set to Nd 1  times the half wavelength and the line lengths of the resonators  106   a  and  107   a  having f 2  and f 3  at the center side of the filter band are set to Nd 2  times the half wavelength. Here, Nd 1 &gt;Nd 2  (Nd 1  is an integer equal to or greater than 2, Nd 2  is an integer equal to or greater than 1).  FIG. 13  shows an example with Nd 1 =2, Nd 2 =1. Setting the line lengths of the resonators  105   a  and  108   a  to twice the half wavelength makes it possible to set power handling capability twice that in the case where the line lengths are set to the half wavelength. In the first embodiment, power handling capability has been improved by increasing the line width, but this embodiment improves power handling capability by increasing the line length. 
         [0086]      FIG. 14  shows a first modification example of the filter circuit according to the first embodiment. 
         [0087]    This filter circuit uses resonators  105   b ,  106   b  and  107   b  having resonance frequencies f 1 , f 2  and f 3 . These resonance frequencies have a relationship of f 1 &lt;f 2 &lt;f 3 . The line widths of the resonators  105   b  and  107   b  located at both ends of the filter band having a large amount of group delay are set to be greater than the line width of the resonator  106   b  having a smaller amount of group delay and the resonators  105   b  and  107   b  are concentrated on one location. This facilitates the layout design of the filter circuit. 
         [0088]      FIG. 15  shows a second modification example of the filter circuit of the first embodiment. 
         [0089]    This filter circuit uses resonators  105   c ,  106   c ,  107   c ,  108   c ,  111   c  and  112   c  having resonance frequencies f 1 , f 2 , f 3 , f 4 , f 5  and f 6 . These resonance frequencies have a relationship of f 1 &lt;f 2 &lt;f 3 &lt;f 4 &lt;f 5 &lt;f 6 . The resonators  105   c  and  112   c  located at both ends of the filter band having a large amount of group delay are assumed to have a first line width, the resonators  107   c  and  108   c  located at the center side of the filter band having a small amount of group delay are assumed to have a second line width which is smaller than the first line width and the resonators  106   c  and  111   c  having a medium amount of group delay are assumed to have a third line width which is smaller than the first line width and greater than the second line width. 
         [0090]    Incidentally, in the first embodiment (see  FIG. 1 ), an arrangement in which the resonators  106  and  108  partially face the lines  121   b  and  131  in parallel to each other is adopted in order to couple the resonators  106  and  108  with the lines  121   b  and  131 . The same applies to the relationship between the resonators  105  and  107 , and lines  121   a  and  109 . In contrast, this modification example adopts an arrangement in which the resonators  108   c ,  111   c  and  112   c  face the lines  141   b  and  151  at one end. The same applies to the arrangement between the resonators  105   c ,  106   c ,  107   c  and the lines  141   a  and  209 . 
         [0091]      FIG. 16  shows an example of combination between the first embodiment and the second embodiment. This shows an example of the filter circuit when both the line width and line length are changed. 
         [0092]    This filter circuit uses resonators  105   d ,  106   d ,  107   d  and  108   d  having resonance frequencies f 1 , f 2 , f 3  and f 4 . These resonance frequencies have a relationship of f 1 &lt;f 2 &lt;f 3 &lt;f 4 . The line widths of the resonators  105   d  and  108   d  located at both ends of the filter band having a large amount of group delay are set to be greater than those of the resonators  106   d  and  107   d  and the line lengths of the resonators  105   d  and  108   d  are set to twice the half wavelength. The line lengths of the resonators  106   d  and  107   d  are half wavelengths. 
         [0093]      FIG. 17  shows a third modification example of the filter circuit of the first embodiment. 
         [0094]    The line widths of resonators  105   e  and  108   e  located on both sides of the filter band having a large amount of group delay are set to be greater than those of the first embodiment. In this way, a filter circuit having higher power handling capability is realized. The line widths of resonators  106   e  and  107   e  located at the center side of the filter band are the same as those of the first embodiment. Furthermore, a delay circuit  309  is interposed between an input line  101  and the resonators  105   e  and  107   e  in this modification example. In this way, a delay circuit may be arranged on any one of the input side and the output side of the resonator. 
         [0095]      FIG. 18  shows a fourth modification example of the filter circuit of the first embodiment. 
         [0096]    Resonators  105   g  and  108   g  located on both sides of the filter band having a large amount of group delay correspond to a strip conductor in a microstrip line having a length of half wavelength which is made wider from both sides toward the center and have a substantially circular planar shape here. The resonance mode includes TM011 mode or TM010 mode. Current concentrates more on parts which are closer to the center of the half wavelength. In this example, current concentrates most on the parts indicated by L1 and L2. Therefore, by increasing the line width for parts which are closer to the center of the half wavelength, that is, by changing the line width according to the degree of concentration of current, it is possible to realize high power handling capability and reduce the area occupied by the resonator. In the case of a resonator having a multi-wavelength structure (half wavelength×n (n is an integer equal to or greater than 2), since current is more concentrated on parts closer to the center of each half wavelength, it is possible to realize high power handling capability and reduce the area occupied by the resonator by widening the line width from both ends of the length of half wavelength toward the center. 
         [0097]      FIG. 19  schematically shows the configuration of a radio communication apparatus as an embodiment of the present invention. More specifically, the configuration of a transmission unit of a radio communication apparatus is schematically shown. 
         [0098]    Data  500  to be transmitted is inputted to a signal processing circuit  501 , subjected to processing such as a digital/analog conversion, coding and modulation and a transmission signal of a baseband or an intermediate frequency (IF) band is generated. 
         [0099]    The transmission signal from the signal processing circuit  501  is inputted to a frequency converter (mixer)  502  and multiplied by a local signal from a local signal generator  503  and thereby converted to a signal of a radio frequency (RF) band, that is, up-converted. 
         [0100]    The RF signal outputted from the mixer  502  is amplified by power an amplifier  504  and then inputted to a band limiting filter (transmission filter)  505 . As the band limiting filter  505 , the filter circuit explained so far can be used. The signal whose band is limited by this band limiting filter  505  and whose unnecessary frequency component has been removed is supplied to an antenna and is radiated out into space as a radio wave.

Technology Category: 5