Patent Document

BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The invention relates to analog to digital converters generally and to the so-called delta-sigma modulators more specifically. 
     2. Description of the Prior Art 
     Modern information transmission systems are often based on the conversion of analog input signals to digital signals for transmission over a discrete channel. Both the analog to digital and digital to analog subsequent reconstruction are subject to errors because the continuum of possible input values must be represented by discrete sets of values in the transmission channel. This error is commonly known as quantizing noise and constitutes one of the major sources of inaccuracies in such systems. 
     In a technical paper (herein called paper No. 1) entitled &#34;Reduction of Quantizing Noise by Use of Feedback,&#34;  H. A. Spang III and P. M. Schultheiss analysed the quantizing noise problem and suggested the use of quantizer feedback as a means of ameliorating the inaccuracies produced by this quantizing noise. That paper appeared in the IRE Transactions on Communication Systems, Vol. CS-10, pp. 373-380 (December 1962). Therein, a general case delta-sigma modulator having a multi-level quantizing characteristic is discussed and analyzed. Presumably, for simplicity of analysis, the authors of that technical paper have chosen to show the sampler and quantizer portions of the delta-sigma modulator as distinct entities. A physical configuration of that type cannot be implemented by conventional analog circuitry. An example of a conventional analog delta-sigma modulator is shown and described in a paper (hereinafter called paper No. 2) entitled &#34;A Telemetering System by Code Modulation - Delta-Sigma (ΔΣ)Modulation - Delta-Sigma (ΔΣ) Modulation&#34; by H. Inose, Y. Yasuda and J. Murakami. That paper was published in the IRE Transactions Space Electronics Telemetry, Vol. SET-8, pp. 204-209 (September 1962). In this latter paper the prior art, basically analog, delta-sigma (ΔΣ) modulator is presented and its utility as an improvement over the prior art delta modulator is described. 
     The authors of the latter paper have pointed out that in the still earlier, so-called delta modulation system, pulses are sent over a transmission line carrying information corresponding to the derivative of the input signal amplitude. At the receiving end, those pulses are integrated to obtain the original waveform. Transmission disturbances such as noise, etc., result in a cumulative error as a transmitted signal is integrated at the receiving end. 
     The so-called delta-sigma modulation system provides for integration of the input signal before it enters the modulator itself so that the output transmitted pulses carry information corresponding to the amplitude of the input signal. 
     In a basically analog implementation, the behavior of a delta-sigma modulator depends on the absolute value of capacitors and resistors in the circuits and, therefore, sensitivity to the detrimental effects of aging and temperature is encountered. Further, operational amplifiers included in analog implementations must be of superior quality, the gain and bandwidth of such amplifiers should not be such as to affect the transfer function of the integrator within the input circuitry of the device. Still further, in an analog implementation the waveform at the output of the digital to analog converter has to be precise and not pattern sensitive, that is, a pulse for an isolated &#34;1&#34; must be substantially identical to pulses imbedded in a series of &#34;1-s&#34;. The circuit intricacies of analog delta-sigma modulator implementations are often the result of that requirement. 
     The manner in which the invention deals with these prior art disadvantages will be evident as this description proceeds. A detailed description of a digital delta-sigma modulator can be found in Applicant&#39;s U.S. Pat. No. 4,270,027, entitled &#34;Telephone Subscriber Line Unit with Sigma-Delta Digital-to-Analog Converter&#34;, which patent is assigned to the same assignee as is the instant invention. 
     SUMMARY OF THE INVENTION 
     In view of the disadvantages of the prior art basically analog delta-sigma modulator, it is the general objective of the present invention to provide a delta-sigma modulator in which performance does not depend on the absolute value of capacitors and resistors as is the case with the prior art analog implementations, but rather depends on the ratios of capacitors. These ratios are relatively unaffected by temperature variations and aging. Further, according to the switched capacitor implementation of the invention, operational amplifiers employed need only be able to charge and discharge circuit capacitors in a nominal time (on the order of one-half the sampling interval). This results in insensitivity to drift of element values caused by temperature and aging. 
     Still further, according to the invention, the switched capacitor implementation is based on an operational sequence of alternate charge and discharge of capacitors in non-overlapping intervals of time, such that a digital delta-sigma modulator of the type described in the aforementioned U.S. Pat. No. 4,270,027 is inherently pattern insensitive. That is, an isolated logical &#34;1&#34; is equivalent to a logical &#34;1&#34; in a string of logical &#34;1&#34;&#39;s in terms of charge transfer. 
     Absolute values of capacitors can be chosen so that the design of operational amplifiers is more flexible. The Delta-Sigma modulating characteristics are affected only by the ratio of capacitors. For example, the larger the value of a capacitor, the larger is the slew rate requirement on the operational amplifier which is charging the capacitor. However, the stray capacitance will be more effectively swamped. Conversely, a small capacitance is more easily charged, but the effect of stray capacitance is more pronounced. By leaving the absolute value of the capacitor as a design parameter, greater design latitude is possible in the overall delta-sigma modulator design. 
     Further, a switched capacitor has an inherent sample and hold function at its input. Actually, the signals at all nodes in the circuit change at discrete instants of time. Consequently, the input to the comparator of the circuit is stable when the comparator makes a decision. 
     Basic circuitry, parameter identification and operational considerations are set forth in the detailed description of a preferred embodiment of the invention hereinafter presented. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a schematic block diagram representative of typical analog implementations according to the prior art. 
     FIG. 2 is a schematic block diagram basically representative of switched capacitor implementations. 
     FIG. 3 is a schematic diagram of an implementation of the delta-sigma modulator according to the invention employing +V and -V reference voltages. 
     FIG. 4 is a timing waveform presentation for the circuit of FIG. 3. 
     FIG. 5 shows a portion of a circuit of FIG. 4 with circuitry adapted to the use of a single reference voltage +V. 
     FIG. 6 illustrates the substitution of the single reference voltage circuitry of FIG. 5 into the device of FIG. 3. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENT 
     At the outset, it should be pointed out that the term delta-sigma modulator is sometimes called a sigma-delta modulator, the transposition of the sigma and delta terms being a matter of author&#39;s preference, the same device going under either name. FIG. 1 is prior art as aforementioned and substantially self-explanatory. In FIG. 1 herewith, the D/A converter would be a pulse shaper providing one of two different pulses in accordance with whether the digital signal is high or low, mathematically +A or -A, the quantity A being related to the conversion between a number and a voltage. The basically lowpass (integrating) device labeled H(s) determines the order of the delta-sigma modulator. H(s) is typically a first order filter if H(s) is =g/s and second order ##EQU1## 
     It may be said that the modulator noise (inaccuracy of conversion of the input function to a digital signal) reduction achieved by the (ΔΣ) modulator is the result of keeping track of all previous conversion errors and feeding this information (as an error related signal) back to correct the next conversion. In this process, a first order modulator attempts to zero the average error over a period of time, whereas a second order modulator not only keeps this average error at zero but also keeps the first derivative of the error signal at zero. 
     In a first order system, only a DC input signal will be accurately represented digitally. However, in a second order system, the bandwidth of the signal which can be continuously represented digitally is increased. 
     According to known filter theory, the integrating filter H(S) can be described for a first order (ΔΣ) modulator as: ##EQU2## and similarly, for a second order filter ##EQU3## 
     Capacitor and other component values are chosen to give appropriate values for the coefficients. ΔΣ modulation performance in terms of noise and stability is related to these coefficients. 
     The paper &#34;Reduction of Quantizing Noise by Use of Feedback&#34; (paper No. 1) is a theoretical paper which provides the basis for ΔΣ Modulation. The second paper, &#34;A Telemetering System by Code Modulation - Δ - Σ Modulation&#34; provides an implementation and the underlying analysis of a &#34;first-order&#34; ΔΣ modulator. FIG. 1 of paper 2 shows the &#34;error signal&#34;, S(t)-P(t) being applied to an integrator, which has a first-order transfer function and consequently is a &#34;first-order ΔΣ M&#34;. FIG. 1 of paper 1 depicts the most general case of ΔΣM, in which each H i  (filter transfer function) could be of order greater than 1 and further, considers a general, multilevel, quantizer characteristic. The authors of paper No. 1 have, most probably for simplicity of analysis, chosen to show the sampler and quantizer as distinct entities. This configuration cannot be implemented by conventional analog circuitry as, for example, described in paper No. 2. The sampler and quantizer can, however, be separated in a switched capacitor implementation as in the invention. 
     In FIG. 1 of paper No. 2, the sampling pulse generator and pulse modulator together form an A/D+D/A operation. The pulse modulator outputs a pulse of known shape whose polarity is determined by the polarity of the analog signal at the input of the pulse modulator at the instant the sampling pulse is asserted. 
     Prior art references, for the most part, describe circuits obtained experimentally--a circuit configuration is chosen, an initial &#34;educated&#34; guess of component values is made and the circuit is then refined on a lab bench. 
     Higher order ΔΣM&#39;s provide, potentially, better noise behavior but are notorious for being unstable and are consequently not recommended. The second order embodiment herein described is regarded as optimum. 
     All switched capacitor implementations are generalized in FIG. 2 and can be, mathematically, reduced to the form shown below. 
     H(z) is a discrete-time transfer function, of the form for &#34;second-order&#34; ΔΣM: ##EQU4## where z -1  is the unit delay operator, unit delay being the time of one sampling interval. For a sampling rate of 1MHz, the sampling interval will be seen to be 1 μsec. The coefficients, which determine the noise performance and stability, are functions of capacitor ratios. The absolute value of each capacitor can be chosen by the circuit designer to optimize amplifier performance, to swamp stray capacitance, etc. 
     The switched capacitor ΔΣM according to the invention is best explained in stages. The principle underlying the operation of an ΔΣM is to provide an analog-to-digital conversion wherein the digital word size is small but the sampling frequency is much higher than the highest signal (speech) frequency. 
     First consider the sampling clock which operates the D type (edge-triggered) flip-flop, 20 in FIG. 3. This clock provides the time reference f s  and also two other clock waveforms, at the sampling frequency, but with duty cycle less than 50%. These are designated θ 1  (&#34;charge&#34;) and θ 2  (&#34;discharge&#34;). FIG. 4 depicts these waveforms in a typical relationship. 
     Consider next a section of FIG. 3 consisting of switches 1 and 3, capacitor C1, amplifier 4 and capacitor C3. Here θ 1  from sampling clock 303 controls the switch 1 and θ 2  controls the switch 3. When θ 1  is low, switch 1 is open (open circuit) and when θ 1  is high, switch 1 is closed (short-circuit). Similarly with θ 2  and switch 3. The non-overlapping nature of θ 1  and θ 2  ensures that switches 1 and 3 will not both be closed at any time. Assuming the input signal u(t) remains constant over the interval [nT, (n+1)T], capacitor C 1  will charge during θ 1 , to a voltage equal to u(nT). Assuming the amplifier, 4, is an ideal op-amp, during θ 2  all the charge on C 1  will be transferred to C 3 , causing a change in the voltage across C 3  of ##EQU5## Consequently, at t=(n+1)T, the op-amp output voltage, x, will be 
     
         x[(n+1)T]=x(nT)-(c.sub.1 /C.sub.3) U(nT) 
    
     the negative increment is because the amplifier inverts. Now consider the addition of switches 6, 7 and 9 and capacitor C 2 . If b(n) is +1, i.e., Qn=HIGH, then during θ 1 , C 2  would charge to +V. In short C 2  would charge to -b(n)·V. During θ 2  this charge would be transferred to C 3 . The overall operation of switches 1, 3, 6, 7 and 9 and capacitors C 1 , C 2  and C 3 , and amplifier 4 can be described by the equation: 
     
         x[(n+1)T]=X(nT)-(C.sub.1 /C.sub.3) U(nT)+b(n)·V·(C.sub.2 /C.sub.3) 
    
     similarly, 
     
         W(n+1)T=W(nT)-(C.sub.4 /C.sub.6) X(nT)-b(n) ·V·(C.sub.5 /C.sub.6) 
    
     The dotted enclosures 301 and 302 may be called reference switching means. The operation of the comparator and D flip-flop is to obtain 
     
         b(n+1)=sqn{W[(n+1)T]} 
    
     The noise performance and stability of the ΔΣ modulator is governed by the capacitor ratios (C 1  /C 3 ), (C 2  /C 3 ), (C 4  /C 6 ) and (C 5  /C 6 ) The voltage V is termed the &#34;reference voltage&#34; and normally all voltages are evaluated as fractions thereof. V is sometimes referred to as the &#34;crash point&#34; of the encoder and is the maximum amplitude of the input signal. An input amplitude of greater than V will cause overload. 
     For a typical ΔΣ modulator, the following capacitor ratios were found to be satisfactory. 
     
         (C.sub.1 /C.sub.2)=(C.sub.2 /C.sub.3)=1/2 
    
     
         (C.sub.4 /C.sub.6)=(C.sub.5 /C.sub.6)=1 
    
     It will be noted that the configuration of FIG. 3 requires two reference voltages, +V and -V. If only one reference, for example +V is available, then the configuration consisting of +V, -V, switches 6, 7 and 9 and capacitor C 2  (similarly with +V, -V, switches 13, 14, 16 and capacitor C 5 ) can be replaced by the circuit shown in FIG. 5 and included in FIG. 6 as dotted block 601. 
     In that variation, and during each sampling interval, C 8  charges to V during θ 1  and during θ 2  causes a change of -(C 8  /C 3 ) V in the output of amplifier 4. During θ 1 , C 7  charges to V volts with the polarity indicated shown on FIGS. 5 and 6. If Q n  were &#34;high&#34;, i.e., b(n)=+1, switches 23 and 22 would close during θ 2  and, because of the polarity reversal, cause a change of +(C 7  /C 3 )V in the output of amplifier 4. If b(n)=-1, then C 7  does not discharge into C 3 . The net effect is then: ##EQU6## If C 7  =2C 8 , then the overall operation of the circuit in FIG. 5 can be described as 
     
         x(n+1)T=x(nT)-(C.sub.1 /C.sub.3)U(nT)+b(n)V·(C.sub.8 /C.sub.3) 
    
     similarly 
     
         W(n+1)T=W(nT)-(C.sub.4 /C.sub.6)x(nT)-b(n)V·(C.sub.10 /C.sub.6) 
    
     Implementing the ΔΣ modulator of the invention according to FIG. 6, the following capacitor ratios would be used: 
     
         (C.sub.8 /C.sub.3)=(C.sub.1 /C.sub.3)=1/2 
    
     
         (C.sub.4 /C.sub.6)=(C.sub.10 /C.sub.6)=1 
    
     
         (C.sub.7 /C.sub.8)=(C.sub.9 /C.sub.10)=2 
    
     The sampling rate must be much higher than the highest frequency component of the input signal U(t). The invention is particularly useful for digitally encoding telephone (speech) band signals for transmission through a discrete telephonic channel. Since such signals require only a few KH z  of bandwidth, and accordingly the sampling rate of 1 MH z  typical for the modulator of the invention fulfills the aforementioned requirement. A second order ΔΣ modulator can be considered to be constructed of a first order ΔΣ modulator embedded in a feedback loop. Conversely, a first order ΔΣ modulator can be considered to be a subset of a second order ΔΣ modulator obtained by &#34;stripping&#34; the second order ΔΣ modulator down. The configuration shown in FIG. 6 can be stripped down to form a first order switched capacitor ΔΣ modulator. If amplifier 4, capacitors C 3 , C 1 , C 2  and their associated switches are removed, what remains is a first order ΔΣ modulator which converts an analog signal at U into a digital signal b(n). 
     It is possible to transmit the bit-stream {b(n)} as is and at the receiving end have a simple digital-to-analog converter which comprises of a pulse shaper which puts out distinct waveforms in a bit-interval according as b(n)=&#34;HIGH&#34; or b(n)=&#34;LOW&#34; followed by a simple analog lowpass filter to smooth the waveform. This, however, would entail the transmission of {b(n)} directly, which is about 1 megabit/sec which is quite high. An alternative method, is to employ a sequence of digital lowpass filters which do the &#34;smoothing&#34; while retaining the digital nature of the signal. As a consequence, the number of bits per word increases, that is the granularity of levels, which can be represented, is made fine. In a line circuit such as that of U.S. Pat. No. 4,270,027 as aforementioned, the lowpass filters permit the resampling of the digital signal at 8 Kilowords per sec with a granularity corresponding to 13 bits per word in a uniform code. Each code word can be converted into an 8-bit code, if so desired, corresponding to either the A-law or μ-law format. The 1-bit device, in this sense, does represent a large range of sample values from u(nt). The 1-bit/word, 1 Mword/sec stream (i.e. a 1-Megabit/sec stream) is sometimes referred to in the art as &#34;PULSE-DENSITY MODULATED&#34; version of the complex voice channel signal. 
     While the present invention has been described in connection with a preferred embodiment thereof, it is to be understood that the invention is not limited to telephone system implementations, and that additional embodiments, modifications and application which will become obvious to those skilled in the art are included within the spirit and scope of the invention as set forth by the claims appended hereto.

Technology Category: 5