Patent Document

BACKGROUND OF THE INVENTION 
     An amplitude modulated signal received over a transmission medium can be distorted due to transmission characteristics of the transmission medium, resulting in attenuation of the signal at different frequencies. A receiver typically includes an equalizer for compensating for the distortion due to the frequency response of the transmission medium. Ideally, the frequency response of the transmission medium should be flat; that is, the magnitude of the sidebands should be equal. However, the frequency response can be downward sloping or upward sloping dependent on frequency and phase distortion in the received signal due to the unequal magnitude of the sideband carriers. The frequency response of a filter in the equalizer is generally selected to approximate the inverse frequency response of the transmission medium in order to equalize or “flatten” the frequency response. 
     The characteristics of the transmission medium or the apparatus used to correct the received signal may vary with time or temperature. Thus, in that case, the frequency response of the equalizer must adapt to varying distortion in the received signal. The frequency response must be continuously measured and the inverse frequency response of the equalizer modified in response to changes in the measured frequency response. One known method for adaptively adjusting the inverse frequency response is to directly measure the frequency response of the transmission medium. However, prior art equalizers with adaptive frequency response based on direct measurement of the frequency response of the transmission medium require characterization of the medium for example, using test tones to “flatten” the frequency response. This may be technically inconvenient, if not impossible. 
     SUMMARY OF THE INVENTION 
     In the present approach, rather than measuring the frequency response directly, the orthogonal property of the received signal is used to measure total energy due to unequal magnitude of sideband carriers. A carrier is transmitted and periodically repeated. The carrier is transmitted and received but no signal orthogonal (i.e., lying at right angles) to the carrier is transmitted. Ideally, any orthogonal leakage signal should exhibit no observable energy. Any “leakage” energy that is observed is due to slope error (or delay error) caused by unequal magnitude (or delay) of sideband carriers. The “leakage” energy is measured directly from the leakage signal and the frequency response adjusted to minimize it. 
     In accordance with the present invention, a receiver has apparatus for adaptively minimizing distortion in a received modulated signal. The apparatus includes a leakage sensor for measuring leakage and an equalizer controller for modifying the frequency response based on the measured leakage. The leakage sensor provides a frequency response error based on leakage measured on a leakage signal orthogonal to the received modulated signal. The receiver also includes an equalizer having a frequency response. The frequency response is selected to minimize distortion in the received modulated signal. The equalizer controller coupled to the leakage sensor and the equalizer adjusts the frequency response of the equalizer in the receiver dependent on the frequency response error to minimize the measured leakage. 
     The frequency response error is a difference between currently measured leakage and previously measured leakage. The leakage sensor also includes a decision circuit. The decision circuit modifies the frequency response error provided by the leakage sensor to increase the rate at which the frequency response error is adjusted by the equalizer controller while the measured leakage is outside a first predetermined window. The leakage sensor includes a search limit establisher which modifies the frequency response error provided by the leakage sensor to increase the rate at which the frequency response error is adjusted by the equalizer controller while the measured leakage is outside a second predetermined window and the first predetermined window. 
     The received modulated signal may be an amplitude modulated signal. In an alternate embodiment, the amplitude modulated signal is one component selected from the I and Q components of a Quadrature Modulated signal and the leakage signal is the other component of the I and Q components. In a Quadrature Amplitude Modulation receiver, the leakage is measured during a periodic clock recovery interval. 
     The measured leakage may be the voltage, current or integrated power of the leakage signal. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The foregoing and other objects, features and advantages of the invention will be apparent from the following more particular description of preferred embodiments of the invention, as illustrated in the accompanying drawings in which like reference characters refer to the same parts throughout the different views. The drawings are not necessarily to scale, emphasis instead being placed upon illustrating the principles of the invention. 
         FIG. 1  is a block diagram of a receiver including a spectral slope-induced leakage sensor according to the principles of the present invention; 
         FIG. 2A  is a graphical representation of an amplitude modulated signal; 
         FIG. 2B  is a graphical representation of the frequency spectrum domain for the amplitude modulated signal shown in  FIG. 2A ; 
         FIG. 2C  is a phasor representation of the frequency spectrum domain shown in  FIG. 2B ; 
         FIG. 2D  is a graphical representation of a phasor resulting from the combination of the phasors shown in  FIG. 2C  when there is phase or amplitude distortion between the sideband phasors; 
         FIG. 3A  is a block diagram of an embodiment of the adaptive correction circuit shown in  FIG. 1 ; 
         FIG. 3B  is a block diagram of an embodiment of the delay circuit shown in  FIG. 3A ; 
         FIG. 4  is a graphical representation of power versus frequency illustrating an upward frequency response tilt for the phasor representation of the frequency domain shown in  FIG. 2B ; 
         FIG. 5  is a plot illustrating measured traces in the embodiment of the spectral-slope-induced leakage sensor shown in  FIG. 3A ; 
         FIG. 6  is a block diagram of a QAM receiver including the adaptive correction circuit according to the principles of the present invention; 
         FIG. 7A  is a block diagram of a frame structure for use in the QAM receiver of  FIG. 6 ; and 
         FIG. 7B  illustrates an example of modulation of a carrier frequency using the frame structure shown in  FIG. 7A . 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     A description of preferred embodiments of the invention follows. 
       FIG. 1  is a block diagram of a receiver  100  including an adaptive correction circuit  106  according to the principles of the present invention. The adaptive correction circuit  106  measures “leakage” on the leakage signal  134 . The “leakage” is an observable effect caused or induced by spectral slope. The spectral slope is any tilt in the frequency domain. For example, if viewed on a spectrum analyzer, the spectrum of a signal with spectral-slope-induced leakage is non-symmetrical in power around a mid-point in the frequency axis. 
     An amplitude modulated signal (“AM”)  104  with distortion induced by channel impairments is processed by a voltage controlled equalizer  110 . The voltage controlled equalizer  110  includes a filter having a frequency response selected to approximate the inverse frequency response of the transmission medium over which the AM signal  104  is being transmitted. The inverse frequency response selected is dependent on the voltage on the equalizer control signal  130 . 
     If the inverse frequency response selected in the voltage controlled equalizer approximates the frequency response of the transmission medium, there is no distortion in the modified amplitude modulated signal  132  output from the voltage controlled equalizer  110 . If there is distortion, the distortion results in “leakage” energy which can be measured in a signal orthogonal (lying at right angles) to the modified amplitude modulated signal  132 . 
     For example, in a Quadrature Amplitude Modulation (“QAM”) scheme, two axis are identified: I (for “in-phase”) and Q (for quadrature). The I and Q axis are at a right angle (90°) from each other. The leakage can be measured by measuring the voltage, current or the integrated power of the leakage signal using methods well-known to those skilled in the art. The adaptive correction circuit  106  measures the leakage and modifies the voltage on the equalizer control signal  130  appropriately to minimize the leakage. The frequency response of the voltage controlled equalizer  110  can be continuously modified to minimize distortion in the modified amplitude modulated signal  132  by minimizing the measured leakage. 
     In a one-axis modulation scheme, such as standard AM radio, the quadrature “leakage” is an unwanted orthogonal signal which can be observed using a quadrature demodulation circuit  118 . The quadrature demodulation circuit  118  is coupled to the voltage controlled equalizer  110  for receiving the modified amplitude modulated signal  132 . The quadrature demodulation circuit  118  provides the leakage signal  134 ; that is, a signal orthogonal to the modified AM signal  132 . The quadrature demodulation circuit  118  can be implemented as a Costas Loop. The Costas Loop is a Phase Lock loop which nulls the average D.C. voltage but not A.C. leakage. The A.C. leakage is a distorted version of the A.C. signal. The output  134  of the quadrature demodulation circuit  118  is leakage because no quadrature signal is required. 
       FIG. 2A  is a graphical representation of an amplitude modulated signal  200  received by the equalizer  110  ( FIG. 1 ). The amplitude modulated signal  200  includes a carrier signal  202  with frequency f c  and a modulating signal  204  with frequency f m . The modulating frequency f m  is lower than the carrier frequency f c . In the amplitude modulated signal  200 , the amplitude of the high frequency carrier  202  is controlled by the low frequency modulating signal  204 . 
       FIG. 2B  is a graphical representation of the frequency spectrum domain for the amplitude modulated signal  200  shown in  FIG. 2A . The frequency of the modulating signal (f m ) and the frequency of the carrier signal (f c ) are expressed in radians (w) where w c =2πf c . The frequency spectrum domain for the amplitude modulated signal  200  can be represented as the sum of three signals each having a different frequency; a carrier having frequency (w c ), an upper sideband having a frequency equal to the sum of the modulating frequency and the carrier frequency (w c +w m ) and a lower sideband having a frequency equal to the difference between the carrier frequency and the modulating frequency (w c −w m ). The amplitude of the upper sideband and the lower sideband is equal if there is no distortion in the amplitude modulated signal  200 . 
     Returning to  FIG. 1 , the adaptive correction circuit  106  is coupled to the quadrature demodulation circuit  118  for receiving the leakage signal  134 . The amplitude of the leakage signal  134  is related to the frequency response error in the voltage controlled equalizer  110 . The dependency can be shown by representing the carrier and sidebands in the modified AM signal  132  shown in  FIG. 2B  as phasors.  FIG. 2C  is a phasor representation of the frequency spectrum domain shown in  FIG. 2B . The carrier with frequency w c  is represented as phasor E c . The lower sideband (w c −w m ) is represented as phasor E L  and the upper sideband (w c +w m ) is represented as phasor E u . The E u  and E L  sideband phasors counter-rotate while the carrier phasor E c  is stationary and the magnitude of the carrier phasor E c  remains constant. If there is no distortion, both sideband phasors E u  and E L  have the same amplitude and retain the same phase difference producing a resultant phasor E R  parallel to carrier phasor E c  The amplitude of the resultant phasor E R  varies dependent on the phase difference between sideband phasors E u  and E L . 
     Phasor E R  can be represented as a complex number with a real and an imaginary component. The real component of the complex number is the attenuated magnitude observed on the modified AM signal  132  ( FIG. 1 ). The imaginary component of the complex number can be observed on the leakage signal  134  ( FIG. 1 ) that is phase shifted by 90 degrees from the modified AM signal  132  (i.e. orthogonal to the modified AM signal.) The relationship between the modified AM signal  132  ( FIG. 1 ) and the leakage signal  132  ( FIG. 2 ) is shown below:
 
 e   jwct =cos  w   c   t+j  sin  w   c   t 
 
Where:
         e jwct  is a counterclockwise rotating phasor of unit length;   cos w c t represents the real part (projected on the x axis) and corresponds to the modified AM signal; and   sin w c t represents the component orthogonal to the modified AM signal and corresponds to the leakage signal.       

     When phasors E u  and E L  are equal, the component orthogonal to the received signal is zero.  FIG. 2D  is a graphical representation of phasor E R  after combining the unequal phasors E L  and E U  shown in  FIG. 2C . Due to the unequal amplitude of phasors E U  and E L , phasor E R  has a component orthogonal to the received signal  254 . The received signal  250  is the projection on one axis, the component orthogonal to the received signal  254  is the projection onto the other (j) axis. With no distortion, the component orthogonal to the received signal is zero. A component with non-zero amplitude orthogonal to the received signal reduces the amplitude of the received signal. As shown, the amplitude of received signal  250  with a non-zero component orthogonal to the received signal  254  is smaller than received signal  252  with no component orthogonal to the received signal. The amplitude of the component orthogonal to the received signal is related to the distortion in the received signal. 
     Thus, the frequency response error in the equalizer  110  ( FIG. 1 ) is related to the amplitude of the leakage signal  134  ( FIG. 1 ). By minimizing the amplitude of the leakage signal  134  ( FIG. 1 ), the frequency response error is minimized and distortion in the modified AM signal  132  is minimized. 
       FIG. 3A  is a block diagram of an embodiment of the adaptive correction circuit  106  shown in  FIG. 1 . The adaptive correction circuit  106  includes an equalizer controller  312  and a spectral-slope-induced leakage sensor. The spectral-slope-induced leakage sensor computes a frequency response error  330  based on leakage measured on the leakage signal  134 . The spectral-slope-induced leakage sensor includes a leakage amplifier  300 , a delay circuit  302 , close enough decision circuit  304 , search limit establisher circuit  310 , a switch  306  and a summer  308 . The equalizer controller  312  modifies the voltage on the equalizer control signal  130  dependent on the frequency response error  330 . 
     The leakage signal  134  is amplified by the leakage amplifier  300  so that the leakage can be detected. The leakage amplifier  300  can be any AC amplifier having sufficient gain to amplify the leakage signal  134  so that it can be used by the delay circuit  302 . As the distortion in the modified amplitude signal  132  ( FIG. 1 ) is minimized, the amplitude of the leakage signal  134  decreases close to zero. The gain of the leakage amplifier  300  is selected to peak near the highest data modulation frequency sidebands because the highest data modulation frequency sidebands are the sidebands most likely to be affected by a non-flat frequency response slope in the voltage controlled equalizer  110  ( FIG. 1 ). In an alternate embodiment, the leakage amplifier  300  may not be required, if the delay circuit  302  is sufficiently sensitive to a low amplitude leakage signal. 
     The leakage amplifier  300  is coupled to the delay circuit  302 . The delay circuit measures the leakage due to the frequency response error. The delay circuit  302  receives the amplified leakage signal  326  and compares the amplitude to a previously measured (stored) amplitude on the amplified leakage signal  326 , in order to determine if the leakage is increasing or decreasing after each subsequent measurement. The amplitude at the output of the delay circuit  302  determines the direction (increase or decrease) in which the amplitude on the equalizer control signal  130  must be modified to correct the frequency response slope and minimize the distortion. The delay circuit  302  computes the “numerical difference” between the current leakage amplitude and the stored leakage amplitude. 
       FIG. 3B  is a block diagram of an embodiment of the delay circuit  302  shown in  FIG. 3A . The delay circuit  302  includes a subtractor  350  and a delay  352 . The delay  352  is the time between measurements. The “leakage” is stored by passing the amplified leakage signal  326  through the delay  352 . In the embodiment shown, a gain of 0.9 is applied to the stored leakage  354  and a gain of −1 is applied to the amplified leakage signal  326 . The stored leakage  354  is compared with the current leakage in the subtractor  350  to compute the “numerical difference”. The delayed path gain is unbalanced compared to the direct path so that a non-zero numerical difference is output even when there is no difference between the current leakage and the stored leakage, so that the adaptive correction circuit  106  continuously monitors the leakage. The unbalance of the gain in the embodiment shown is 10% (−1+0.9). As noted above, the numerical difference is used to determine the direction in which the control voltage  130  should be modified to minimize distortion in the modified amplitude modulated signal  132  by minimizing the measured leakage in the leakage signal  134 . In the embodiment shown, the stored leakage and current leakage is a voltage. However, in alternate embodiments, the leakage can be current, power or a numerical value. 
     Returning to  FIG. 3A , the output of the delay circuit  302  is coupled to the close enough decision circuit  304 . The close enough decision circuit  304  is a symmetrical window comparator and includes a full-wave rectifier  314  and a comparator  316 . The full wave rectifier  314  includes two diodes  346 ,  348  and an inverter  349 . The full wave rectifier  314  provides a D.C. level voltage from the A.C. voltage on the output  334  of the delay circuit  302 . 
     The full-wave rectifier  314  is coupled to comparator  316 . Comparator  316  compares the D.C. voltage output by the full-wave rectifier  314  with a D.C. threshold voltage  347 . The close enough decision circuit  304  indicates whether the numerical difference is “close enough”; that is, inside a predetermined voltage range. If the numerical difference is inside the predetermined voltage range or window, the voltage on the equalizer control voltage signal  130  is “close enough” to the voltage required to minimize distortion. Further reduction of the distortion is not necessary. For example, the window can be selected so that the “close enough” value reduces the leakage signal to eliminate 90% of the distortion. 
     In one embodiment, the output of the close enough decision circuit  304  toggles between 0V and 1V. The output switches to 1V when the input voltage increases above 0.1625V and switches to 0V when the input voltage decreases below 0.15V. Thus, output of the close enough decision circuit  304  remains at 0V while the voltage on the output  334  of the delay circuit  302  is below the threshold voltage of 0.15V, that is, while the distortion is about 10%. While the output of the close enough decision circuit  304  is 0V, the magnitude of the frequency response error  330  is the same as the magnitude of the leakage signal  334  coupled through the summer. While the output of the close enough decision circuit  304  is 1V, the magnitude of the frequency response error  330  is increased by 1V. 
     The search limit establisher circuit  310  is a symmetrical window comparator. In the embodiment shown, the symmetrical window comparator includes a full-wave rectifier  320  and comparator  322 . The operation of the search limit establisher circuit  310  is the same as already discussed for the close enough decision circuit  304 . 
     The output of the close enough decision circuit  304  is coupled to the A-input of a switch  306 . If numerical difference is not “close enough”, the 1V on the output of the close enough decision circuit  304  closes the switch  306 , allowing the output of the search limit establisher circuit  310  to be superimposed on the delay circuit output  334 . The search limit establisher circuit  310  is switched in to establish a search limit. 
     The output  336  of search limit establisher circuit  310  is a D.C. voltage much larger than the voltage on the output of the delay circuit  302 . The D.C. voltage at the output of switch  306  is summed with the output  334  from the delay circuit  302  in the summer  308 . The large D.C. voltage on the output  330  of the summer  308  input to the equalizer controller  312  causes the equalizer controller  312  to ramp (“slew”). 
     The large D.C. voltage is applied to the equalizer controller  312  in a sustained manner in order to keep the equalizer controller  312  slewing by providing hysteresis in the search limit establisher circuit  310 . In one embodiment, the trigger points of the search limit establisher circuit  310  are ±05V and the output of the search limit establisher circuit  310  is ±8V. The trigger points cause the equalizer controller ramp to reverse direction if the input voltage is ±0.5V. There is a large hysteresis of ±0.5V. If the control voltage moves outside this ±0.5V window, the output of the search limit establisher circuit  310  switches voltage level to provide either +8V or −8V at the input of the equalizer controller. 
     Thus, a D.C. voltage of ±8V is applied to the input to the equalizer controller  312  while the numerical difference output  334  from the delay circuit  302  is outside the window established by the close enough decision circuit  304 . 
     Once the numerical difference output  334  of the delay circuit  302  is within the window defined by the close enough decision circuit  304 , the output of the close enough decision circuit  304  switches to 0V. The 0V at the A input of the switch  306  opens the switch  306 . While switch  306  is open, the output of the search limit establisher circuit  310  is decoupled from the equalizer controller. However, the output of delay circuit  302  remains coupled to the equalizer controller  312  through the B input of summer  308 . As discussed, in the delay circuit  302 , the stored leakage and the current leakage are amplified by different gain values. Thus, even when the “leakage” is minimized there is still a non-zero voltage at the output of the delay circuit  302  which is provided to the input of the equalizer controller  312 . This small voltage allows the leakage to be continuously monitored so that the leakage signal can be continuously polled to detect when the distortion increases above 10%. 
     The equalizer controller  312  can be implemented as a linear integrator (analog integrator) or a non-linear integrator (digital integrator). For example, the digital integrator can be implemented as an up-down counter with a plurality of voltage steps corresponding to the count value. In an alternate embodiment, the equalizer controller can be implemented as a hunt-and-seek algorithm using successive approximation. The algorithm monitors divergence of the frequency response error and reverses the direction of the equalizer control upon detecting divergence. In the embodiment shown, the equalizer controller is an analog integrator and the output voltage is linear as shown in  FIG. 5 . 
     The voltage of the leakage signal  104  is related to the slope of the frequency response in the equalizer  110 .  FIG. 4  is a graphical representation of power (dB) versus frequency illustrating an upward frequency response tilt for the phasor representation of the frequency domain shown in  FIG. 2B . As shown, the upper sideband power is greater than the lower sideband power, indicating a frequency response tilt with a positive slope. 
     An embodiment of the invention has been described for an AM signal. However, the invention is not limited to Amplitude Modulation. The invention can also be used to minimize distortion in any orthogonal modulation scheme, for example, DQPSK (Differential Quadrature Phase Shift Keying), QPSK (Quadrature Phase Shift Keying), QAM (Quadrative Amplitude Modulator), 9QPR (9-point Quadrative Partial-Response Modulator), 16QAM, 256QAM and Phase Modulation. 
       FIG. 5  is a plot illustrating measured traces in the embodiment of the spectral-slope-induced leakage sensor  106  shown in  FIG. 3A . Three traces are shown. Trace  500  corresponds to the voltage of the equalizer control signal  130  at the output of the equalizer controller  312 . Trace  502  corresponds to the close enough signal  332  at the output of the close enough decision circuit  304 . Trace  504  corresponds to the output  334  of the delay circuit  302 . 
     As shown, trace  504  is −1V at time=1 second, indicating a change in leakage signal detected by the delay circuit  302  requiring modification of the equalizer control signal voltage. The −1V results in switching the output of the close enough decision circuit (trace  502 ) to 1V. Trace  502  remains at 1V until the output of the delay circuit  302  shown on trace  504  decreases below −0.15V as the equalizer control signal voltage decreases as shown on trace  500 . 
     Once the delay circuit output voltage (trace  504 ) decreases below 0.15V, the output of the close enough decision circuit  304  (trace  502 ) switches to 0V. The 0V on the A input to the switch  306  acts to “disconnect” the search circuit limit establisher circuit  310  from the feedback loop by multiplying the output of the search circuit limit establisher  310  by 0. The output of the search circuit limit establisher circuit  310  remains at the same state while the search circuit limit establisher circuit is disconnected. 
     The output  334  of the delay circuit  302  is coupled to the summer  308 . Thus, any change in the voltage on the leakage signal  134  results in a corresponding change in the input voltage to the equalizer controller  312 . The change in the input voltage results in a corresponding change in the equalizer control signal voltage  130  at the output of the equalizer controller  312 . 
     A decrease in the equalizer control voltage  130  at the output of the equalizer controller  312  results in a corresponding decrease in the voltage on the delay circuit output  334 . At 2.75 seconds, the voltage on the delay circuit output  334  reaches the hysteresis trigger point of single sided comparator  316 , resulting in 1V on the close enough decision circuit output  332 . The 1V opens the switch  306  and the voltage on the output of the search limit establisher  310  controls the voltage on the equalizer controller input  330 . 
     The search limit establisher output  336  decreases the equalizer control voltage in a negative direction until the hysteresis trigger point for the search limit establisher  310  is reached. As shown, at 2.85 seconds, the hysteresis trigger point is reached and the equalizer control voltage starts to increase in the positive (correct) direction. Equalizer control voltage  500  continues to increase until the close enough decision circuit output  332  switches to 0V and the search circuit limit establisher  310  is “disconnected” from the feedback loop controlling the equalizer controller input voltage  330 . 
     The search circuit limit establisher output  336  is now in the correct state and is connected as needed to increase the control voltage to overcome the steady state positive bias caused by the delay circuit  302 . The search circuit limit establisher output  336  is only connected briefly as shown by the positive spikes on the timing diagram because the search circuit limit establisher output is the correct polarity; that is, negative. 
     In an alternate embodiment, if the search circuit limit establisher output  336  is initially negative, the equalizer control voltage remains positive, eliminating the negative spike at 2.75 seconds. 
     The 0V on the output of the close enough decision circuit  304  results in a small decrease in the equalizer control voltage. At 2.7 seconds, the equalizer control voltage decreases to a voltage level resulting in an increase in the delay circuit voltage  334  and the output of the close enough decision circuit  304  switches to 1V until the delay circuit output  334  decreases below 0.15V. 
     The equalizer control voltage  130  continues to change to provide a difference in the delay circuit  302  to allow adjustment of the equalizer control voltage  130  to reduce the distortion. Positive feedback exists around the loop, which creates a “flip-flop” behavior outside the hysteresis window (0.1625V–0.15V). 
     As discussed in conjunction with  FIG. 3B , there is an unbalance in the delay circuit  302  due to the unequal gain in the stored leakage and the current level of leakage. When the leakage is minimized, the 0.15V on the output of the delay circuit  302  results in 0.75V at the input of the equalizer controller  312 . Thus, the equalizer controller output (equalizer control) will increase resulting in the delay circuit output voltage  334  greater than 0.15V. In the embodiment shown, the equalizer controller is an analog integrator. A very small “hunting” results as shown by the positive pulses on trace  502 . The hunting can be reduced by replacing the analog integrator with a digital integrator or by implementing a non-linear integrator in software. 
       FIG. 6  is a block diagram of a QAM receiver  604  including the adaptive correction circuit  600  according to the principles of the present invention. The QAM receiver  604  is described in U.S. patent application Ser. No. 09/952,321 filed Sep. 13, 2001 entitled “Broadband System With Topology Discovery”, by Gautam Desai, et al, the entire teachings of which are incorporated herein by reference. 
     The receiver  604  receives a quadrature-multiplexed signal  602  which includes in-phase (I) and quadrature (Q) carriers. At the front end, the receiver  604  includes low-noise amplifier (LNA)  650 , equalizer  652  and automatic gain control (AGC)  654 . The received signal  602  is boosted in the LNA  650  and corrected for frequency-dependent line loss in the equalizer  652 . The equalized signal is passed through the AGC stage  654  to I and Q multiplier stages  656 ,  658 , low pass filters  660  and analog-to-digital converters (ADC)  662 . After down-conversion in the multiplier stages  656 ,  658  and low-pass filtering, the I and Q channels are digitized and passed on to the QAM-to-byte mapper  629  for conversion to a byte-wide data stream. 
     Carrier and clock recovery, for use in synchronization at symbol and frame levels, are performed during periodic training periods. A carrier recovery PLL circuit  668  provides the I and Q carriers from the RF carrier (RFin)  620  to the multipliers  656 ,  658 . The RF carrier  620  includes the I and Q carriers. A clock recovery delay locked loop (DLL) circuit  676  provides a clock to the QAM-to-byte mapper  649 . During each training period, to perform carrier and clock recovery, PLL and DLL paths that include F(s) block  674  and voltage controlled oscillator (VCXO)  670  are switched in using normally open switch  673  under control of SYNC timing circuit  672  in order to provide updated samples of phase/delay error correction information. 
     In the embodiment shown, only the I carrier is transmitted during the training period. With no distortion, the measured leakage on the Q carrier orthogonal to or in quadrature with the I-signal is close to zero. After the carriers have been recovered by the carrier recovery PLL  668 , the adaptive correction circuit  600  measures the leakage on the Q carrier. The frequency response of the equalizer  652  is modified to minimize leakage on the Q carrier and thus, minimize the distortion. In an alternative embodiment, the Q carrier is transmitted during the training interval and the leakage on the I carrier is measured and minimized after the carriers have been recovered. 
       FIG. 7A  is a block diagram of a frame structure  720  for use in the QAM receiver  604  of  FIG. 6 . The frame structure  720  is used to transmit a frame over the network. The frame structure  720  includes frame synchronization (or carrier synchronization)  700 , symbol synchronization  702  and a data phase  704 . In a particular embodiment, frame and symbol synchronization is performed every 10 micro seconds (μs) followed by 1280 bytes of Data Phase  721 , with frame synchronization (FS)  600  for 1 micro second (μs) and the symbol synchronization (SS)  702  for 400 nano seconds (ns). There is a quiet interval before and after the frame synchronization  700 . The quiet interval before the frame synchronization is referred to as the front porch  700 A frame and the quiet time after the frame synchronization is referred to as the back porch  700 B. The front porch  700 A and back porch  700 B provide sufficient time for closing or opening a switch, for example, switch  673  ( FIG. 6 ) at the beginning and end of the frame synchronization  700  period. In one embodiment the front and back porches are about 300 ns for a frame synchronization period of about 1 μs. It should be understood that other frame structures are possible and the frame structure described is only an example. 
     During the frame synchronization period  700 , the carrier recovery PLL  668  recovers the I carrier. After the I carrier is recovered, the leakage on the Q carrier is measured during the symbol synchronization period  702 , to minimize distortion in the recovered carrier. 
       FIG. 7B  illustrates an example of modulation of a carrier frequency using the frame structure shown in  FIG. 7A . An adaptive correction circuit  600  ( FIG. 6 ) uses the orthogonal property of the received QAM signal to measure the frequency response. During training intervals, a carrier is transmitted and periodically repeated. As discussed in conjunction with  FIG. 7A , in a particular embodiment, symbol synchronization can be performed every 10 μs for 400 ns. During symbol synchronization, the carrier is transmitted only on the in-phase (I) or the quadrature (Q) channel, but not both. The other channel ‘Q’ or ‘I’ should exhibit no observable energy. Any energy observed on the other channel is due to frequency response error. 
     The I and Q modulation is interrupted once every 10 μs so that the receiver can be synchronized to the transmitter. The frame synchronization period  700  is used to recover the carriers. During the symbol synchronization period  702  after the carriers have been recovered a 155.52 MHz A.C. signal is received on the I channel. The symbol synchronization period is used to measure leakage in the Q carrier. The synchronization includes carrier, symbol and bit rate synchronization. The repetitive interruption of the symbol synchronization every 10 μs is creates 100 KHz sidebands as discussed in conjunction with  FIG. 2B . The 100 KHz sidebands are symmetric around 155.52 MHz. Thus, any leakage in the Q channel due to distortion in the I channel also includes the 100 KHz sidebands. The leakage is measured on the Q channel after the carrier recovery PLL  468  ( FIG. 6 ) is locked; that is, after the carriers have been recovered. 
     While this invention has been particularly shown and described with references to preferred embodiments thereof, it will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the scope of the invention encompassed by the appended claims.

Technology Category: 5