Patent Document

BACKGROUND OF THE INVENTION 
     In communication systems, signal generators such as transmitters and receivers commonly employ filters to remove noise in the signal path. The systems typically employ static filters that are designed to remove noise in specified frequency ranges. For example, noise in a transmitter may include spurious noise frequencies (also referred to as spurs) resulting from component mismatches and signal leaks, as well as harmonic noise due to mixing the intermediate frequency signal with the reference frequency signal. The noise signal may vary for different signal bands. Therefore, the frequency range that should be filtered is often variable. A static filter is usually unable to eliminate the harmonic noise frequencies that falls within the filter&#39;s bandwidth. It would be desirable to have a technique that could reduce variable frequency noise. It would also be desirable if the circuitry could be implemented without significantly increasing the complexity of the existing systems. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Various embodiments of the invention are disclosed in the following detailed description and the accompanying drawings. 
         FIG. 1  is a diagram illustrating a signal generator embodiment. 
         FIG. 2  is a block diagram illustrating a transmitter embodiment. 
         FIGS. 3A-3C  are diagrams illustrating changing filter characteristics for different f IF , according to some embodiments. 
         FIGS. 4A-4B  illustrate the difference in transfer function between a reference path low pass filter and a feedback path low pass filter according to one embodiment. 
     
    
    
     DETAILED DESCRIPTION 
     The invention can be implemented in numerous ways, including as a process, an apparatus, a system, a composition of matter, a computer readable medium such as a computer readable storage medium or a computer network wherein program instructions are sent over optical or electronic communication links. In this specification, these implementations, or any other form that the invention may take, may be referred to as techniques. In general, the order of the steps of disclosed processes may be altered within the scope of the invention. 
     A detailed description of one or more embodiments of the invention is provided below along with accompanying figures that illustrate the principles of the invention. The invention is described in connection with such embodiments, but the invention is not limited to any embodiment. The scope of the invention is limited only by the claims and the invention encompasses numerous alternatives, modifications and equivalents. Numerous specific details are set forth in the following description in order to provide a thorough understanding of the invention. These details are provided for the purpose of example and the invention may be practiced according to the claims without some or all of these specific details. For the purpose of clarity, technical material that is known in the technical fields related to the invention has not been described in detail so that the invention is not unnecessarily obscured. 
     A technique of generating an output signal is disclosed. In some embodiments, an input signal is mixed with a reference signal to obtain an intermediate frequency signal, which is filtered by a filter with a filter characteristic that is configured according to the intermediate frequency. One example of the configurable filter characteristic includes a bandwidth requirement that may change for different inputs. The filtered signal is frequency translated to obtain the desired output signal. In some embodiments, an input signal and a filtered intermediate frequency signal are compared, and the difference is used by a voltage controlled oscillator to generate an output signal. Frequency translation is performed on the output signal to generate an intermediate frequency signal having an intermediate frequency. The intermediate frequency signal is filtered by a configurable filter to generate the filtered intermediate frequency signal. The configurable filter has a filter characteristic that is configured according to the intermediate frequency signal. 
       FIG. 1  is a diagram illustrating a signal generator embodiment. In this example, transmitter  100  includes an intermediate frequency (IF) generator  104 , a frequency translation loop  106  and a reference frequency generator  102 . IF generator  104  receives an input signal  108  and mixes the input signal with a local oscillator frequency signal f LO1    116  to generate an intermediate frequency signal  110  with a center frequency at f IF . In the example shown, the input signal is at baseband, thus f IF  is approximately equal to f LO1 . The intermediate frequency signal is sent to a frequency translation loop  106  to be modulated to radio frequency (RF). RF output  112  is generated by frequency translation loop  106 . 
     In the example shown, a reference source such as a temperature controlled crystal oscillator (TCXO) generates a reference frequency signal f TCXO    114 . Reference frequency generator  102  uses f TCXO    114  to generate local oscillator frequencies f LO1  and f LO2 , which are required by IF generator  104  and frequency translation loop  106 , respectively. Noise signals such as harmonics of f IF , f LO1 , f LO2  or combinations thereof may vary for different input frequencies. As will be shown in more details below, IF generator  104  and/or frequency translation loop  106  use filters with configurable filter characteristics to remove variable frequency noise. 
       FIG. 2  is a block diagram illustrating a transmitter embodiment. In this example, transmitter  200  includes an IF generator  204 , a frequency translation loop  206  and a reference frequency generator  202 . The inphase component (I in ) and the quadrature component (Q in ) of the input signal are sent to baseband mixers  208  and  210  respectively. The input components are mixed with a local oscillator frequency f LO1  and the mixer outputs are then combined by combiner  211 . The path of the input signal via the IF generator is sometimes referred to as the reference path, thus low pass filter (LPF)  212  is referred to as the reference path LPF. Reference path LPF  212  is configured to filter combined signal  213  to generate an IF signal f IF    244 . 
     The filter signal is sent to frequency translation loop  206 . The frequency translation loop is so named because during operation, the circuitry translates the loop input into RF. In the example shown, the path of the signal through the frequency translation loop is referred to as the feedback path. Frequency translation loop  206  is formed by a phase-locked loop that includes a phase frequency detector (PFD)  214 , a loop filter  216 , a voltage controlled oscillator  218 , a mixer  222  and a feedback path LPF  220 . The output of phase frequency detector  214  is filtered by loop filter  216  and then sent to voltage controlled oscillator  218  to generate an RF output signal  240  with a center frequency f RF . The output is fed back to mixer  222 , which demodulates the output from RF to IF. The demodulated signal is filtered by feedback path LPF  220  and the filtered signal is sent to PFD  214  for comparison of phase and frequency. Once the phase locked loop enters the locking state, inputs to PFD  214  (i.e. signals  242  and  244 ) will track each other. In other words, signal  242  will have approximately the same phase and frequency as f IF    244 . 
     Local oscillator frequencies f LO1    250  and f LO2    252  are supplied by reference frequency generator  202 . The reference frequency is sent to a fractional PLL, which includes phase frequency detector  224 , loop filter  226 , voltage controlled oscillator  228  and fractional divider  230 . The input of the PLL is reference signal f TCXO    246  and the output of the PLL is f PLL    248 . When the PLL is in its locking state, the frequency of the output signal generated by the fractional PLL is equal to the frequency of the reference signal multiplied by the value of fractional-divider  230 . For example, if f TCXO  has a frequency of 26 MHz and the fractional divider has a fractional value (K.f) of 153.8, then the frequency of f PLL  is approximately 4 GHz. The PLL output is then frequency divided by dividers  232  and  234  to generate mixer local oscillator signals f LO1    250  and f LO2    252  respectively. Divider  232  has a value of R and divider  234  has a value of N. In the example shown, R and N are integers. Fractional R and N values are also possible in some embodiments. 
     Sometimes it is useful to vary the frequency of f IF  during transmission to avoid spurs that are substantially close to the mixer local oscillator signal f LO2  and may degrade output signal (such spurs are sometimes referred to as close-in spurs). The close-in spurs typically arise due to imperfections in the reference frequency generator  202 . Examples of such imperfections include the coupling of other signals (such as harmonics of the f TCXO ) into the PLL and charge-pump current mismatch in the PFD. For a fractional PLL, the spurs tend to have frequency of N×f TCXO , where N is an integer. Spurs may also appear at 0.5×N×f TCXO , 0.25×N×f TCXO  or 0.125×N×f TCXO , etc. In general, for a fractional PLL, spurs may appear at (½ M )×N×f TCXO , where M and N are positive integers. 
     Strong close-in spurs are problematic since they will mix with the signal at f RF , resulting in undesired signal components. The undesired signal components are added to the desired output. The addition will degrade the quality of the output signal since feedback path LPF  256  cannot separate the undesired and the desired components. The spurs closer in frequency to the desired signal and the spurs corresponding to smaller values of M are typically stronger. For example, consider a fractional PLL with f TCXO  of 26 MHz and output frequency f PLL =3534 MHz=135.9230769×26 MHz. For M=0, 1 and 2, spurs may appear at 135×26=3510 MHz, 135.25×26=3516.5 MHz, 135.5×26=3523 MHz, 135.75×26=3529.5 MHz, and 136×26=3536 MHz. Because of frequency division, spurs substantially close to f PLL  will result in close-in spurs of f LO2  and cause distortion. In this case, the spur appearing at 3536 MHz will introduce the most distortion since it is only 2 MHz away from the desired f PLL  frequency of 3534 MHz. The presence of the spur in the output of the PLL leads to a close-in spur in the reference signal generated. Other potential spurs further away from 3534 MHz are less problematic since they tend to be weaker. 
     In the embodiment shown, IF generator  204  is configurable, allowing f IF  to shift when the spurs degrade output quality. During operation, if any strong close-in spur is present, the IF generator is reconfigured to generate a different f IF . Changing f IF  changes the PLL output frequency f PLL  and results in a different spur profile. When an appropriate f IF  is chosen, the resulting spurs move further away from f IF  and can be more easily filtered. In some embodiments, the f IF  is chosen with the additional constraint of also moving harmonic frequencies away from f PLL  and f LO2 . Reference path LPF  212  and feedback path LPF  220  are configurable in the embodiment shown. The parameters of one or both of the LPFs are configured according to f IF  to better remove the noise. 
     Control signals  254  and  256  are used to adjust the parameters of reference path LPF  212  and feedback path LPF  220  respectively to achieve appropriate filter characteristics. In some embodiments, the control signals are derived from the values of N and R. The frequency relationship of the signals associated with the frequency translation loop PLL is expressed as:
 
 f   LO2   −f   IF   =f   RF   (equation 1),
 
where
 
 f   LO2   =f   PLL   /N   (equation 2), and
 
 f   IF   =f   LO1   =f   PLL   /R   (equation 3).
 
     Given a specific output frequency, an appropriate f IF  can be selected to avoid strong close-in spurs. Consider again the numerical example given above. The desired RF frequency is 824.6 MHz. One way to generate the desired f RF  is to choose f IF =58.9 MHz and f LO2 =883.5 MHz. In this case, N=4, R=60, and f PLL =3534 MHz. Based on equation 1, f RF =f LO2 −f IF , or 824.6=3534/4−3534/60=883.5−58.9. 
     As shown above, there is a potentially strong spur that is 2 MHz away from the desired PLL frequency of 3534 MHz. This spur translates to a close-in spur of 884 MHz. One way to avoid this spur is to choose a new f IF =63.43076923 MHz and f LO2 =888.0307692 MHz. For this case, N=4, R=56, and f PLL =3552.123077 MHz=136.6201183×26 MHz. Considering M=0, 1 and 2, spurs may appear at 136×26=2536 MHz, 136.25×26=3542.5 MHz, 136.5×26=3549 MHz, 136.75×26=3555.5 MHz, and 137×26=3562 MHz. In this case, the closest spur to is weaker since it is both further away (3.123 MHz vs 2 MHz)), and the value of M is larger (M=1 vs. M=0). 
     In some embodiments, R is substantially greater than N. In an example with a 4 GHz f PLL , N is 4 and R is approximately 50. For configuration purposes, f RF , f PLL  and N are chosen to be fixed values while the value for R is adjusted to obtain the desired f IF . The filter characteristics are adjusted according to f IF . In some embodiments, the bandwidth of the LPF is configured such that f IF  falls substantially within the bandwidth while the harmonic noise frequencies are located substantially outside the bandwidth. Details of the tuning process are discussed in U.S. patent application Ser. No. 10/854,027 filed May 25, 2004, entitled DIGITAL NOISE COUPLING REDUCTION AND VARIABLE INTERMEDIATE FREQUENCY GENERATION IN MIXED SIGNAL CIRCUITS, which is incorporated herein by reference for all purposes. 
       FIGS. 3A-3C  are diagrams illustrating changing filter characteristics for different f IF , according to some embodiments. In the example shown, the 3 dB roll off point of the low pass filter&#39;s transfer function is set to be 10% greater than f IF . Other values may be used in different embodiments. As f IF  changes, the bandwidth of the filter also changes to ensure that frequency of interest, f IF , is preserved in the filter output and the noise harmonics associated with f IF  are removed. 
     In some embodiments, the harmonic noise to be filtered by feedback path low pass filter  220  are spaced further apart than the harmonic noise to be filtered by reference path low pass filter  212 . The requirements of feedback path LPF  220  can be relaxed consequently so that the feedback path LPF can be implemented more efficiently.  FIGS. 4A-4B  illustrate the difference in transfer function between a reference path low pass filter and a feedback path low pass filter according to one embodiment. The harmonics of f IF  are spaced relatively close, thus filter transfer function  402  of the reference path LPF is required to have a relatively steep roll off in order to reject the harmonic frequencies. In contrast, in the feedback path, the harmonics of f LO2  are spaced further apart, hence filter transfer function  404  of the feedback path LPF has a roll off that is more gradual. The difference in the filter transfer functions indicate that the feedback path LPF can be implemented as a lower order filter than the reference path LPF. The resulting feedback path LPF has smaller circuit area and consumes less power than the reference path LPF, achieving over all cost and power savings. 
     Although the foregoing embodiments have been described in some detail for purposes of clarity of understanding, the invention is not limited to the details provided. There are many alternative ways of implementing the invention. The disclosed embodiments are illustrative and not restrictive.

Technology Category: h