Patent Document

FIELD OF THE INVENTION  
       [0001]     The present invention relates to high bandwidth, high transconductance electronic integrated circuits, and particularly to high bandwidth, high transconductance source follower amplifiers having class AB operation characteristics.  
       BACKGROUND OF THE INVENTION  
       [0002]     High frequency electronic integrated circuits (IC) used in modern electronic communications and computing devices require high bandwidth amplifiers. Furthermore, the low voltages used in modem IC designs lead to a need for high bandwidth amplifiers having a high transconductance, that is, amplifiers capable of generating significant change in output current for a relatively small change in input voltage. Such devices ideally should also have high input impedance and a choice of high or low output impedance. And, because modem electronic devices are increasingly portable, battery-powered devices, it is also highly desirous that the high bandwidth amplifiers be as energy efficient as possible.  
         [0003]     Traditionally, source follower amplifiers have been used to provide high bandwidth amplifier outputs. These circuits are well known in the art and are typically used to buffer signals and provide low output impedance circuits capable of driving loads at high frequency. However, traditionally designed source followers have relatively low transconductance, which limits their ability to drive loads, especially in modem, low voltage IC design. Traditional source followers also have the limitation that they are class A type amplifiers, with a maximum load current limited to the quiescent current in the buffer. Having a type A amplifier with large quiescent current is inherently energy inefficient.  
         [0004]     Previous attempts to increase the load drive capability of low voltage source followers have focused on sensing the current in the source follower drain and folding it back to increase the effective transconductance of the device. These prior art high bandwidth, low voltage gain cells include, for instance, the circuits described in U.S. Patent Application Publication US 2002/0175761 A1 titled “High-Bandwidth Low-Voltage Gain Cell and Voltage Follower Having an Enhanced Transconductance” by Bach et al, the contents of which are hereby incorporated by reference.  
         [0005]     These circuits achieve the required low voltage operation with high bandwidth and high transconductance, but operate essentially in a class A type mode, drawing significant current even when there is no input signal. Moreover, such circuits are limited in drive capability in one direction by the quiescent current.  
         [0006]     To improve power consumption, there is a need for a high bandwidth, high transconductance source follower circuit that operates in a class AB mode, in which only a small current is drawn in the absence of an input signal and in which the drive current is not limited by the quiescent current in either the sink or source direction.  
       SUMMARY OF THE INVENTION  
       [0007]     The current invention is a low voltage, high bandwidth, enhanced transconductance source follower circuit constructed from Metal Oxide Silicon Field Effect Transistors (MOSFET) devices, which operates in a class AB mode. The class AB mode operation allows significantly more than the quiescent current to be available for both sourcing and sinking the output, resulting in significantly reduced average power consumption when driving the same load. Moreover, both the enhanced transconductance and the class AB mode operation of this invention are achieved while maintaining the large bandwidth of traditional source follower designs. The circuit of this invention can be used in a variety of applications, including voltage regulators and current conveyors  
         [0008]     The high bandwidth, high conductance class AB source follower circuit of this invention uses a folded cascode device to effectively sense the drain current of the source follower device. The folded cascode sensing device feeds its source current to the gate of a common source device of the same type (NMOS or PMOS) as the source follower. The connections are such that this results in a current that is effectively a multiple of the sensed current being directed to the output load. Over-limit current load at the source follower drain is sensed by a common source device of the opposite type (NMOS or PMOS) and any necessary extra current from the over-limit sensor is added to the output load.  
         [0009]     In one embodiment of the invention, a current source is gate connected to the drain of a current sensing device of the same type (PMOS or NMOS) such that sensing the need for increased current drives more current from the current source. This allows a drive current greater than the quiescent current. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0010]      FIG. 1  is a schematic drawing of a type A enhanced transconductance source follower.  
         [0011]      FIG. 2  is a schematic drawing of a type AB enhanced transconductance source follower in accordance with the present invention, implemented as a modification to the type A circuit of  FIG. 1 .  
         [0012]      FIG. 3  is a schematic drawing of an NMOS type AB enhanced transconductance source follower with additional speed up transistors, in accordance with the present invention.  
         [0013]      FIG. 4  is a schematic drawing of an NMOS source follower with class AB output in accordance with the present invention, operating as a current conveyor with replica output devices into a cascode device with a load.  
         [0014]      FIG. 5  is a schematic drawing of a PMOS source follower with class AB output in accordance with the present invention, operating as a current conveyor with replica output devices into a cascode device with a load.  
         [0015]      FIG. 6  is a schematic drawing of a class AB source follower in accordance with the present invention, used to buffer a regulator output and provide high current drive with low output impedance and low quiescent power.  
     
    
     DETAILED DESCRIPTION  
       [0016]     During the course of this description like numbers will be used to identify like elements according to the different views that illustrate the invention.  
         [0017]      FIG. 1  is a schematic drawing of an enhanced transconductance source follower having class A behavior. Class A type amplifiers are limited to a maximum current output equal to the quiescent current, which makes their average power consumption very high. The class A type circuit of  FIG. 1  provides a reference point for understanding the implementation and functioning of class AB circuits in accordance with this invention, as shown in  FIGS. 2-6 .  
         [0018]     The circuit of  FIG. 1  includes an NMOS source follower, transistor  12 , having an input signal applied to its gate, or control terminal, at  13 , and its source, a current flow terminal, is coupled to the output at  16 . The circuit also has a current source, or supply, comprising the PMOS transistor  18 . The drain, another current flow terminal, of PMOS  18  is connected to the rail voltage  20 , and its gate is biased to a fixed voltage at  28 . In this configuration, PMOS  18  acts as a current supply providing an essentially fixed current i 1 . The current i 2  flowing into the drain of source follower NMOS  12  is sensed by the folded-cascode PMOS transistor  22 . The sensing function of PMOS  22  results from constant current i 1  being the supply both to the drain of NMOS  12  and the source of PMOS  22 . This results in PMOS  22  drain current i 3  effectively being the difference between current i 2  an i 1 . The drain of PMOS  22  is connected to the gate of NMOS  24 , with diode connected NMOS  26  functioning essentially as a biasing diode. The drain of NMOS  24  is connected to the source of the source follower, NMOS  12 .  
         [0019]     The operation of the type A enhanced transconductance source follower shown in  FIG. 1  can be qualitatively understood by considering the currents that flow in response to a small change in signal voltage. When there is no signal voltage applied at  14 , quiescent currents flow through all the transistors. In the quiescent state, bias voltages at  14 ,  28  and  30  are chosen such that the current i 2  flowing into NMOS  12  essentially matches the current i 5  flowing into the drain of NMOS  24 . This tends to minimize the quiescent current i 4  flowing to the load and, with a sufficiently high impedance load, the quiescent value of current i 4  is essentially zero. When an input signal is applied at  14 , the results is in an increase from the quiescent bias voltage at  14 , the current flowing from drain to source in NMOS  12  increases and consequently current i 2  increases. Because source current i 1  is essentially fixed by the fixed bias voltage at  28 , the increase in the drain current i 2  of NMOS  12 &#39;s is sensed by PMOS  22  as a decrease in it&#39;s drain current i 3  This decrease in current i 3  results in a lower voltage drop across diode connected NMOS  26 , and consequently a decrease in the gate voltage on NMOS  24 . This decreased gate voltage results in a lowering of the drain to source current across NMOS, which means that current i 5  decreases. So the net result of increase in signal voltage at  14 , is an increase in current i 2  and a decrease in current i 5 . The result is that the increase in the current i 4  flowing out to the load at  16  is not merely a function of the increase in current i 2 . Instead it is a function of the combination of the increase in i 2  plus the difference between the quiescent value of i 5  and the now decreased value of i 5 . This results in increased transconductance, that is increased change in current supplied to the output for a given change in signal voltage. However, in the circuit of  FIG. 1 , the maximum current that can be supplied to the output is limited by the fixed current i 1  flowing through PMOS  18 . Moreover, that fixed current i 1  is also the quiescent current flowing through the circuit.  
         [0020]      FIG. 2  is a schematic drawing of a first embodiment of a type AB enhanced transconductance source follower of this invention, implemented essentially as a modification to the type A enhanced transconductance source follower of  FIG. 1 . The class AB operation means that more that quiescent current is available for both sourcing and sinking the output, resulting in significantly reduced average power consumption in driving the same load. A significant difference between the class A amplifier of  FIG. 1  and the class AB type amplifier of  FIG. 2  is that in the circuit of  FIG. 2 , the gate of the current source PMOS  18  is tied directly to the gate of PMOS  22  by the conducting connecter  38 . This allows the same cascode voltage at  30  to be applied to the gates of both PMOS  18  and PMOS  22 . The result of this modification may be understood quantitatively by considering how currents in the circuit change from their quiescent values in response to a signal voltage applied at  14 .  
         [0021]     In a quiescent state, with no signal voltage at input  14 , current i 1  is still essentially equal to the sum of currents i 2  and i 3 . Furthermore, components and bias voltages are chosen so that in the quiescent state, current i 5  is set essentially equal to current i 2  so that current i 4  directed to a high impedance load is essentially zero. When the signal voltage at  14  increases, the drain to source current i 2  increases. This is sensed by folded cascode connected PMOS  22  as a decreased current i 3 . A reduced current i 3  also means a reduced voltage to the gate of common source NMOS  24  because of the reduced current through diode connected NMOS  26 . This in turn results in a reduced current i 5 , giving improved transconductance as in the circuit of  FIG. 1 . However, the gate of the current source PMOS  18  is now tied to the gate of the current sensing PMOS  22 . As the current in PMOS  22  decreases, the voltage difference between the gate of PMOS  18  and the rail voltage at  20  increases, and produces an increase in current i 1 . This increase in current i 1  may be seen as a consequence of the voltage at  30  being essentially the sum of the voltage across diode connected NMOS  26  and the gate to drain voltage of PMOS  22 . A decrease in current i 3  effectively decreases both of these voltages and results in an increased voltage difference between point  30  and the fixed upper rail  20 . This means that the maximum load current of the circuit is not limited to the quiescent value of current i 1 . In the circuit of  FIG. 2 , the maximum load current is now essentially equal to the maximum current that can be carried by PMOS  18  when fully turned on. The direct link  38  of the gate of current source PMOS  18  to the gate of current sensing PMOS  22  effectively gives the source follower NMOS  12  type AB amplifier characteristics. The ability to have a significantly lower quiescent current to enable driving the same load current means that the average power consumption of the circuit can be considerably reduced.  
         [0022]     Cascode NMOS  33  is optionally included to more closely control conductance by effectively having two transistors  33  and  24  in parallel with the output load and having a fixed bias voltage applied to transistor  33 .  
         [0023]     In a further embodiment of this invention, shown in  FIG. 3 , the frequency response of the amplifier output can be further improved by placing impedances at the gates of the transistors  18  and  26 . Having impedances at their gates also effectively makes the two transistors  18  and  26  current mirrors. In particular, short circuit  38  of  FIG. 2  may be replaced by a PMOS transistor  19  with source connected to the gate of PMOS  18 , drain connected to the drain of PMOS  18 , gate connected to the gate of PMOS  22  and the tub or body connected to voltage point  20 . The effect of this additional transistor  19  includes speeding up the response of PMOS  18  when a change in current is detected by PMOS  22 . A further aspect of the embodiment of  FIG. 3  is an additional PMOS transistor  25 . This additional transistor  25  adds capacitance to the base of PMOS  26 . This is done as shown in  FIG. 3  by having the drain of PMOS  25  connected to the gate of PMOS  26 , while the tub or body of PMOS  25  is connected to ground. The source of PMOS  25  is connected to the drain of NMOS  22 . The gate of PMOS  25  is biased to the same voltage as cascode transistor  33 . The net result of theses connections is an improvement in the high frequency response of the amplifier output circuit as compared to the circuit of  FIG. 2 .  
         [0024]      FIG. 4  is a schematic drawing of an NMOS source follower with class AB output operating as a current conveyor with replica output devices into a cascode device with a load.  
         [0025]     The five transistors  40 ,  46 ,  50 ,  52  and  54  of this circuit function in a manner that is broadly analogous to the behavior of the five transistor class A amplifier having enhanced transconductance diagramed in  FIG. 1 . The functioning of transistors  40 ,  46 ,  50 ,  52  and  54  may be understood at a qualitative level in the same manner. NMOS  40  is connected as a source follower with a signal input and quiescent offset voltage being applied to its gate at  42 . The source and tub of NMOS  40  are connected to output  44 . PMOS  46  acts essentially as a current source, or supply, in a manner analogous to PMOS  18  in  FIG. 1 . A constant bias voltage at  48  is applied to the gate of PMOS  46 , resulting in a constant current following from source to drain. The drain of PMOS  46  is connected to both the drain of source follower NMOS  40  and to the source of current sensing PMOS  50 , which is connected in a folded cascode manner. The drain of PMOS  50  is connected to the source of diode connected NMOS  52  and to the gate of common source NMOS  54 . As before, when the signal input at  42  is zero, quiescent currents are such that the current steered toward output  44  is essentially zero. When the signal voltage increases, the current through NMOS  40  increases, which is sensed by PMOS  50 . This results in a reduction in current flowing from the drain of PMOS  50  that is a function of the increase in current flowing through NMOS  40 . This decrease in current is effectively multiplied by common source NMOS  54 . The combined increase current from NMOS  40  and the related but multiplied decrease in current flowing across NMOS  54 , are both steered toward output  44 , in a similar fashion to the circuit of  FIG. 1 . The difference between the circuit of  FIG. 3  and that of  FIG. 1  is, that when the input signal at  42  increases to the point where all of the current from current source PMOS  46  is flowing through source follower NMOS  40  the circuit does not “max out”. Instead, there is an additional common source transistor, PMOS  56 , which acts effectively both as an over limit current sensor and as an auxiliarly current source or supply. PMOS  56  is turned off in the quiescent state, but when PMOS  50  turns off, because of an over limit current condition in which essentially all the current supplied by PMOS  46  is flowing through NMOS  40 , the reduced voltage drop across diode connected PMOS  52  turns auxiliary current source, or supply, PMOS  56  on. The current flowing through PMOS  56  is steered towards the output  44 , adding the necessary additional current to correspond to the increased input signal at  42 . The addition of PMOS  56  effectively allows the source follower NMOS  40  to act in a class AB manner in the supply or push mode. In addition, if common source NMOS  54  has a larger current capacity than current source PMOS  46 , the circuit functions essentially in a class AB manner in the sink or pull mode. The net result is that the six transistor  40 ,  46 ,  50 ,  52 ,  54  and  56  portion of the circuit of  FIG. 4  can supply a total drive current significantly greater than the quiescent current in the circuit. This can be seen from the fact that the quiescent current is effectively the fixed current flowing through current source PMOS  46 , whereas the available drive current includes the additional current that can be supplied through PMOS  56 , which is turned off in the quiescent state.  
         [0026]     If PMOS transistors  46  and  56  are matched, twice the quiescent current can flow, significantly reducing average power consumption and battery requirements.  
         [0027]     The remaining components in the circuit of  FIG. 4 , including PMOS  58 , PMOS  68 , NMOS  60  and NMOS  62  form a current conveyor, providing a mirrored current to output  64  in addition to the voltage following output at  44 . In the output at  64 , the current is a precise conversion of the input voltage signal. In contrast, at the output at  44 , the voltage is a precise conversion of the input signal voltage.  
         [0028]      FIG. 5  is a schematic drawing of an embodiment of the present invention in which a PMOS source follower with class AB output operates as a current conveyor with replica output devices into a cascode device with a load.  
         [0029]     In  FIG. 6 , transistors  70 ,  74 ,  76 ,  78 ,  80 ,  82  and  84  form the class AB source follower. PMOS  70  is connected as a source follower, with signal and quiescent bias voltage supplied at  72 , and output to  74 . NMOS  74  acts as a current source or supply. NMOS  76  acts as a current sensing device, sensing a change in the drain current of source follower PMOS  70  when the input signal at connector  72  changes. The current change sensed by NMOS  76  is multiplied by common source NMOS  78  and steered towards the output at  74 . Common source NMOS  80  is turned off in the quiescent state, and acts both as an over current limit sensing device and as a second, auxiliary current supply. When all available current supplied by current source PMOS  74  is steered through the source follower  70 , NMOS  80  turns on and steers the necessary extra current directly to the output connector  74 , providing class AB operation.  
         [0030]     Transistors  86 ,  88 ,  90  and  90  operate as a current conveyor providing a replica current-to-current output  94 .  
         [0031]      FIG. 6  is a schematic drawing of a class AB source follower, in accordance with the present invention, used to buffer a regulator output and provide high current drive with low output impedance and low quiescent power.  
         [0032]     PMOS  100  acts as a source follower with signal and quiescent voltage supplied to its gate at  102 .  
         [0033]     The remainder of the circuit in  FIG. 6  acts so that a regulator voltage is buffered and a high current drive is supplied with a low impedance output at  110 .  
         [0034]     While the invention has been described with reference to the preferred embodiment thereof, it will be appreciated by those of ordinary skill in the art that modifications can be made to the structure and elements of the invention without departing from the spirit and scope of the invention as a whole.

Technology Category: 5