Patent Document

BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates generally to voltage references. In particular, the present invention relates to low temperature coefficient low power programmable complementary metal oxide semiconductor (CMOS) voltage references with simple and efficient architecture. 
     2. Description of the Related Art 
     Historically, the design of voltage references entailed the use of circuits containing hundreds of transistors. Due to this, the complexity and costs involved in designing such voltage references can be significantly high. Bandgap voltage reference circuits are most commonly used in CMOS mixed signal circuits to provide for a stable voltage reference. A standard bandgap reference such as, for example, LM330 3-Terminal Positive Regulator from National Semiconductor Corporation, has a variation of approximately 0.2% in output voltage over a temperature range from room temperature to 80° C. Moreover, as voltage references, erasable programmable read-only memories (EPROMs) are designed to support leakage on the order of 1 volt over a ten (10) year lifetime. 
     However, the bandgap circuit has a number of shortcomings. Some of these shortcomings include large dc power consumption, the need for parasitic, and usually unsupported, pnp or npn transistors, and the need to match the base emitter voltage (V BE ) for these parasitic devices. Additionally, several design iterations are often necessary to get the bandgap circuit to operate due to lack of good models for the second order effects upon which the bandgap performance is based. Moreover, the stability of the bandgap references over time has been a reliability issue for some CMOS products, for example, in telecommunications devices. 
     SUMMARY OF THE INVENTION 
     In view of the foregoing, the present invention discloses a low temperature coefficient low power programmable CMOS voltage reference. 
     In particular, an apparatus including a voltage reference in accordance with one embodiment of the present invention includes a reference terminal; an output terminal; a first transistor having first and second terminals; a second transistor having first and second terminals, said second transistor first terminal coupled to said output terminal; a capacitance coupled between said first transistor first terminal and said reference terminal configured to store a pre-determined reference signal; and a variable signal generator coupled to said second transistor first terminal configured to provide a variable signal and in accordance thereto provide an output signal to said output terminal; wherein said variable signal is configured to vary such that said output signal is substantially equal in magnitude to said predetermined reference signal. 
     An apparatus including a voltage reference in accordance with another embodiment of the present invention includes a reference terminal; a first transistor having first, second and third terminals; a second transistor having second and third terminals; a capacitance coupled between said first transistor first terminal and said reference terminal, said capacitance configured to store a reference signal; and a programming signal source coupled to said first and second transistor third terminals configured to provide a programming signal thereto, and in accordance therewith, generates said reference signal across said capacitance. 
     An apparatus including a voltage reference in accordance with yet another embodiment of the present invention includes a reference terminal; an output terminal; a first transistor having first, second and third terminals; a second transistor having first, second and third terminals, said second transistor first terminal coupled to said output terminal; a capacitance coupled between said first transistor first terminal and said reference terminal configured to store a reference signal; a programming signal source coupled to said first and second transistor third terminals configured to provide a programming signal thereto, and in accordance therewith, generating said reference signal across said capacitance; and a variable signal generator coupled to said second transistor first terminal configured to provide a variable signal and in accordance thereto provide an output signal to said output terminal; wherein said variable signal is configured to vary such that said output signal is substantially equal in magnitude to said reference signal. 
     Accordingly, the present invention provides an architecturally simple and efficient voltage reference. These and other features and advantages of the present invention will be understood upon consideration of the following detailed description of the invention and the accompanying drawings. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 shows a conceptual illustration of the voltage reference in accordance with one embodiment of the present invention. 
     FIG. 2 illustrates the voltage reference circuit in accordance with one embodiment of the the present invention. 
     FIG. 3 illustrates an implementation layout of the voltage reference in accordance with one embodiment of the present invention. 
     FIG. 4 illustrates a cross-coupled transistor implementation layout of the voltage reference in accordance with one embodiment of the present invention. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENT 
     FIG. 1 shows a conceptual illustration of the voltage reference in accordance with one embodiment of the present invention. As shown, there are provided a MOS input operational amplifier  101  having an input terminal  102  and an output terminal  104 . Coupled to the input terminal  102  of the amplifier  101  is a reference capacitor  103  which is further coupled to a reference terminal such as a ground terminal GND. In this manner, a reference voltage V REF  stored in the reference capacitor  103  is buffered by the MOS amplifier  101  and provided to the output terminal  104  of the amplifier  101 . Since the reference capacitor  103  itself does not draw any current, in can be seen that the overall architecture requires less power to operate. 
     FIG. 2 illustrates a capacitive voltage reference in accordance with one embodiment of the present invention. As shown, there are provided two transistors  202 ,  203  in a differential amplifier configuration whose drain terminals are coupled at node  208 . Also coupled to node  208  is a reference signal source  207  configured to provide a reference signal thereto. The gate terminal of transistor  202  is coupled to a reference capacitor  201  at node  209 . The reference capacitor  201  is further coupled to a reference terminal which can be a ground terminal GND. 
     The gate terminal of transistor  203  is coupled to a variable voltage source  204  at node  212  to provide variable voltages thereto. The variable voltage source  204  coupled to node  212  can be variably adjusted as discussed in further detail below so that the voltages at the nodes  210  and  211  are matched. In this matched state, the differential amplifier (transistors  202  and  203 ) has substantially the same current at nodes  210  and  211  thereby providing a balanced differential amplifier. When the differential amplifier is balanced, node  212  is matched with node  209  such that any voltage variation at node  209  is reflected as voltage variation at node  212 . Accordingly, the reference voltage pre-programmed in the reference capacitor  201  can be sampled at node  212  without substantially draining the stored voltage across the reference capacitor  201  and thereby avoiding potential leakage. In this manner, by measuring or sampling the voltage level at node  212 , it is possible to determine the voltage level at node  209  representing the level of the voltage reference stored in the reference capacitor  201 . 
     Referring to FIGS. 1 and 2, in accordance with one embodiment of the present invention, a programming signal can be directly applied to the gate dielectric of the transistor  203  to program the desired reference voltage for the voltage reference circuit. In particular, a voltage can be applied, for example, from the reference signal source  207  to node  208  to induce a current flow known as Fowler/Nordheim tunneling current across the source-gate barrier of transistor  202 ; i.e., between the source and gate terminals of transistor  202 . The application of the voltage at node  208 , in turn, varies the charge level on node  209  such that the top plate  201   a  of the reference capacitor  201  is more positively charged as compared to the bottom plate  201   b  of the reference capacitor  201 , thereby programming a voltage at node  209  with respect to the reference terminal (such as the ground terminal GND) coupled to the bottom plate  201   b  of the reference capacitor  201 . It is to be noted that the actual voltage level at node  209  depends largely upon the intended application for the voltage reference. Naturally, using the approach described above, the programmable voltage level can be modified as necessary. 
     It is also to be noted that programming of the initial reference voltage across the reference capacitor  201  to configure the voltage reference for use as a voltage reference as described above can also be achieved by lowering the voltage of the bottom plate  201   b  of the reference capacitor  201 . In specific, a voltage across the gate dielectric of transistor  202  can be applied which will change the total amount of charge on node  209 . In other words, if node  208  is made relatively negative with respect to node  209 , electrons will be injected to node  209  to make node  209  relatively more negative, and conversely, if node  208  is made relatively positive with respect to node  209 , electrons will be pulled out of node  209 , thus rendering node  209  relatively more positive with respect to node  208 . For example, approximately 10 Mvolts/cm will be required to induce current flow across the gate dielectric of transistor  202 . In the case of a 100 angstrom oxide (of transistor  202 ), 10 volts will be required to pull current across the gate dielectric of transistor  202 . 
     Further shown in FIG. 2 are resistor  205  and variable resistor  206  which are coupled to the corresponding source terminals of the transistor pair  202 ,  203  at nodes  210  and  211 , respectively. During the initial programming stage of the voltage reference in accordance with one embodiment of the present invention, the variable resistor  206  coupled to the source terminal of transistor  203  at node  211  is variably adjusted off-chip at room temperature to match resistor  205  which is coupled to the source terminal of transistor  202  at node  210 . Additionally, the variable voltage source  204  coupled to the gate terminal of transistor  203  at node  212  is likewise adjusted to minimize any offset between the two source terminals of transistors  202  and  203  such that the voltage at node  212  substantially equals the voltage at node  209  which couples the reference capacitor  201  to the gate terminal of transistor  202 , and also, to minimize any offsets between the two transistors  202  and  203 . 
     Then, the temperature of the chip is increased and the variable voltage source  204  is further adjusted to minimize offsets between the source terminals of the two transistors  202 ,  203  at nodes  210  and  211 , respectively. Accordingly, the variation in the variable voltage source  204  applied to minimize offsets between the source terminals of the transistors  202 ,  203  coupled at nodes  210  and  211 , respectively, in response to the temperature increase in the chip then corresponds to the sum of the temperature coefficient T C  of the reference capacitor  201  and any component offsets between transistors  202  and  203  which has a dependence upon variations in temperature. For example, as the on-chip junction temperature changes, the temperature-dependent variations in the transistor pair  202 ,  203  will rise and fall together such that the signal at node  212  will track the signal at node  209  irrespective of the chip temperature. 
     Accordingly, by variably adjusting the voltage with the variable voltage source  204  at node  212 , any variation in the on-chip temperature will have the same effect, if any, upon both transistors  202 ,  203  because the differential amplifier is in a substantially balanced condition. Moreover, the amount of charge loss across the reference capacitor  201  can be determined by comparing the initial temperature coefficient T C  of the reference capacitor  201  with a subsequent measurement of the same after a predetermined exposure to a temperature of, for example, 250° C. 
     The reference voltage programming across reference capacitor  201  as described above, however, may disturb the symmetry of the differential amplifier. More specifically, because the programming operation is fairly stressful upon the gate dielectrics of transistors  202  and  203 , this operation may cause a shift in the symmetry between the two transistors  202 ,  203 . Thus, to regain symmetry between transistor  202  and transistor  203  of the differential amplifier, a dummy programming can be used. 
     For example, one approach to providing a dummy programming includes providing a same current to both transistors  202  and  203  to equilibrate them and then proceed with the programming of the transistor  202  as described above. The dummy programming is performed as the final test stage and can be used to program any analog voltage if a feedback is applied during programming. As discussed previously, the voltage at node  208  is initially raised to a predetermined level sufficient to provide a current through the gate dielectric of transistor  202  (the forcing function). Due to symmetry obtained between the two transistors  202  and  203 , the raised voltage at node  208  provides the current to both transistors  202  and  203 , and by measuring the voltage at node  212 , the programming voltage can be determined. Accordingly, the shift in the transistors  202  and  203  remain relatively consistent with one another thereby preserving the symmetry therebetween. 
     For example, in the case of a 2-volt voltage reference with an initial voltage at node  209  of 1 volt (which can be determined by measuring the voltage at node  212 ), a voltage of 5 volts, for example, is applied by the reference signal source  207  at node  208  which is sufficient to provide a current through the gate dielectric of transistor  202 . By measuring the voltage at node  212 , the voltage increase in node  209  from the initial 1 volt, for example, to a higher level such as 1.5 volts, can determined. Since this measured value is still below the desired reference voltage of 2 volts, the same procedure as outlined above is repeated until the voltage at node  209  is raised to the desired voltage for the voltage reference, in this instance, 2 volts. 
     Thus, as can be seen, by increasing the voltage across the gate dielectric of transistor  202  until it conducts current, the long term threshold voltage V T  offset drift of the matched transistor pair  202 ,  203  can be significantly improved. Moreover, in accordance with the present invention, the programming time would not pose a significant problem as is the case with EPROMs since there is only one capacitor to program, and since this programming step would take the place of trimming which would normally be applied to precision voltage references. 
     FIG. 3 illustrates an implementation layout of the CMOS voltage reference circuit in accordance with one embodiment of the present invention. As shown, nodes  301 ,  302  and  303  correspond to nodes  208 ,  210 , and  211 , respectively, of the voltage reference circuit illustrated in FIG.  2 . In particular, node  301  in FIG. 3 corresponds to the node at which the drain terminals of the two transistors  202 ,  203  (FIG. 2) are coupled, and also coupled thereto is the reference signal source  207  (FIG.  2 ). Further, nodes  302  and  303  respectively correspond nodes  210  and  211  of the embodiment shown in FIG. 2 where the source terminals of the transistor pair  202 ,  203  in the differential amplifier are respectively coupled. 
     Further shown in FIG. 3 is capacitor C 1  which corresponds to the reference capacitor  201  of FIG.  2 . The top plate  304  and the bottom plate  305  of the capacitor C 1  correspond respectively to the top plate  201   a  and bottom plate  201   b  of the reference capacitor  201  of FIG.  2 . The top plate  304  of the capacitor C 1  includes a floating polysilicon layer in order to achieve minimal temperature coefficient T C . Moreover, as can be seen, the top plate  304  of capacitor C 1  is extended over the edge of the polysilicon layer to form first transistor  307  which corresponds to transistor  202  of the voltage reference as illustrated in FIG.  2 . Thus, it can be seen that node  209  coupling the reference capacitor  201  to the gate terminal of transistor  202  in FIG. 2 corresponds to node  310  corresponding to the contact with the area on the top plate  304  as shown in FIG.  3 . 
     Likewise, second transistor  308  corresponding to transistor  203  in the voltage reference circuit as shown in FIG. 2 is formed by extending the upper polysilicon layer of a second capacitor C 2  as shown in FIG.  3 . In this manner, the tracking gate at node  309  which is connected to the top plate  306  of the second capacitor C 2  corresponds to node  212  of FIG. 2 which is coupled to the gate terminal of transistor  203  and the variable voltage source  204  (FIG.  2 ). Moreover, as can be seen, second transistor  308  is connected to the second capacitor C 2  to maintain symmetry with first transistor  307  up to the contact mask. The connection to the second capacitor C 2  further contributes to improving the matching of first and second transistors  307 ,  308 . Additionally, a dummy bottom plate  311  is provided to the second capacitor C 2  to further improve matching between transistors  307  and  308 . 
     Furthermore, if a floating node such as node  310  is FIG. 3 is connected to any kind of metal interconnections, the retention capability of the floating node is drastically reduced primarily due to the fact that the metal interconnections generally protrude through the dielectric layers and thus are not as well protected by the passivation layers that are normally deposited. For example, after the second polysilicon layer is deposited, a layer containing high concentration of phosphorus is deposited such that the phosphorus minimizes any sodium ions contamination which may be introduced during the process steps. Metal interconnections coupled to the floating node  310  will protrude through the layer of high concentration phosphorus and become exposed to sodium ions. Therefore, using a double layer of polysilicon as shown in the present invention rather than a single polysilicon layer maximizes the retention time of the floating nodes where the floating nodes are coupled to transistors and the like via metal interconnections. 
     Moreover, in accordance with the present invention, the area of the floating polysilicon layer for both capacitors C 1  and C 2  which does not cover the corresponding lower polysilicon layer is kept to a minimum to preserve a good temperature coefficient T C . 
     One issue with respect to the operating conditions of the voltage reference in accordance with the present invention is how well transistor  307  and transistor  308  are matched. This is significant since transistor  308  is essentially configured to replicate the operation of transistor  307 , and thus, any offset in the matching of transistors  307  and  308  will offset the operability of the voltage reference circuit itself. 
     To provide improved matching of transistors  307  and  308 , a cross coupled transistor in accordance with one embodiment of the present invention is provided as illustrated in FIG.  4 . In particular, the first and second transistors  307 ,  308  of FIG. 3 (which corresponds to transistors  202 ,  203  comprising the differential amplifier as illustrated in FIG. 2) are provided in a cross coupled configuration in the implementation layout. It is to be noted that like elements shown in FIG. 4 which correspond to the counterpart elements in FIG. 3 are labelled as such. The implementation layout of transistors  307  and  308  of FIG. 3 is now provided in a cross-coupled manner as shown in FIG. 4 such that the gradient across the wafer cancels thus providing a more stable transistor mismatch over time. 
     Accordingly, the cross-coupled transistor embodiment as shown in FIG. 4 provides improved transistor matching, which results in improved offset drift over time, and therefore, the voltage reference circuit incorporating the cross-coupled transistor configuration will be more stable. On the other hand, such cross-coupled configuration would require more area of the floating polysilicon layer which is not situated over the lower polysilicon layer. Consequently, this cross-coupling transistor configuration may have potential adverse effect on the temperature coefficient T C . Nevertheless, the cross-coupled transistor configuration shown in FIG. 4 provides a more stable reference voltage circuit. 
     As illustrated above, in accordance with the present invention, the reference voltage is stored over the reference capacitor with low leakage such as an EPROM capacitor, for example, a few millivolts over the product&#39;s life. The stored voltage on an EPROM capacitor is then buffered through a pMOS input voltage follower to provide a reference voltage with 8 ppm/° C. 
     Since a poly-to-poly capacitor has a temperature coefficient T C  of 8 parts per million (ppm) over a temperature range of 60° C., the temperature-dependent variation of the reference capacitor would be 480 ppm or 0.05% over the temperature range. Thus, a reference voltage stored in such poly-to-poly capacitor would only change by 0.05% over the given temperature range of 60° C. 
     As compared with the temperature coefficient T C  of bandgap references such as the LM330 3-Terminal Positive Regulator, the poly-to-poly capacitor has a temperature coefficient which is four times better. Furthermore, since the only transistor that the stored charge node in the reference capacitor is connected to is a pMOS transistor (instead of a nMOS transistor as is the case with EPROMs), there is a much lower probability of hole channel current passing through the gate oxide in the pMOS transistor than electron channel current in the NMOS transistor. Additionally, since the area of the reference capacitor in accordance with the present invention is many times that of an EPROM capacitor (in the order of picoFarads rather than femtoFarads) such that the area-to-perimeter ratio is much higher, the effects of perimeter-related leakage such as migration of ions are significantly reduced. Moreover, in accordance with the present invention, only one capacitor is conditioned to not leak, whereas EPROMs contain millions of cells which must all be operational. Therefore, some guard banding is built into the EPROM relaxation requirement of 1 volt drop in the 10 year lifetime period of a product. In others words, EPROMs must allow for worst case whereas the present invention can assume typical leakage. 
     Further, the voltage reference circuit according to the present invention need only be programmed once, not multiple times as is the case for most EPROMs. This is an advantage since multiple programming increases leakage in the gate dielectric through which the reference voltage programming is completed. Moreover, the voltage reference circuit shown in the present invention is also fully static during operation, thus eliminating many kinds of read disturb seen in EPROMs. Also, voltage across the source-gate and drain-gate oxide of the first transistor (transistor  202  in FIG. 2, for example) can be designed to be very low. 
     The electric field in the reference capacitor in accordance with the present invention is relatively low (approximately one volt) compared to several volts in EPROM. In turn, tunneling leakage decreases very rapidly with reduction in voltage. However, the voltage reference circuit of the present invention can operate with a low signal voltage because the capacitance is relatively large compared to an EPROM such that the signal which is proportional to the stored charge in the reference capacitor can be large even if the voltage is small. Also, since the area of the reference capacitor is not a constraint as in the case with the design of an EPROM cell, novel layout techniques may be used to improve leakage from the floating gate. For example, the guard bands around the voltage reference circuit may be used to provide a barrier for lateral charge migration. 
     As is known, EPROM floating gates are programmed negative with electron injection. Thus, sodium (Na) and other positive ions are attracted to the gate. This is one of the limiting mechanisms for EPROM lifetime. By contrast, the present invention allows the floating gate to be programmed positive which will repel ionic contamination. Finally, the voltage reference cell of the present invention allows for a very small active gate area compared to the capacitor area (e.g., 100:1). With most EPROMs, the ratio of the active gate area to storage capacitor areas is on the order of one to one (1:1). 
     Various other modifications and alterations in the structure and method of operation of this invention will be apparent to those skilled in the art without departing from the scope and spirit of the invention. Although the invention has been described in connection with specific preferred embodiments, it should be understood that the invention as claimed should not be unduly limited to such specific embodiments. It is intended that the following claims define the scope of the present invention and that structures and methods within the scope of these claims and their equivalents be covered thereby.

Technology Category: g