Patent Document

RELATED APPLICATION 
     This application is a continuation of U.S. application Ser. No. 14/718,597, filed on May 21, 2015, entitled “CHARGE PUMP CIRCUIT FOR A PHASE LOCKED LOOP”, the contents of which are hereby incorporated by reference in their entirety. 
    
    
     TECHNICAL FIELD 
     This disclosure is related to the field of phase locked loops, and, more particularly, to a charge pump circuit for phase locked loops. 
     BACKGROUND 
     A phase locked loop (PLL) is a control system that generates an output signal whose phase is related to the phase of an input signal. A typical phase locked loop includes a variable frequency oscillator and a phase detector. The oscillator generates a periodic signal. The phase detector compares the phase of the input signal with the phase of the periodic signal and generates control signals that adjust the oscillator to keep the phases matched. 
     Keeping the input and output phases locked also implies keeping the input and output frequencies the same. Consequently, in addition to synchronizing phases between signals, a phase locked loop can track an input frequency, or it can generate a frequency that is a multiple of the input frequency. 
     Such phase locked loops are widely employed in radio, telecommunications, computers and other electronic applications. They can be used to demodulate a signal, recover a signal from a noisy communication channel, generate a stable frequency at multiples of an input frequency (frequency synthesis), or distribute precisely timed clock pulses in digital logic circuits such as microprocessors. Since a single integrated circuit can provide a complete phase locked loop building block, phase locked loops are widely used in modern electronic devices, with output frequencies from a fraction of a hertz up to many gigahertz. 
     In some cases, it may be desirable for a phase locked loop to be operable over a wide band of frequencies. In order to produce such wide band phase locked loops, a charge pump circuit is typically employed in the loop to generate the control signals sent to the oscillator. However, such charge pump circuits may be noisy, resulting in an undesirable amount of in-band noise. 
     Therefore, new phase locked loop designs with new charge pump circuits are desirable. 
     SUMMARY 
     This summary is provided to introduce a selection of concepts that are further described below in the detailed description. This summary is not intended to identify key or essential features of the claimed subject matter, nor is it intended to be used as an aid in limiting the scope of the claimed subject matter. 
     Disclosed herein is a circuit including a phase frequency detector (PFD) configured to compare phases of an input signal and a feedback signal, and to generate first and second control signals as a function of that comparison. An attenuation circuit includes a capacitor coupled in series between a node and a switching node, and is configured to charge the capacitor and disconnect the switching node from ground based on assertion of the first control signal, and discharge the capacitor and connect the switching node to ground based on assertion of the second control signal. 
     A phase locked loop includes a phase frequency detector (PFD) configured to compare phases of an input signal and a feedback signal, and to generate therefrom control signals. A an attenuation circuit is coupled in series with the PFD and includes first and second current sources, and a loop filter coupled between a voltage controlled oscillator (VCO) control node and a ground node. An amplifier has an input coupled to the VCO control node. An impedance network is coupled to the VCO control node and includes at least one impedance element configured to be coupled to the first current source such that voltage at the VCO control node increases, based upon the control signals indicating that the phase of the input signal leads the phase of the feedback signal, and coupled to the second current source such that the voltage at the VCO control node decreases, based upon the control signals indicating that the phase of the feedback signal leads the phase of the input signal. A VCO is coupled to the VCO control node and to generate an output signal based upon a signal at the VCO control node, with the phase of the output signal matching the phase of the input signal. The feedback signal is based upon the output signal. 
     Another aspect is directed to a circuit including a first current source coupled between a power supply node and a first node, and a first switch coupled between the first node and a second node and controlled by a first control signal. A second switch is coupled between the second node and a third node and controlled by a second control signal. A second current source is coupled between the third node and a ground node. A third switch is coupled between the first node and an output node and controlled by a complement of the first control signal. A fourth switch is coupled between the second node and the output node and controlled by a third control signal. A fifth switch is coupled between the second node and a fourth node and controlled by the third control signal. A first capacitor coupled between the second node and the fourth node, and a second capacitor coupled between the second node and ground. A sixth switch is coupled between the fourth node and a fifth node and controlled by an inverse of the third control signal. A loop filter coupled between the fifth node and ground. An amplifier has a non-inverting terminal coupled to the fifth node, an inverting terminal coupled to the output node, and an output terminal coupled to the output node. A seventh switch coupled between the output node and the third node and controlled by an inverse of the second control signal. 
     A further aspect is directed to a circuit including a first current source coupled between a power supply node and a first node, and a first switch coupled between the first node and a second node and controlled by a first control signal. A second switch is coupled between the second node and a third node and is controlled by a second control signal. A second current source is coupled between the third node and a ground node. A first resistor is coupled between the second node and a fifth node. A second resistor is coupled between the second node and a fourth node. A third switch is coupled between the fourth node and a sixth node and controlled by a third control signal. A loop filter is coupled between the fifth node and ground. An amplifier has a non-inverting terminal coupled to the fifth node, an inverting terminal coupled to the sixth node, and an output terminal coupled to the sixth node. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a block diagram of a phase locked loop. 
         FIG. 2  is a schematic diagram of a charge pump circuit such as may be used in the phase locked loop of  FIG. 1 . 
         FIG. 2A  is a block diagram of a circuit that may be used to generate the control signals H and HB for the charge pump circuit of  FIG. 2 . 
         FIG. 3  is a schematic diagram of a loop filter such as may be used in the phase locked loops of  FIGS. 1 and 4 . 
         FIG. 4  is a schematic diagram of another charge pump circuit such as may be used in the phase locked loop of  FIG. 1 . 
         FIG. 5  is a schematic diagram of a loop filter such as may be used in the phase locked loop of  FIG. 4 . 
         FIG. 6  is a graph showing output current noise for the charge pump circuits described herein vs. conventional charge pump circuits. 
         FIG. 7  is another graph showing output current noise for the charge pump circuits described herein vs. conventional charge pump circuits. 
         FIG. 8  is a schematic diagram of another phase locked loop such as may employ the charge pump circuits of  FIGS. 2 and 4 . 
         FIG. 9  is a schematic diagram of another charge pump circuit such as may be used in the phase locked loop of  FIG. 8 . 
         FIG. 10  is a block diagram of a logic circuit to generate the selection signal for the charge pump circuit of  FIG. 9 . 
     
    
    
     DETAILED DESCRIPTION 
     One or more embodiments will be described below. These described embodiments are only examples of implementation techniques, as defined solely by the attached claims. Additionally, in an effort to provide a focused description, irrelevant features of an actual implementation may not be described in the specification. 
     With reference to  FIG. 1 , a phase locked loop  100  is now described. The phase locked loop  100  includes a phase frequency detector (PFD)  110 , which receives an input signal Fin having an input frequency, and an output signal Fout having an output frequency. The output signal Fout received by the phase frequency detector  110  is the output signal Fout of the phase locked loop  100 . The phase frequency detector  110  has outputs UP, DN coupled to a charge pump  200  or  300 , also referred to as an attenuation circuit. The charge pump  200  or  300  in turn has an output coupled to a loop filter Z, which is in turn coupled to a voltage controlled oscillator (VCO)  120 . The output of the VCO  120  is coupled to the input of the phase frequency detector  110  via an optional divider  130 . 
     In operation, the phase frequency detector  110  compares the input signal Fin to the output signal Fout, and generates the control signals UP, DN for the charge pump  200  or  300  based thereupon. When the phase of the input signal Fin leads the phase of the output signal Fout, the control signal UP is asserted at a logic high, while the control signal DN remains at a logic low. Conversely, then when the phase is the input signal Fin lags the phase of the output signal Fout, the control signal DN is asserted at a logic high, while the control signal UP remains at a logic low. When the phase of the input signal Fin and the phase of the output signal Fout match, neither UP nor DN are asserted. 
     The charge pump  200  or  300  generates a control signal for the VCO  120 , which is passed through the loop filter Z, which extracts the low frequency content of the control signal. The VCO  120 , based on the control signal, adjusts the phase and frequency of the output signal Fout. When UP is asserted, the charge pump  200  or  300  increases the voltage of the control signal, as opposed to decreasing the voltage of the control signal when DN is asserted. Those of skill in the art will appreciate that since the phase of the input signal Fin cannot both lead and lag the phase of the output signal Fout, the phase frequency detector  110  will not simultaneously assert both UP and DN. 
     An optional divider  130  may be included in the feedback loop coupling the output signal Fout to the phase frequency detector  110 . The divider  130  serves to divide the frequency of the output signal Fout, thereby causing the frequency of the output signal Fout to be generated by the VCO  120  as a multiple of the frequency of the input signal Fin. For example, if the divider  130  divides the frequency by 2, in order for the phase frequency detector  110  to see that the input signal Fin and the feedback signal (the output signal Fout after being fed through the divider  130 ) have a same frequency, the output signal Fout would have a frequency twice that of the input signal Fin. If the divider  130  is not present, or if the divider divides by 1, then the frequency of the output signal Fout will match the frequency of the input signal Fin. 
     Details of the charge pump  200  and loop filter Z will now be given with reference to  FIGS. 2-3 . The charge pump  200  includes a first current source  202  coupled between a power supply node Vcc and a node  204 . Switch S 1  is coupled between node  204  and node  206 . Switch S 2  is coupled between node  206  and node  208 . A second current source  210  is coupled between node  208  and ground. Switch S 3  is coupled between node  204  and node  218 . Switch S 4  is coupled between node  206  and node  218 . Switch S 7  is coupled between node  218  and node  208 . 
     A first capacitor Cs is coupled between node  206  and node  212 , and switch S 5  is coupled in parallel with the first capacitor Cs between node  206  and node  212 . A second capacitor Cs 2  is coupled between node  206  and ground GND. Switch S 6  is coupled between nodes  212  and  214 , and the loop filter Z is coupled between node  214  and ground. In addition, the non-inverting terminal of an amplifier  216  is coupled to node  214 , while the inverting terminal and output terminal of the amplifier  216  is coupled to the node  218 . The capacitors Cs and Cs 2  have a capacitance value less than a capacitance value of impedance elements used in the loop filter Z. The value of Cs 2  differs from that of Cs by a factor of one less than a desired gain A of the charge pump circuit  100 . That is, the value of Cs 2  is Cs*(A−1). 
     The loop filter Z, details of which are shown in  FIG. 3 , includes a resistor R 1  and capacitor C 1  coupled in series between node  214  and ground. A capacitor C 2  is coupled between node  214  and ground, and a resistor R 2  and capacitor C 3  are coupled in series between node  214  and ground. 
     In operation, switch S 1  is triggered in response to assertion of UP, while switch S 2  is triggered in response to assertion of DN. Switch S 3  is triggered in response to assertion of a complement of UP, noted as NUP, while switch S 7  is triggered in response to assertion of a complement of DN, noted as NDN. Switch S 6  is triggered in response to assertion of a signal representing a logical NAND operation between the complement of UP and the complement of DN HB (shown in  FIG. 2A ), while switches S 4  and S 5  are triggered in response to assertion of a signal H which is a complement of that signal. 
     Thus, when the phase of the input signal Fin leads the phase of the output signal Fout, the phase frequency detector  110  asserts UP while keeping DN low. The switches S 1 , S 6 , and S 7  are closed and the other switches opened, resulting in the flow of current from the first current source  202  through nodes  204  and  206  into the second capacitor Cs. This serves to charge up the second capacitor Cs with a voltage seen at node  214 . The amplifier  216  has a unity gain, and thus passes the voltage seen at node  214  to its output at node  218 . The control signal for the VCO  120  is output from node  214 . 
     On the other hand, when the phase of the input signal Fin lags the phase of the output signal Fout, the phase frequency detector  110  asserts DN while keeping UP low. The switches S 2 , S 3 , and S 6  are thus closed and the other switches opened, resulting in the sinking of current from node  206 , and thus the discharge of the voltage at the second capacitor Cs. Therefore, the voltage at node  214  falls, which the amplifier  216  passes to its output at node  218 . The control signal for the VCO  120  is output from node  214 . 
     Where the phase of the input signal Fin is matched to the phase of the output signal Fout, the phase frequency detector  110  asserts neither UP nor DN. Thus, switches S 3 , S 4 , S 5 , and S 7  close, while the other switches remain open. This serves to pass the current from the first current source  202  through the node  204 , into node  218 , into node  208 , and to ground GND through the second current source  210 . 
     The charge pump circuit  200  described above provide a variety of advantages over traditional charge pump circuits. For example, the charge pump circuit  200  uses a charge-pump current  202  and  210  that is higher by a factor of A, but preserves the overall PLL loop gain by an attenuation factor of 1/A which is achieved via capacitive division. This is illustrated in  FIGS. 6-7 . Shown in  FIGS. 8A-8C  is how noise suppression increases as A increases. In addition, the thermal noise in the charge pump circuit  200  from the current sources  202  and  210  is reduced by a factor of A. Amplifier noise feedthrough to the loop filter Z is proportional to Cs*Vamp*Fin, where Fin is the input frequency to the PLL and where Vamp is the voltage at the non-inverting terminal of the amplifier  216 , and should be less than the noise from the current sources  202  and  210 . Thus, for the same loop gain in the charge pump circuit  200 , the noise entering the loop filter Z is reduced. This also serves to reduce the in-band phase noise. The reduction in output noise over conventional charge pump circuits is on the order of 1/A and can be seen in  FIGS. 6-7 . 
     An alternate design for the charge pump circuit  300  is now described with reference to  FIG. 4 . The charge pump circuit  300  includes a first current source  302  coupled between the power supply node Vcc and node  304 , and a switch S 1  coupled between the node  304  and a node  306 . A switch S 2  is coupled between the node  306  and a node  308 . A second current source  310  is coupled between the node  308  and ground GND. A resistor R 3  is coupled between the node  306  and a node  312 , and the loop filter Z is coupled between the node  312  and ground GND. A resistor R 4  is coupled between the node  306  and a node  314 , through switch S 3 . A amplifier  316  has its non-inverting terminal coupled to node  312 , and its inverting terminal and its output coupled to the node  314 . The values of the resistor of the attenuation filter Z is high. A switch S 4  is coupled between node  304  and node  314 . Node  314  is coupled to node  311 . 
     The resistance of the resistor R 3  may equal (A−1)*R 4 , while the resistance of R 4  is chosen to reduce the noise contribution from the resistive attenuation network and make its noise contribution less than that of current sources  302  and  310 . To do so, R 4 &gt;A/Gm, where Gm is the transconductance of the current sources  302  and  310 . This causes 1/A of the current from the current sources  302 ,  310  to flow across R 3  and into the attenuation filter Z. The current sources  302 ,  310  conduct A times more current than conventional charge pump current sources, thus the transconductance of the current sources  302 ,  310  can be A times more than that of conventional charge pump current sources. In addition, when the resistors R 3  and R 4  have large values, the noise from the amplifier  316  that enters the attenuation filter Z is reduced. 
     In operation, switch S 1  is triggered in response to assertion of UP, while switch S 2  is triggered in response to assertion of DN. Switch S 3  is triggered in response to assertion of a logical NAND operation between complements of UP and DN, denoted as HB, while switch S 4  is triggered in response to a complement of assertion of UP and switch S 5  is triggered in response to a complement of assertion of DN. 
     Therefore, when the phase of the input signal Fin leads the phase of the output signal Fout, the phase frequency detector  110  asserts UP while keeping DN low. Switch S 1 , S 3 , and S 5  are then closed while switch S 2  and S 4  are open, resulting in the flow of current from the first current source  302  through node  306 , into the resistor R 3 , and into node  312 , thereby generating a voltage across the resistor R 3 , which is seen by the non-inverting terminal of the amplifier  316  at node  312 , which passes the voltage at node  312  to its output at node  314 . The control signal for the VCO  120  is output from node  312 . 
     When the phase of the input signal Fin lags the phase of the output signal Fout, the phase frequency detector  110  asserts DN while keeping UP low. The switches S 2 , S 3 , and S 4  close while the switches S 1  and S 5  open, resulting in the sinking of current from node  306 . Therefore, the voltage at node  312 , and thus the voltage of the control signal for the VCO  120 , falls. 
     When the phase of the input signal Fin matches the phase of the output signal Fout, the phase frequency detector  110  asserts neither UP nor DN. Thus, switches S 4 , S 5  are closed, while switches S 1 , S 2 , S 3  remain open. This serves to couple output of the amplifier  316  to the non-inverting terminal of the amplifier  316  and to ground, lowering the voltage at node  312 , and thus the voltage of the control signal for the VCO  120 . 
     The loop filter Z of  FIG. 5  is usable with the charge pump circuit  300 , and comprises a resistor R coupled in series with a capacitor C. When the loop filter Z is employed, the value of the resistor R 3  differs from that of the resistor R 4  by a factor of one less than a desired gain A of the charge pump circuit  300 . That is, the value of R 3  is R 4 *(A−1). 
     The charge pump circuit  300  has the same advantages as the charge pump circuit  200  described above. As stated, the charge pump circuit  300  offers an increased gain over conventional charge pumps by a factor of A, yet reduces the loop gain within the charge pump circuit  300  by a factor of 1/A, so the overall loop gain for the phase locked loop  100  is preserved. In addition, the thermal current noise in the charge pump circuit  300  is increased by a factor of A or √{square root over (A)}, but is attenuated by 
             1     A   2           
when entering the loop filter Z. The noise feed through from the amplifier  316  to the loop filter Z is proportional to
 
               Vamp     A   *   R   ⁢           ⁢   4       .         
Thus, for the same loop gain in the charge pump circuit  300 , the noise entering the loop filter Z is reduced.
 
     An embodiment where the phase locked loop  100  employs one of the charge pump circuits  200 ,  300  described above as well as an additional charge pump circuit  400  is now described with reference to  FIG. 8 . The phase locked loop  100  operates as the phase locked loop of  FIG. 1 , however the additional charge pump circuit  400  is coupled in series between the PFD  110  and the loop filter Z before the phase locked loop  100  locks, while one of the charge pump circuits  200 ,  300  is coupled in series between the PFD  110  and the loop filter Z after the phase locked loop  100  locks. The purpose of this selection between charge pump circuits  200 ,  300  or  400  is so as to assist quick locking of the phase locked loop  100  while still receiving the advantages of the charge pump circuits  200 ,  300  as described above. It should be noted that if the current output by the charge pump  400  is I, then the current output by the charge pump circuits  200 ,  300  would be I*A. Selection of the charge pump circuit  200 ,  300  or  400  is based upon a selection signal. 
     As shown in  FIG. 10 , a selection signal LOCK is generated based on a lock detector detecting whether or not the phase locked loop  100  has locked, by comparing the input frequency Fin to the feedback signal. An inverse of this selection signal ENH is used to enable the charge pump circuit  400 , while an inverse of that signal ENL is used to enable the charge pump circuits  200 ,  300 . 
     The charge pump circuit  400 , as shown in  FIG. 9 , includes a first current source  402  coupled between a power supply node and node  404 . A first switch S 1  is coupled between node  404  and node  406 . An amplifier  416  has a non-inverting terminal coupled to node  406 . A loop filter Z is coupled between node  406  and ground. The inverting terminal of the amplifier  416  is coupled to its output at node  414  so as to bias the amplifier  416  in a unity gain mode. 
     A switch S 2  is coupled between node  406  and node  408 . A second current source  410  is coupled between node  408  and ground. A switch S 3  is coupled between node  404  and node  414 , while a switch S 4  is coupled between node  414  and node  408 . In operation, switch S 1  is actuated by assertion of UP, while switch S 2  is actuated by assertion of DN. Switch S 3  is actuated by an inverse of UP, NUP, while switch S 4  is actuated by an inverse of DN, NDN. 
     When the phase of the input signal Fin leads the phase of the output signal Fout, the phase frequency detector  110  asserts UP while keeping DN low. The switches S 1 , S 4  are closed and the other switches opened, resulting in the flow of current from the first current source  402  through nodes  404  and  406  into the loop filter Z and the non-inverting terminal of the amplifier  416 , thereby increasing the voltage seen at the non-inverting terminal. Due to the unity gain of the amplifier  416 . The voltage seen at node  406  is passed to its output at node  414 . The control signal for the VCO  120  is at node  406 . 
     When the phase of the input signal Fin lags the phase of the output signal Fout, the phase frequency detector  110  asserts DN while keeping UP low. The switches S 2 , S 3  are thus closed and the other switches opened, resulting in the sinking of current from node  406 . Therefore, the voltage at node  406  falls, which the amplifier  416  passes to its output at node  414 . The control signal for the VCO  120  is at node  406 . 
     Where the phase of the input signal Fin is matched to the phase of the output signal Fout, the phase frequency detector  110  asserts neither UP nor DN. Thus, switches S 3 , S 4  while the other switches remain open. This serves to pass the current from the first current source  402  through the node  404 , into node  414 , into node  408 , and to ground GND through the second current source  410 . 
     It should be understood that any of the loop filters Z described herein may be used with any of the embodiments described herein, and that other types of loop filters (i.e. active loop filters utilizing operational amplifiers) are also usable with any of the embodiments described herein. 
     While the disclosure has been described with respect to a limited number of embodiments, those skilled in the art, having benefit of this disclosure, will appreciate that other embodiments can be envisioned that do not depart from the scope of the disclosure as disclosed herein. Accordingly, the scope of the disclosure shall be limited only by the attached claims.

Technology Category: 5