Patent Document

FIELD OF THE INVENTION 
     The present invention relates to a thyristor switched current source circuit with compensation for power supply voltage and ambient temperature variations. It can be used to provide constant current to LED and other loads. 
     WORD AND SYMBOL EXPLANATIONS 
     Whenever the following words or symbols are encountered in this document their true meaning would be as stated below. 
     LED: Light emitting diode. 
     Load: Electrical load 
     LED load: One LED or more LEDs in series or parallel connections that would allow the LEDs when activated to be forward biased. 
     deg. C.: Degree or degrees Celsius. 
     mA: Milliampere or milliamperes. The milliampere is a unit of current equal to one thousandth of an ampere. 
     mV: Millivolt or millivolts. The millivolt is a unit of voltage equal to one thousandth of a volt. 
     kohm: Kiloohm or kiloohms. The kiloohm is a unit of electrical resistance equal to one thousand ohms. 
     BACKGROUND OF THE INVENTION 
     In an LED driver circuit it is essential for the LED current to be controlled or kept constant over the expected range of power supply voltage variations and ambient temperature changes. The simplest method to somewhat stabilize the current of an LED would be to use a resistor in series connection with an LED load. But this method has serious drawbacks. In order for the resistor to act more like a current source its value must be large. This would keep the power dissipation on the resistor low but would limit the number of LEDs that can be driven with one resistor. If a low value resistor is used more LEDs can be accommodated but at higher temperatures or supply voltages the power dissipation on the resistor may turn out to be excessive. Hence, with the resistor methodology there is not really current control and at high ambient temperatures or high line voltage conditions the change in the LED load current can shorten the life of the LEDs and can even result in thermal run-away, poor illumination, degradation and sometimes even a total damage to the LEDs. 
     There is a large number of circuits in prior art designed to stabilize the LED current for power supply voltage and/or ambient temperature variations. These circuits are driven from low AC voltage sources that are rectified with bridge rectifiers before being converted to DC. In order to obtain a pure DC voltage and eliminate rectifier bridge ripple voltage, high quality smoothing capacitors must be used. Such capacitors are bulky, expensive and have a short lifetime as well. Circuits for power factor correction (PFC) are also required. Variable duty cycle control circuits may be included as well most likely for controlling LED illumination by means of pulse width modulation (PWM) techniques that control the LED load current. Also in order for these circuits to provide a constant current for the LED load over a wide DC power supply voltage range, linear regulators or the switched type of regulators like boost or buck converters are used. 
     BRIEF SUMMARY OF THE INVENTION 
     The present invention is a thyristor switched current source designed to provide constant current to LED and other loads. The thyristor switch can be an SCR or a TRIAC or an equivalent circuit for an SCR or a TRIAC. This circuit, by utilizing a triggered thyristor, is driven from an unsmoothed rectified AC voltage source. Without a high quality smoothing electrolytic capacitor the LED load voltage is a 120 Hz rectified AC voltage waveform. This type of a voltage is actually more desirable than a pure DC by virtue of the fact that LED lifetime is increased if the LED is not fully on constantly. Furthermore, the LED load voltage waveform is sinusoidal and in phase with the AC supply voltage waveform. Hence, no coils and expensive electrolytic capacitors or any other hardware for power factor correction (PFC) is necessary. There is also no problem with the LED illumination exhibiting flicker. With the LED current being switched at a rate of 120 Hz and in the absence of any pulse width modulation (PWM) the LED illumination would appear continuous to the human eye. It should also be noted here that thyristor circuits, as is the case with other high frequency switching LED circuits, generate electromagnetic interference (EMI). However if the thyristor circuit in this invention is driven from the secondary side of a transformer, as should be the case with low voltage applications, the conductive part of EMI can be prevented from getting to the primary side of the transformer and to other electrical lines. As for the radiated part of EMI none was detected in the most critical AM band of radio with the thyristor circuit driven from the secondary side of a transformer. 
     In the present invention the LED or any other load is driven from a power device such as a MOSFET, a JFET or a transistor. The power device is biased for activation from a resistive string which essentially constitutes the load of the thyristor. Now if the thyristor is set to trigger only at a predetermined voltage point on the AC supply voltage waveform the LED load current can remain fairly constant with power supply voltage fluctuations or loading. In other words, with power supply voltage variations the bias on the resistive string also varies by the amount needed for the current in the power device or the LED load to remain constant. The bias change in the resistive string with power supply voltage variations is attributed to shifts in the thyristor trigger angle. The thyristor performs still another function by making possible that the bias change with temperature in the resistive string enables the LED load current to remain constant with ambient temperature variations as well. Without the thyristor compensation for changes in the LED load current for AC supply voltage and ambient temperature variations could not have been achieved with conventional biasing schemes. Most likely two different sets of hardware would have been required. The type of temperature compensation here would be for the current in the power device to either remain constant or change as required. This can be done by using in the trigger circuit of the thyristor a zener diode or another electrical component with positive or negative temperature coefficient. For example, a component with negative temperature coefficient can cause the current in the power device to start falling off at higher temperatures a feature which with an LED load can be desirable for preventing any deterioration in the life and illumination of the LEDs. Temperature compensation, as is the case with the AC supply voltage variations, is attributed to shifts in the thyristor trigger angle except that in this case the trigger angle of the thyristor is not confined to a specific point on the AC supply voltage waveform. However, any shift in the triggering point on the AC supply voltage waveform would be minor and would not affect the proper operation with AC supply voltage variations. 
     Therefore, a general object of this invention is to provide a current source circuit for LED and other loads where the load current must stay constant with power supply voltage fluctuations. 
     Another object of this invention is to provide a current source circuit for LED and other loads where the load current must stay constant with power supply loading. 
     Another object of this invention is to provide a current source circuit for LED and other loads where the load current must stay constant with ambient temperature variations. 
     Another object of this invention is to provide a thyristor voltage source for biasing external power devices driving LED and other loads that require for the load current to stay constant with power supply voltage and ambient temperature variations. 
     Yet another object of this invention is to provide a current source circuit for LED and other loads where the load current is allowed to vary with ambient temperature variations. 
     Still another object of this invention is to maximize the useful life of the LED by insuring that the LED current stays within the allowable limits with power supply voltage and temperature variations. 
     A further object of this invention is to provide a constant current source driver circuit which is durable, versatile and cost effective. This circuit is composed of discrete components but with the bill of materials being very short and in the absence of large capacitors and coils the manufacturing cost would be comparable to a circuit with an integrated driver. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  shows a schematic diagram of a prior art P-CHNNEL MOSFET LED driver with the LED current staying constant with power supply voltage variations or loading. 
         FIG. 2  shows a schematic diagram of a prior art P-CHNNEL MOSFET LED driver with the LED current staying constant with ambient temperature changes. 
         FIG. 3  shows a schematic diagram of a thyristor switched P-CHNNEL MOSFET current source LED driver according to at least one embodiment of the present invent ion. 
         FIG. 4  shows a schematic diagram of a thyristor switched N-CHNNEL MOSFET current source LED driver according to still another embodiment of the present invention. 
         FIGS. 5A, 5B and 5C  show half-cycle of rectified AC voltage versus time waveforms of the load and thyristor triggering for a P-CHNNEL MOSFET current source LED driver. 
         FIGS. 6A, 6B and 6C  show half-cycle of rectified AC voltage versus time waveforms of the load and thyristor triggering for an N-CHNNEL MOSFET current source LED driver. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Shown in  FIG. 1  is a prior art current source driver for LED and other loads. The load current in this circuit is independent of supply voltage changes but not for ambient temperature variations. In the circuit both transistors are the P-CHANNEL MOSFET type with source, gate and drain terminals as depicted. These transistors can also be substituted with the PNP type of transistors with emitter, base and collector terminals being the MOSFET source, gate and drain terminals respectively. AC supply Vcc 1  is the positive side of a DC voltage source or the positive side of an unsmoothed rectified AC voltage source. The negative side of voltage source Vcc 1  is connected to ground GND 1 . The source terminal of transistor  100  is connected to supply Vcc 1  by way of resistor  109 . The drain terminal of transistor  100  is returned to ground GND 1  via load LED 1 . Resistor  107  is connecting the gate terminal of transistor  100  to node  111  on the bias string comprising diode  101 , diode connected transistor  103  and resistor  105 . The gate-to-source voltage of transistor  100  is compensated with the gate-to-source voltage of transistor  103 . Hence, the load LED 1  current is primarily set with the forward voltage of diode  101  appearing across resistor  109 . Now since the diode  101  forward voltage hardly changes with supply Vcc 1  voltage variations or loading the load LED 1  current stays constant. With ambient temperature changes the gate-to-source voltage variation of transistor  100  can easily be compensated with a similar variation of the gate-to-source voltage of transistor  103 . This is fine but the variation with temperature of the forward voltage of diode  101 , responsible for setting the load LED 1  current, is a problem. If diode  101  is bipolar with a temperature coefficient of −2 mV/deg. C. the load LED 1  current can vary by as much as 30% with a change of 100 deg. C. in ambient temperature. Furthermore and in order to minimize the voltage drop across resistor  109  if diode  101  is a low forward voltage Schottky diode with a temperature coefficient of −1.2 mV/deg. C. the load LED 1  current can vary by as much as 50% for a 100 deg. C. change in ambient temperature. 
     Shown in  FIG. 2  is a prior art current source driver for LED and other loads. The load current in this circuit is independent of ambient temperature variations but not for supply voltage fluctuations. In the circuit both transistors are the P-CHANNEL MOSFET type with source, gate and drain terminals as shown. These transistors can also be substituted with the PNP type of transistors with emitter, base and collector terminals being the MOSFET source, gate and drain terminals respectively. AC supply Vcc 2  is the positive side of a DC voltage source or the positive side of an unsmoothed rectified AC voltage source. The negative side of voltage source Vcc 2  is connected to ground GND 2 . The source terminal of transistor  200  is connected to Vcc 2  by way of resistor  209 . The drain terminal of transistor  200  is returned to ground GND 2  via load LED 2 . Resistor  207  is connecting the gate terminal of transistor  200  to node  211  on the bias string comprising diode  201 , diode connected transistor  203  and resistor  205 . The gate-to-source voltage of transistor  200  is compensated with the gate-to-source voltage of transistor  203 . Hence, the load LED 2  current is set with the voltage across resistor  201  also appearing across resistor  209 . In this circuit the load LED 2  current is independent of ambient temperature variations. The reason is that transistors  200  and  203  by being properly matched have similar temperature characteristics. Any variation with ambient temperature in the gate-to-source voltage of transistor  200  would be compensated with a similar variation in the gate-to-source voltage of transistor  203  in such a way that the bias at node  211  would not be affected. However, compensation for changes in the load LED 2  current with supply Vcc 2  voltage variations or loading is not expected to be the case in this circuit. Any change in the supply Vcc 2  voltage would affect the voltage across resistor  209  and hence the load LED 2  current. For a 20% change in the supply Vcc 2  voltage there would be a 30% change in the load LED 2  current. 
     According to at least one embodiment of the present invention depicted in  FIG. 3  is a schematic diagram of a thyristor switched current source driver for LED and other loads. In this circuit the load current is independent of supply voltage and ambient temperature variations. The circuit is driven by an unsmoothed rectified AC voltage source having positive and negative sides connected to supply Vcc 3  and ground GND 3  terminals respectively. Thyristor  310  having input, output and control terminals is driven from supply Vcc 3  shown connected to thyristor  310  input terminal. The output terminal of thyristor  310  is returned to ground GND 3  via resistors  311  and  313  in series connection. Thyristor  310  control terminal is connected to supply Vcc 3  by way of a trigger circuit comprising a series combination of resistor  305  and zener diode  303  operable in reverse or breakdown mode. Included in the trigger circuit of thyristor  310  is capacitor  301  connected between the thyristor  310  control terminal and ground GND 3 . Transistor  300  is a P-CHNNEL MOSFET having source, gate and drain terminals as depicted in a conventional diagram for a MOSFET. Transistor  300  can also be a PNP transistor with emitter, base and collector terminals being the MOSFET source, gate and drain terminals respectively. The source terminal of transistor  300  is connected to supply Vcc 3  by way of resistor  309 . The transistor  300  drain terminal is returned to ground GND 3  via load LED 3 . Resistor  307  is connecting the gate terminal of transistor  300  to the voltage divider common point of resistors  311  and  313 . The reason for this divider is to obtain the correct bias for the transistor  300  gate terminal without allowing a large initial voltage drop across thyristor  310 . Another reason is that by utilizing resistor  311  dimming can be introduced in the LEDs of load LED 3 . By adjusting lower resistor  311  the bias at the gate terminal of transistor  300  is increased and this would cause the voltage across resistor  309  and the load LED 3  current to be lowered. Another way of introducing dimming in the LEDs of load LED 3  is by adjusting lower resistor  305 . This time the voltage across thyristor  310  is reduced and this would cause the bias at the gate and source terminals of transistor  300  to go higher and the load LED 3  current to go lower. 
     Half-cycle rectified AC oscilloscope voltage versus time waveforms for thyristor  310  triggering and load LED 3  are shown in  FIGS. 5A, 5B and 5C  for supply Vcc 3  voltage changes form high to nominal to low respectively. Any change in the supply Vcc 3  voltage would affect the voltage across resistor  309  and hence the load LED 3  current. What actually happens is that a change in supply Vcc 3  voltage, as a result of shifts in the thyristor  310  trigger angle, would affect the voltage across thyristor  310  as well as the bias on the divider of resistors  311  and  313  and through resistor  307  the bias of the transistor  300  gate and source terminals. As shown in the thyristor  310  voltage waveforms of  FIGS. 5A, 5B and 5C , triggering on the AC supply Vcc 3  voltage waveform occurs at the same voltage point VP 1  which, for example, may be taken to be about 15 volts. By properly selecting the triggering point VP 1  on the supply Vcc 3  voltage waveform and keeping it constant for supply Vcc 3  voltage variations, the correct bias is obtained at the gate and source terminals of transistor  300  to prevent the load LED 3  current from changing. Thyristor  310  by controlling the voltage on the bias string of resistors  311  and  313  enables the load LED 3  current to stay constant for ambient temperature variations as well. This would not have been the case with conventional biasing schemes. Transistor  300  gate terminal is returned to ground GND 3  by way of resistors  307  and  313 . This would enable transistor  300  to be operable even before any triggering of thyristor  310  as shown in  FIGS. 5A, 5B and 5C  load LED 3  voltage waveforms. Also indicated in  FIGS. 5A, 5B and 5C  is that all load LED 3  voltage waveforms are of the same amplitude, sinusoidal and in phase with the supply Vcc 3  voltage waveforms. Capacitor  301  can be omitted if thyristor  310  triggering is not needed beyond 90 electrical degrees. As an example, in  FIGS. 5A, 5B and 5C  thyristor  310  may be looked at as triggering at about 40, 50, and 60 electrical degrees respectively. For sensitive thyristor  310  the control terminal input current is low and being the only current through resistor  305  requires for resistor  305  have a large value. It is actually more desirable for resistor  305  to have a lower value in which case triggering desensitization of thyristor  310  can be employed. This is done by connecting a small value resistor between the control and output terminals of thyristor  310 . 
     Temperature compensation must be carried out for variations in the load LED 3  current as well as the voltage in the thyristor  310  load of resistors  311  and  313 . This compensation can be carried out by utilizing the positive temperature coefficient of the operable in reverse or breakdown mode zener diode  303  voltage to offset temperature related changes due to negative temperature coefficients of the thyristor  310  control terminal input current and control terminal to output terminal voltage. The change in the thyristor  310  control terminal input current is manifested as a voltage change across resistor  305 . Compensation must also be carried out for the transistor  300  source-to-gate voltage variation with temperature. This again can be done by utilizing the variations with temperature of the zener diode  303  voltage and the thyristor  310  control terminal input current. However with MOSFET devices the temperature coefficient of the gate-to-source voltage can be both positive and negative and cancel each other out at some value of drain current. Therefore, if transistor  300  is set to operate at the zero temperature coefficient point for the gate-to-source voltage then temperature compensation would reestablish the selected triggering point VP 1  on the AC supply Vcc 3  voltage waveform and the load LED 3  current would be constant for ambient temperature changes as well as supply Vcc 3  voltage variations. In the event of having variation with temperature in the gate-to-source voltage of transistor  300  temperature compensation can still be achieved but with a shift in the triggering point VP 1  on the AC supply Vcc 3  voltage waveform. However, if the components used are selected with the proper electrical characteristics, the shift with temperature of the triggering point VP 1  on the AC supply Vcc 3  voltage waveform, would have only a minor effect on the change of the load LED 3  current with supply Vcc 3  voltage variations. Also the triggering point VP 1  on the AC supply Vcc 3  voltage waveform can always be adjusted to accommodate the variation with temperature of the transistor  300  gate-to-source voltage. With a sensitive thyristor  310  the control terminal input current is very low and being the only current through resistor  305  requires for the value of resistor  305  to be large. Actually in this case and for less troublesome temperature compensation it would be better if the value of resistor  305  is not very large. One way of doing this would be to desensitize the triggering of thyristor  310 . This can be done by connecting a small value resistor between the thyristor  310  control terminal and output terminal. The result would be additional current in resistor  305  and this would require for resistor  305  to have a lower value. For example, a resistor of 1 kohm would increase the current in resistor  305  by 0.65 mA if the thyristor  310  control terminal to output terminal voltage is 0.65 volts. 
     Excessive shifts with temperature of the predetermined triggering point VP 1  on the AC supply Vcc 3  voltage waveform can still be realized if a large change in the load LED 3  current is required. For example, for the triggering point VP 1  to shift lower for reduction in the load LED 3  current at higher temperatures may require for the zener diode  303  voltage to be lower or the zener diode  303  be replaced with a negative temperature coefficient resistor. 
     According to another embodiment the circuit depicted in  FIG. 4  is the N-CHNNEL MOSFET version of the circuit in  FIG. 3 . It is a thyristor switched current source driver for LED and other loads. In this circuit the load current is also independent of supply voltage and ambient temperature variations. The circuit is driven by an unsmoothed rectified AC voltage source having positive and negative sides connected to supply Vcc 4  and ground GND 4  terminals respectively. Thyristor  410  having input, output and control terminals is driven from supply Vcc 4  shown connected to thyristor  410  input terminal by way of resistor  413 . The output terminal of thyristor  410  is returned to ground GND 4  via resistor  411 . Thyristor  410  control terminal is connected to supply Vcc 4  by way of a trigger circuit comprising a series combination of resistor  405  and zener diode  403  operable in reverse or breakdown mode. Included in the trigger circuit of thyristor  410  is capacitor  401  connected between the thyristor  410  control terminal and ground GND 4 . The reason for the thyristor load of resistors  411  and  413  being split up is for the gate terminal of transistor  400  to have the proper bias without a large voltage drop across thyristor  410 . This is no different form the way it is done in the circuit of  FIG. 3  where the bias at the gate terminal of transistor  300  is set with the voltage drop across thyristor  310  and resistor  311 . Another reason is that with resistor  411  dimming can be introduces in the LEDs of load LED 4 . By adjusting lower resistor  411  the bias at the gate terminal of transistor  400  is lowered and this would cause the voltage across resistor  409  and the load LED 4  current to be lowered. Another way of introducing dimming in the LEDs of load LED 4  is by adjusting lower resistor  405 . This time it is the reduction of the voltage across thyristor  410  that would cause the bias at the gate terminal of transistor  400  to go lower. Transistor  400  is an N-CHNNEL MOSFET having source, gate and drain terminals as depicted in a conventional diagram for this type of a MOSFET. Transistor  400  can be an NPN transistor with emitter, base and collector terminals being the MOSFET source, gate and drain terminals respectively. Transistor  400  can also be an integrated base bipolar transistor (IGBT) with emitter, base and collector terminals being the MOSFET source, gate and drain terminals respectively. The source terminal of transistor  400  is connected to ground GND 4  by way of resistor  409 . The transistor  400  drain terminal is returned to supply Vcc 4  via load LED 4 . Resistor  407  is connecting the gate terminal of transistor  400  to thyristor  410  input terminal. The reason for that is for the gate of transistor  400  to be biased from supply Vcc 4  through resistors  407  and  413  in order for transistor  400  to be operable even before any triggering of thyristor  410 . 
     The operation of the circuit in  FIG. 4  is similar to the P-CHANNEL MOSFET circuit of  FIG. 3  in spite of the fact that some components are rearranged with respect to supply Vcc 4  and ground GND 4 . The load LED 4  current is set with resistor  409  which this time is returning the source terminal of transistor  400  to ground GND 4 . Load LED 4  is now connected between supply Vcc 4  and the drain terminal of transistor  400 . In this circuit the load LED 4  voltage waveform is referenced to supply Vcc 4 . It is also sinusoidal and in phase with the AC supply Vcc 4  voltage waveform. Again by properly selecting a triggering point VP 2  on the supply Vcc 4  voltage waveform and keeping it constant with supply Vcc 4  voltage variations the correct bias is obtained at the gate and source terminals of transistor  400  to prevent the load LED 4  current from changing. It is actually the variation of the voltage across thyristor  410  that enables the biasing scheme used to work properly, that is, to prevent the load LED 4  current from changing not only for supply Vcc 4  voltage fluctuations but also for ambient temperature variations. Capacitor  401  can be omitted if no thyristor  410  triggering is needed beyond 90 electrical degrees. Half-cycle rectified AC oscilloscope voltage versus time waveforms for thyristor  410  triggering and load LED 4  are shown in  FIGS. 6A, 6B and 6C  for supply Vcc 4  voltage changes form high to nominal to low respectively. 
     Compensation for temperature related changes in the current of load LED 4  and the voltage in the thyristor  410  load of resistors  411  and  413  can be carried out the same way as with the circuit of  FIG. 3 . This compensation can be done by utilizing the positive temperature coefficient of the operable in reverse or breakdown mode zener diode  403  voltage to offset temperature related changes in the thyristor  410  control terminal input current and control terminal to output terminal voltage and the transistor  400  gate-to-source voltage. The change in the thyristor  410  control terminal input current can be looked at as voltage change across resistor  405 . Resistors with positive or negative temperature coefficients can also be utilized in place of zener diode  403  to obtain large changes in the load LED 4  current. Also with a sensitive thyristor  410  having low control terminal input current it may be necessary to desensitize the triggering of thyristor  410  in order to increase the current through resistor  405 . This can be done by connecting a small value resistor between the control terminal and output terminal of thyristor  410 . The result would be a higher current through resistor  405  which would enable resistor  405  to have a lower value and make compensation for temperature related changes in the load LED 4  current easy to achieve. 
     It could be apparent to those skilled in the art that modifications and variations can be made to the preferred embodiments of this invention without departing from the scope or spirit of the invention as defined by the appended claims. One very obvious modification would be to duplicate the preferred embodiments of this invention by referring to the actual components of a thyristor equivalent circuit. Another modification would be to have the thyristor circuit incorporated in the same substrate as the MOSFET.

Technology Category: 5