Patent Document

CROSS-REFERENCE TO RELATED APPLICATIONS  
       [0001]     This Application is related to U.S. Non-Provisional Application Ser. No. ______, entitled “DUAL PDFE SYSTEM WITH FORWARD-BACKWARD VITERBI” and filed on Apr. 17, 2006, which application is incorporated herein by reference and for all purposes. 
     
    
     BACKGROUND OF THE INVENTION  
       [0002]     1. Field of the Invention  
         [0003]     The present invention relates to decoding of trellis-encoded signals and more particularly to systems and methods of symbol correction in prediction decision feedback equalization architectures  
         [0004]     2. Description of Related Art  
         [0005]     Since the adoption of the Advanced Television Systems Committee (“ATSC”) digital television (“DTV”) standard in 1996, there has been an ongoing effort to improve the design of receivers built for the ATSC DTV signal as described in the ATSC standard A/54 (see U.S. patent application publication 20050163209 for . . . ). Designers face major obstacles in designing receivers that might achieve good reception in the presence of multipath interference in the channel. Multipath interference affects the ability of the receiver to correctly decode transmitted symbols. Therefore, designers often add equalizers to receivers in order to cancel the effects of multipath interference and thereby improve signal reception.  
         [0006]     Referring to  FIG. 1 , in the ATSC DTV transmission system, data is transmitted in frames  10 . Each frame  10  is composed of 2 fields  11  and  12 , each field  11  and  12  having 313 segments, and each segment having 832 symbols. The first four symbols in each segment are segment sync symbols  13  having the sequence [+5, −5, −5, +5]. The first segment in each field is a field sync segment  14  and  15 .  
         [0007]     Referring to figure shown in more detail in  FIG. 2 , field sync  20  comprises segment sync  21 , a 511 symbol pseudo noise (PN 511 ) sequence  22 , a 63 symbol pseudo noise (PN 63 ) sequence  23 , a second PN 63  sequence  24 , a third PN 63  sequence  25 , and a 128 symbol sequence  26  composed of various mode, reserved, and precode symbols. The four PN sequences  22 - 25  are composed of symbols from the set {+5, −5}. In alternate fields, the three PN 63  sequences  23 - 25  are the same. In the remaining fields, the first PN 63   23  and third PN 63   25  are the same while the second PN 63   24  is inverted.  
         [0008]     As shown in  FIG. 3 , subsequent  312  segments  30  of the field  11  and  12  (referred to as data segments) are structured such that 828 symbols  32  following the four segment sync symbols  31  are trellis encoded by a 12 phase trellis encoder described in detail in ATSC standard A/54. This results in 8 level symbols derived from the alphabet {−7 −5 −3 −1 +1 +3 +5 +7}.  
         [0009]     Consider now an 8T-VSB transmitter such as is illustrated in  FIG. 4 . Input data  40  is first randomized  41 , Reed-Solomon byte wise encoded  42 , and then byte interleaved  43 . Next the data is trellis encoded by a 12-phase trellis encoder  44 . A multiplexer  45  adds the segment sync symbols and the field sync symbols to the trellis coded data at the appropriate times in the frame. Then, a pilot is inserted  46  by adding a DC level to the baseband signal and a modulator  47  modulates the resulting symbols to IF. Finally a RF upconverter  48  converts the signal for RF transmission as a vestigial sideband (VSB) signal at a symbol rate of 10.76 MHz.  
         [0010]     Now consider a baseband model of the transmission channel fed by the above transmitter. The transmitted signal has a root raised cosine spectrum with a nominal bandwidth of 5.38 MHz and an excess bandwidth of 11.5% centered at one fourth of the symbol rate (i.e., 2.69 MHz). Thus the transmitted pulse shape q(t) is complex and given by 
 
 q ( t )= e   jπF     x     t/2   q   RRC   (t) , 
 
 where F s  is the symbol frequency, and q RRC (t) is a real square root raised cosine pulse with an excess bandwidth of 11.5% of the channel. The pulse q(t) is referred to as the “complex root raised cosine pulse”. For the 8T-VSB system, the transmitter pulse shape q(t) and the receiver matched filter pulse shape q*(−t) are identical since q(t) is conjugate-symmetric. Thus the raised cosine pulse p(t), referred to as the “complex raised cosine pulse”, is given by 
 
 p ( t )= q ( t )* q *(− t ) 
 
 where * denotes convolution, and * denotes complex conjugation. The transmitted baseband signal of data rate 1/T symbols/sec can be represented as:  
           s   ⁡     (   t   )       =       ∑   k     ⁢       I   k     ⁢     q   ⁡     (     t   -   kT     )             ,       
 
 where {I k  ∈ A≡{α 1 , . . . α 8 }⊂R 1 } is the transmitted data sequence, which is a discrete 8-ary sequence taking values on the real 8-ary alphabet A. The physical channel between the transmitter and receiver is denoted c(t) and can be described by:  
         c   ⁡     (   t   )       =       ∑     k   =     -     L   ha           L   hc       ⁢       c   k     ⁢     δ   ⁡     (     t   -     τ   k       )               
 
 where {c k (τ)}⊂C 1 , L ha  and L hc  are the number of maximum anti-casual and casual multipath delays, τ k  is multipath delay, and δ(t) is the Dirac delta function. Hence, the overall channel impulse response is  
         h   ⁡     (   t   )       =         p   ⁡     (   t   )       *     c   ⁡     (   t   )         =       ∑     -     L   ha         L   hc       ⁢       c   k     ⁢     p   ⁡     (     t   -     τ   k       )                 
 
         [0011]      FIG. 5  shows an 8T-VSB receiver block diagram. A tuner  50  and IF filter  51  demodulate an RF signal to baseband. Next, timing and synchronization recovery is performed  52  along with the rejection of any NTSC interference  53 . The data is then equalized  54  and sent through a phase tracker  55  and trellis decoded  56 , de-interleaved  57 , Reed-Solomon decoded  58 , and finally de-randomized  59 . The matched filter output y(t) in the receiver prior to equalization is:  
           y   ⁡     (   t   )       =         (       ∑   k     ⁢     δ   ⁡     (     t   -   kT     )         )     *     h   ⁡     (   t   )         +     v   ⁡     (   t   )           ,         
 where 
   v ( t )=η( t )* q *(− t )  
 denotes the complex (colored) noise process after the pulse matched filter, with η(t) being a zero-mean white Gaussian noise process with spectral density σ n   2  per real and imaginary part. Sampling the matched filter output y(t) at the symbol rate produces the discrete time representation of the overall communication system according to the following equation: 
   y[n]≡y ( t )| t=nT   =ΣI   k   h[n−k]+v[n]   
         [0012]     Broadcast television channels are a relatively severe multipath environment due to a variety of conditions encountered in the channel and at the receiver. Only 728 symbols of a VSB field sync segment are known a priori and can be used as a training sequence for an adaptive equalizer. The channel is not known a priori, so the equalizer in the receiver must be able to adaptively identify and combat the various channel conditions. Since multipath signals in the broadcast channel may arrive many symbols after the main signal, the decision feedback equalizer (DFE) is invariably used in 8T-VSB applications. Another DFE structure that is well known is the noise predictive decision feedback equalizer (pDFE). Although both DFEs and pDFEs are good at combating multipath channels, both have the problem of error propagation. Error propagation occurs when there are errors in the feedback path. This, in turn, feeds erroneous data into the decision device resulting in incorrect symbol decisions. For 8T-VSB applications, the most commonly used decision device is the Viterbi Decoder. Therefore it is important to mitigate the effects of error propagation.  
         [0013]     Since the 8T-VSB symbols are convolutionally coded, they may be decoded in the receiver with a Viterbi decoder [ATSC Standard A/54, U.S. Pat. No. 5,600,677, U.S. Pat. No. 5,583,889]. The Viterbi Algorithm (VA) for maximum likelihood sequence estimation of transmitted symbols corrupted by white noise is very well known (see “The Viterbi Algorithm”, G. D. Forney, Jr., Proc. IEEE, vol. 61, pp. 268 -278, March 1973, “Digital Communications—Fundamentals and Applications”, Bernard Sklar, Prentice-Hall, 1988). The decoder may equivalently provide estimates of the encoded bit pairs or estimates of the mapped 8 level symbols, the later being utilized in the context of an equalizer. As is well known, the VA requires a path history memory for each state and involves add, compare, select operations based on trellis path metrics determined from sums of Euclidean distance branch metrics. As time advances, the most likely trellis paths (as indicated by the lowest path metrics) into each state of the trellis are saved, the rest are discarded. If the decoding algorithm searches back sufficiently deep in the trellis path memory, the result of discarding less likely paths—leaving only survivor paths—is a single surviving branch which defines the most likely symbol (hard symbol decision) at that prior point in time. At shallower path memory trace back depths (closer to the present time), there is a higher likelihood of multiple surviving branches with symbol probabilities proportional to the corresponding path metrics.  
       BRIEF SUMMARY OF THE INVENTION  
       [0014]     The present invention provides a novel technique for improving the performance of equalizers by reducing the effects of error propagation in equalizers that use a Viterbi Decoder. Systems and methods of symbol correction in prediction decision feedback equalization (“pDFE”) architectures are provided. Particularly, embodiments of the invention are described that include an enhanced Viterbi decoder and novel methods of symbol correction to obtain better system performance. The use of a blending algorithm is described to reduce errors in symbol decoding.  
         [0015]     The Viterbi Algorithm (“VA”) can be described as follows for an S state trellis with a path memory of length M for each state that holds a sequence of state transitions and associated branch metrics: 
        At each time increment n.     For each trellis state k.     Calculate the Euclidean branch metric for each branch into state k from all possible prior states at time (n−1).     Add the above branch metrics to the associated path metrics for each possible prior state at time (n−1).     Choose the path into state k at time n with the best path metric and store it in the path memory associated with state k (overwriting the previous stored path).     Decode symbol.     Examine path memory back to time (n-M); if M is large enough, the path memories for each of the S states will show the same state transition at time (n-M) and hence indicate the same symbol; choose that symbol as the hard decision.     If the state transitions for time (n-M) are not the same, choose the state transition (and hence the symbol) corresponding to the path that has the best path metric from time (n-M) to time n. 
 
 Instead of outputting a single symbol decision delayed by M, the decoder can output a M+1 long vector of symbol decisions with delays ranging from zero (corresponding to time n) to M as follows. For each time increment n, the Viterbi decoder updates the metrics and returns a vector of symbols whose length is M+1, where M is referred to as the trace back depth. As described above, the deep trace back depth symbol decisions will be more accurate than the shallow trace back depth symbols. 
       
 
         [0024]     As will be explained in more detail below, trellis decoders may use this advantage in the feedback path of an adaptive equalizer. For a given time n, it can be beneficial to update all M+1 symbols in the equalizer feedback path such that M trace back depth symbols can overwrite M previously decoded symbols in the feedback path, thereby updating symbol decisions for times (n-M) through n. Such updating of more accurate symbols can facilitate a reduction in error propagation in the equalizer feedback path. Consequently, the present invention provides novel techniques for improving the performance of equalizers by reducing the effects of error propagation in equalizers that use a Viterbi Decoder.  
         [0025]     The foregoing and other aspects of various embodiments of the present invention will be apparent through examination of the following detailed description thereof in conjunction with the accompanying drawings. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0026]     The present invention is illustrated by way of example, and not limitation, in the figures of the accompanying drawings in which like references denote similar elements, and in which:  
         [0027]      FIG. 1  depicts a frame in an ATSC DTV transmission system;  
         [0028]      FIG. 2  depicts a field sync segment in a frame in an ATSC DTV transmission system;  
         [0029]      FIG. 3  depicts a data segment in a frame in an ATSC DTV transmission system;  
         [0030]      FIG. 4  illustrates an 8T-VSB transmitter;  
         [0031]      FIG. 5  shows an 8T-VSB receiver block diagram;  
         [0032]      FIG. 6  illustrates prediction decision feedback equalization as included in certain embodiments of the invention; and  
         [0033]      FIG. 7  is an example of a system according to one embodiment of the invention.  
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0034]     Embodiments of the present invention will now be described in detail with reference to the drawings, which are provided as illustrative examples so as to enable those skilled in the art to practice the invention. Notably, the figures and examples below are not meant to limit the scope of the present invention. Wherever convenient, the same reference numbers will be used throughout the drawings to refer to same or like parts. Where certain elements of these embodiments can be partially or fully implemented using known components, only those portions of such known components that are necessary for an understanding of the present invention will be described, and detailed descriptions of other portions of such known components will be omitted so as not to obscure the invention. Further, the present invention encompasses present and future known equivalents to the components referred to herein by way of illustration.  
         [0035]     Certain embodiments provide systems and methods of symbol correction in prediction decision feedback equalization (“pDFE”) architectures. Certain of the methods and systems described can also be applied to conventional decision feedback equalization (“DFE”) architectures.  
         [0036]     Referring to  FIG. 6 , an example of a pDFE architecture is illustrated. A feed forward filter  61  performs block based frequency domain filtering on data received at an input  60 . Summing element  63  adds output  62  from input filter  61  to output  69  from feedback filter  68 . The summed output is then provided to Viterbi decoder  64 . Viterbi decoder  64  provides the pDFE output  65 . Typically, frequency domain filter  61  is block based and output  65  consequently comprises a block of symbols. Output  65  is then added to input filter output  62  using summer  66  to provide an input  67  to feedback filter  68 . Feedback filter  68  can include a noise predictor (not shown). The noise predictor in the feedback loop can estimate colored (non-white) noise from error signal  67 . Adder  63  may then subtract the estimated colored noise from the equalized data, thereby helping Viterbi decoder  64  to make better decisions.  
         [0037]     In certain embodiments, Viterbi Decoder  64  can store metrics for a plurality of states including a smallest metric obtained, a previous state, and a current state. As discussed above, the metrics are typically used to configure or adjust a Viterbi algorithm that requires a path history memory for each state. The metrics can be based on trellis path metrics determined from sums of Euclidean distance branch metrics. The condition of the plurality of stored metrics is used to determine which symbol is decoded. If a delay is incurred, Viterbi Decoder  64  may be able to correct some symbols using trace back depth decoding.  
         [0038]     Referring now to  FIG. 7 , embodiments of the invention include an enhanced Viterbi Decoder  70  as a decision device coupled with a novel method of symbol correction that can deliver better system performance. In certain embodiments a Viterbi algorithm operates on an S state trellis with a path memory of length M for each state that holds a sequence of state transitions and associated branch metrics and can output an M+1 long vector of symbol decisions with delays ranging from zero (corresponding to time n) to M as follows. For each time increment n, the Viterbi decoder may update the metrics and return a vector of symbols whose length is M+1, where M is referred to as the trace back depth. Deep trace back depth symbol decisions can be more accurate than shallow trace back depth symbols.  
         [0039]     In certain embodiments, Viterbi Decoder  70  can determine metrics associated with potential decoding paths wherein the metrics can be used to assess reliability of decoded symbols. Viterbi Decoder  70  can output a first vector  71  representing the most likely decoded symbols. The most likely symbols are typically determined by considering trace back depth decoding. Additionally, Viterbi Decoder  70  can output a second vector  72  representing second most likely decoded symbols and a difference metric (“diff_metric”)  73  quantifying a difference in estimated reliability of first vector  71  and second vector  72 . Diff_metric  73  can be used to ascertain the reliability of a trellis decoded symbol such that a large diff_metric  73  value may indicate reliability of decoded symbols while small a diff_metric  73  value can be indicative of decoded symbols that are unreliable. In certain embodiments, a blender  74  can apply a blending algorithm on first vector  71  and second vector  72  based on the diff_metric  73 . Having received decoded symbols from Viterbi Decoder  70 , blender  74  can blend the decoded symbols with long delayed trellis symbols from Viterbi Decoder  70 .  
         [0040]     More particularly, diff_metric  73  is a measure of reliability that can be calculated as the difference of the two smallest surviving path metrics. For deep trace back depth symbols, the corresponding symbols of first vector  71  and second vector  72  can often be identical, indicating that a single surviving path exists at that point. However, for shallow trace back depth symbols, it is more likely that the corresponding symbols from first vector  71  and second vector  72  will be different, indicating that multiple surviving paths exist at that point. Path metrics can be calculated that indicate variances of the decoded path from a measured signal path and it will be appreciated that a smallest path metric typically indicates the most probable path. However, where multiple surviving paths exist, the smallest metric surviving path may generate errors in the decoded symbols. Diff_metric  73  can quantify the probability of errors by indicating the difference in path metrics between most likely surviving paths. In the example of  FIG. 7 , diff_metric  73  is calculated as the difference in path metrics between first vector  71  and second vector  72 .  
         [0041]     In certain embodiments, diff_metric  73  is used to assess the reliability of first vector  71 . A large diff_metric may be interpreted as an indication that the decoded symbols are reliable. On the other hand, a small diff_metric may be interpreted as an indication that the decoded symbols are unreliable. This reliability information can be provided to blender  74  for executing a blending algorithm on first vector  71  and second vector  72 . The blending algorithm may apply weighting factors based on one or more successive diff-metric  73  values to generate error compensation in the pDFE of  FIG. 7 .  
         [0042]     Referring to Table 1, an example may better illustrate weighting as employed by blender  74  in certain embodiments. Taking a scalar weighting factor a as weight for the most likely path while the scalar weighting factor b weights the second most likely path. Then a and b can be selected depending on the value of diff_metric  73  as shown in Table 1.  
                                                         TABLE 1                                       1.0 &lt;   2.0 &lt;               diff_metric &lt;=   diff_metric &lt;=   diff_metric &lt;=   3.0 &lt;           1.0   2.0   3.0   diff_metric                                    a   0.5   0.625   0.75   1.0       b   0.5   0.375   0.25   0.0                  
 
         [0043]     Consider another example in which a trace back depth of 8 is assumed. Where the symbols for the most likely path of a Viterbi Decoder are [−5 1 3 5 1 −3 3 −3 −7], the symbols for the second most likely path of that same Viterbi Decoder are [−7 −1 3 3 −1 1 3 −3 −7] and the corresponding diff_metric  73  is 2.6. Then, according to Table 1 above, the weighting factors are a=0.75 and b=0.25. In this example, the resulting new trace back depth decoded vector would be: 
 
0.75*[−5 1 3 5 1 −3 3 −3 −7]+0.25*[−7 −1 3 3 −1 1 3 −3 −7]=[−5.5 0.5 3.0 4.5 0.5 −2.0 3.0 −3.0 −7.0]. 
 
 This new “soft” vector can then be used in the feedback path  67 - 69  of  FIG. 7  instead of the most likely path vector  71 . Consequently, is able to more accurately predict the noise and improve the performance of the pDFE. 
 
         [0044]     In many embodiments, implementation of the described methods of decoding symbols can require little additional hardware. For example, the Viterbi Decoder block may be implemented using an additional M+1 memory units for storing second trace back depth symbols, where M represents the trace back depth. An adder may also be needed to calculate the diff 13  metric. For the Blend block, three comparators, 8 preset taps (for a and b), 2M multipliers and M adders may be needed.  
         [0045]     Simulations have been performed using the example depicted in  FIG. 7 , with the criterion of SER &lt;=0.0050 for a trace back depth of 16. Six different channels were tested. In comparison to the pDFE system in  FIG. 6 , the pDFE in  FIG. 7  performs better. For Brazil A, there is a 0.1 dB improvement (signal-to-noise ratio “SNR” input can be 0.1 dB less). Brazil B shows a 0.2 dB improvement, Brazil C shows a 0.7 dB improvement, Brazil D shows a 0.4 dB improvement and Brazil E shows a 0.9 dB improvement. This improvement incurs a minor cost in hardware.  
         [0046]     It will be appreciated that the Brazil channels A-E are difficult multipath channels used for reference testing. They include ghosts of varying delays and amplitudes. They are referenced in Interferencia por Multipercurso—Simulacao De Canais Tipo:“Rice”, “Rayleigh”, Harbour Apartment” E “SNF”, SET-ABERT, Feb. 11, 1999.  
         [0047]     It is apparent that the above embodiments may be altered in many ways without departing from the scope of the invention. Further, various aspects of a particular embodiment may contain patentably subject matter without regard to other aspects of the same embodiment. Additionally, various aspects of different embodiments can be combined together. Also, those skilled in the art will understand that variations can be made in the number and arrangement of components illustrated in the above diagrams. It is intended that the appended claims include such changes and modifications.

Technology Category: h