Patent Document

CROSS-REFERENCE TO RELATED APPLICATIONS  
       [0001]     This application claims priority to Application No. 60/854,783 filed on Oct. 27, 2006. 
     
    
     BACKGROUND OF THE INVENTION  
       [0002]     1. Technical Field of the Invention  
         [0003]     The subject matter disclosed generally relates to the field of electronic systems and methods. More specifically, the subject matter disclosed relates to electronic arrangements that allow cost reduction and increased utility for microwave and millimeter-wave imaging applications.  
         [0004]     2. Background of Related Art  
         [0005]     Imaging sensors utilizing electromagnetic signals in the millimeter-wave frequency spectrum have demonstrated the ability to detect objects obscured from sight such as handguns concealed underneath clothing and metal objects inside of bags, due to the ability of signals in this frequency range to penetrate clothing and various other materials. For this discussion, the term “millimeter-wave” includes the microwave spectrum and refers to frequencies in the range of, but not limited to, 1 GHz-1 THz. Active versions of these sensors typically utilize a transmitted signal that reflects from objects, and is then received and processed to create an image of the differences of reflectivity of the objects. Passive versions of these sensors typically utilize a high sensitivity receiver to receive naturally emitted energy from objects to create an image of the thermal differences of the objects. Passive sensors can have an advantage of not requiring transmit circuitry, but typically have poorer image contrast in indoor applications, and also typically require a higher sensitivity receiver which can add cost.  
         [0006]     The utility and performance of images created by millimeter-wave signals in detecting concealed weapons or identification of objects is typically related to image resolution. To increase the number of image pixels, reduce the size of image pixels, or provide higher image resolution, typically the size and cost of an imaging sensor is increased. This can be detrimental to application and deployment. In addition, image quality and weapon detection capabilities can be enhanced through the use of frequency modulation or multiple frequency operation of the sensor. This, however, typically adds cost and complexity to the sensor.  
         [0007]     To facilitate mass deployment of millimeter-wave imaging sensors, reduction of the sensor cost, size, and weight, and improvement of the imaging performance are desirable. Some prior-art millimeter-wave imaging sensor methods for cost reduction utilize mechanical scanning of a fixed antenna or array of antennas to create a two-dimensional scanned image with a reduced number of millimeter-wave components. However, there exists a finite speed in which mechanical scanning can be performed, often on the order of one second or longer for practical systems. Such slow rates can cause image blurring or reduced performance in applications where the object being imaged is not still, or in handheld applications where the imaging sensor is not still during this scanning period. Electrically sequenced, or scanned, two-dimensional antenna arrays can provide a much faster scanning time, typically on the order of tens of milliseconds, but can suffer from high cost due to the millimeter-wave hardware for realization of the electrically sequenced or scanned two-dimensional antenna array.  
         [0008]     It would be desirable to have an electrically sequenced or scanned two-dimensional active millimeter-wave imaging array with fast scan time and frequency modulation capability in a low-cost, mass-production-capable design. In addition, it would be desirable to have a low-cost mechanically-scanned array for applications where body or object motion blurring is not of concern. Also, it would be desirable to have a two-dimensional active millimeter-wave imaging array with complex signal sampling and frequency modulation capability compatible with digital beam-forming, super-resolution, two-dimensional and three-dimensional image processing techniques well known in the art, as well as weapon signature detection techniques such as, but not limited to, frequency response signatures. In addition, an implementation which has multiple polarization capability can have additional advantage. Furthermore, an implementation which has a small size, light weight, and portability can have further advantage.  
       BRIEF SUMMARY OF THE INVENTION  
       [0009]     An antennae system for a detector. The antennae system includes a two-dimensional electro-magnetic transmitter array that has an x number of transmitter elements, and a two-dimensional electro-magnetic receiver array that has a y number of receiver elements. The two-dimensional electro-magnetic transmitter and receiver arrays have a spatial relationship such that at least one subset of the two-dimensional electro-magnetic transmitter and receiver arrays forms a regular array of spatial displacements of z pairwise combinations of transmitter and receiver elements, where z is greater than the sum of x and y.  
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0010]     The accompanying drawings are for the purpose of illustrating and expounding the features involved in the present invention for a more complete understanding, and not meant to be considered as a limitation, wherein:  
         [0011]      FIG. 1A  is a block diagram illustrating features of one embodiment of an imaging sensor architecture according to aspects of the present invention.  
         [0012]      FIG. 1B  is a block diagram illustrating features of another embodiment of an imaging sensor architecture according to aspects of the present invention.  
         [0013]      FIG. 1C  is a block diagram illustrating features of an alternate embodiment of an imaging sensor architecture according to aspects of the present invention.  
         [0014]      FIG. 1D  is a block diagram illustrating features of an antenna arrangement according to aspects of the present invention.  
         [0015]      FIG. 2A  is a diagram illustrating received signal phase from an object across an antenna array aperture according to aspects of the present invention.  
         [0016]      FIG. 2B  is a diagram illustrating a transmit and receive antenna arrangement according to aspects of the present invention.  
         [0017]      FIG. 2C  is a diagram illustrating a one-dimensional virtual array arrangement according to aspects of the present invention.  
         [0018]      FIG. 2D  is a diagram illustrating a two-dimensional thinned-array transmit and receive antenna arrangement according to aspects of the present invention.  
         [0019]      FIG. 2E  is a diagram illustrating a two-dimensional virtual array arrangement according to aspects of the present invention.  
         [0020]      FIG. 3A  is a block diagram illustrating features of one embodiment of an antenna network according to aspects of the present invention.  
         [0021]      FIG. 3B  is a block diagram illustrating features of another embodiment of an antenna network according to aspects of the present invention.  
         [0022]      FIG. 3C  is a block diagram illustrating features of a further embodiment of an antenna network according to aspects of the present invention.  
         [0023]      FIG. 4A  is an electrical block diagram illustrating features of one embodiment of an imaging sensor architecture according to aspects of the present invention.  
         [0024]      FIG. 4B  is an electrical block diagram illustrating features of another embodiment of an imaging sensor architecture according to aspects of the present invention.  
         [0025]      FIG. 5  is a diagram illustrating operational timing according to aspects of the present invention.  
         [0026]      FIG. 6A  is an electrical block diagram illustrating features of a further embodiment of an imaging sensor architecture according to aspects of the present invention.  
         [0027]      FIG. 6B  is an electrical block diagram illustrating features of a yet further embodiment of an imaging sensor architecture according to aspects of the present invention.  
         [0028]      FIG. 6C  is a diagram illustrating a two-dimensional thinned-array, dual-polarized transmit and receive antenna arrangement according to aspects of the present invention.  
         [0029]      FIG. 7A  is a block diagram illustrating features of one embodiment of the signal generator  405  according to aspects of the present invention.  
         [0030]      FIG. 7B  is a block diagram illustrating features of another embodiment of the signal generator  405  according to aspects of the present invention.  
         [0031]      FIG. 7C  is a block diagram illustrating features of a further embodiment of the signal generator  405  according to aspects of the present invention.  
         [0032]      FIG. 7D  is a block diagram illustrating features of a yet further embodiment of the signal generator  405  according to aspects of the present invention.  
         [0033]      FIG. 7E  is a block diagram illustrating features of another embodiment of the signal generator  405  according to aspects of the present invention.  
         [0034]      FIG. 7F  is a block diagram illustrating features of one embodiment of the TX &amp; LO signal generator  407  according to aspects of the present invention.  
         [0035]      FIG. 7G  is a block diagram illustrating features of another embodiment of the TX &amp; LO signal generator  407  according to aspects of the present invention.  
         [0036]      FIG. 8A  illustrates an output waveform from the signal generator  405  or TX &amp; LO signal generator  407  in accordance with one embodiment of the present invention.  
         [0037]      FIG. 8B  illustrates an output waveform from the signal generator  405  or TX &amp; LO signal generator  407  in accordance with another embodiment of the present invention.  
         [0038]      FIG. 8C  illustrates an output waveform from the signal generator  405  or TX &amp; LO signal generator  407  in accordance with a further embodiment of the present invention.  
         [0039]      FIG. 8D  illustrates an output waveform from the signal generator  405  or TX &amp; LO signal generator  407  in accordance with a yet further embodiment of the present invention.  
         [0040]      FIG. 9A  is a diagram illustrating antenna selection timing according to aspects of the present invention.  
         [0041]      FIG. 9B  is a diagram illustrating antenna selection timing according to aspects of the present invention.  
         [0042]      FIG. 9C  is a diagram illustrating antenna selection timing according to aspects of the present invention.  
         [0043]      FIG. 9D  is a diagram illustrating antenna selection timing according to aspects of the present invention.  
         [0044]      FIG. 9E  is a diagram illustrating antenna selection timing according to aspects of the present invention.  
         [0045]      FIG. 9F  is a diagram illustrating A/D converter sample timing according to aspects of the present invention.  
     
    
     DETAILED DESCRIPTION  
       [0046]     An imaging sensor arrangement is presented in  FIG. 1A  as one embodiment of aspects of the present invention. In this arrangement, the transmit signal generator  650  outputs u signals to a multi-dimensional thinned transmit antenna network  601  for electromagnetic transmission, where u is an integer greater than or equal to 1. A typical frequency of the transmitted signal from the multi-dimensional thinned transmit antenna network  601  can be within, but is not limited to, the frequency range of 1 GHz-1 THz, and can be a fixed frequency or be frequency modulated. The imaging sensor&#39;s total occupied transmit spectral bandwidth is dependent on the frequency modulation bandwidth, and can be wideband (WB) or ultra-wideband (UWB) in order to achieve adequate range resolution for some applications. A typical WB bandwidth value can be, but is not limited to, a value greater than 100 MHz. A typical UWB bandwidth value can be, but is not limited to, a value greater than 1 GHz. The reflected signal from an object is received by the multi-dimensional thinned receive antenna network  621 , which outputs v signals to a receiver/down-converter  670 , where v is an integer greater than or equal to 1. The receiver/down-converter  670  also accepts q signals from the transmit signal generator  650 , where q is an integer greater than or equal to 1, and outputs one or a plurality of signals each comprising at least one of the frequency or phase difference between components of the transmitted signal and corresponding received reflected signal from an object as an input to a signal processor  690 . The receiver/down-converter  670  can utilize one or a plurality of down-conversion operations in generating the output difference signals. The transmit signal generator  650  can include, but is not limited to, generation of one or a plurality of fixed frequency or frequency modulated signals, intermediate frequency signal generation, local oscillator signal generation, transmit and/or receive gating signal generation, or transmit pulsing signal generation. The multi-dimensional thinned transmit antenna network  601  can include, but is not limited to, a two-dimensional array of spatially separated antennas, multiple one-dimensional arrays arranged in multiple axes, a conformal array of spatially separated antennas, a three-dimensional array of spatially separated antennas, or one or a plurality of groups of spatially separated antennas with one or a plurality of antennas simultaneously selected for transmission of one or a plurality of signals, wherein at least two adjacent antennas have a distance between them that is different than at least two adjacent antennas in the multi-dimensional thinned receive antenna network  621 . The multi-dimensional thinned receive antenna network  621  can include, but is not limited to, a two-dimensional array of spatially separated antennas, multiple one-dimensional arrays arranged in multiple axes, a conformal array of spatially separated antennas, a three-dimensional array of spatially separated antennas or one or a plurality of groups of spatially separated antennas with one or a plurality of antennas simultaneously selected for reception of one or a plurality of signals, wherein at least two adjacent antennas have a distance between them that is different than at least two adjacent antennas in the multi-dimensional thinned transmit antenna network  601 . According to aspects of the present invention, the multi-dimensional thinned transmit antenna network  601  and the multi-dimensional thinned receive antenna network  621  are utilized to synthesize an array having more elements than the sum of the elements contained in multi-dimensional thinned transmit antenna network  601  and the multi-dimensional thinned receive antenna network  621 , for the purpose of reducing the sensor hardware necessary for imaging applications. The term “thinned” in this application refers to the utilization of a lower number of physical transmit and receive antenna elements to synthesize an array with a larger number of synthesized or virtual elements than the sum of the physical transmit and receive elements. The term “imaging” in this application includes, but is not limited to, multi-dimensional object image construction, detection or identification of objects using, but not limited to, image processing or image recognition techniques, and/or object signatures such as, but not limited to, radar cross-section signatures, angular cross-section signatures, range cross-section signatures, wideband or ultra-wideband frequency response signatures, wideband or ultra-wideband frequency resonance signatures, or polarization signatures. Examples of objects that may be detected using imaging techniques can include, but are not limited to, concealed weapons, guns, knives, explosives, contraband, or improvised explosive devices (IED&#39;s).  
         [0047]     Signal processor  690  may comprise a single or plurality of individual processors. Signal processor  690  may perform, but is not limited to, any single or combination of the functions of signal or image processing, real or complex DFT or FFT signal processing, CFAR threshold detection, spectral peak detection, target peak association, frequency measurement, magnitude measurement, phase measurement, magnitude scaling, phase shifting, spatial FFT processing, digital beam-forming (DBF) processing, super-resolution processing such as, but not limited to, the use of the multiple signal classification algorithm (MUSIC) or the estimation of signal parameters via rotational invariance techniques (ESPRIT) algorithm, neural network processing, two-dimensional image processing, three-dimensional image processing, two or three-dimensional image reconstruction processing, microwave or millimeter-wave holography processing, backward-wave reconstruction processing, wavefront reconstruction processing, synthetic aperture radar (SAR) processing, or Kirchoff diffraction integral processing. Additional processing techniques used in the above-mentioned functions may include, but are not limited to, windowing, digital filtering, Hilbert transform, least squares algorithms, or non-linear least squares algorithms. Furthermore, one or a combination of object signature methods can be used to determine the presence or identification of potential threats, weapons or contraband such as, but not limited to, radial cross-section characteristics, angular cross-section characteristics, strength of returns, wideband or ultra-wideband frequency response characteristics, wideband or ultra-wideband frequency resonance characteristics, polarization response characteristics, spectral absorption characteristics, or image shape characteristics, and such signatures may be determined for the entire object or for one or more regions of an object or detection zones. The signal processor may include, but is not limited to, one or more digital signal processors (DSPs), microprocessors, micro-controllers, electrical control units, or other suitable processor blocks.  
         [0048]     An imaging sensor arrangement is presented in  FIG. 1B  as another embodiment of aspects of the present invention. The arrangement in  FIG. 1B  is similar to the arrangement in  FIG. 1A , except that instead of the multi-dimensional thinned transmit antenna network  601  and multi-dimensional thinned receive antenna network  621 , a mechanically scanned thinned transmit antenna network  601   b  and mechanically scanned thinned receive antenna network  621   b  are utilized. The same components are denoted by the same reference numerals, and will not be explained again. In this arrangement, the transmit signal generator  650  outputs u signals to a mechanically scanned thinned transmit antenna network  601   b  for electromagnetic transmission, where u is an integer greater than or equal to 1. A typical frequency of the transmitted signal from the mechanically scanned thinned transmit antenna network  601   b  can be within, but is not limited to, the frequency range of 1 GHz-1 THz, and can be a fixed frequency or be frequency modulated. The imaging sensor&#39;s total occupied transmit spectral bandwidth is dependent on the frequency modulation bandwidth, and can be wideband (WB) or ultra-wideband (UWB) in order to achieve adequate range resolution for some applications. The reflected signal from an object is received by the mechanically scanned thinned receive antenna network  621   b , which outputs v signals to a receiver/down-converter  670 , where v is an integer greater than or equal to 1. The receiver/down-converter  670  also accepts q signals from the transmit signal generator  650 , where q is an integer greater than or equal to 1, and outputs one or a plurality of signals each comprising at least one of the frequency or phase difference between components of the transmitted signal and corresponding received reflected signal from an object as an input to a signal processor  690 . This arrangement utilizes a one-dimensional or multi-dimensional thinned transmit array and a one-dimensional or multi-dimensional thinned receive array, mechanically scanned or dithered in one or more directions for the purpose of sampling different spatial positions for the elements along the mechanically scanned or dithered direction. For example, not meant as a limitation, a one-dimensional azimuth line-array consisting of a transmit array with spacing D and a receive array with spacing different than D and a position where there is no overlap in the azimuth dimension between transmit and receive arrays, is mechanically scanned in the elevation dimension. Through spatial sampling at various positions in elevation during the mechanical scanning in that dimension, a two dimensional set of array measurements is achieved and can be utilized for image processing. In another example, not meant as a limitation, a two-dimensional thinned transmit array and a two-dimensional thinned receive array are utilized, where one or both of the arrays utilize positional dithering in one or more directions in order to provide additional spatial sampling positions in the synthesis of a virtual array. The thinned transmit and receive arrays are utilized to reduce the hardware necessary for the imaging sensor, as is the mechanical scanning and spatial sampling along the mechanical scanning path. When the thinned array and mechanical scanning methods are utilized in combination, the sensor hardware required and/or sensor cost can be reduced for applications where the mechanical scan time is acceptable.  
         [0049]     An imaging sensor arrangement is presented in  FIG. 1C  as an alternate embodiment of aspects of the present invention. The arrangement in  FIG. 1C  is similar to the arrangement in  FIG. 1A , except that a processor  690   a  provides u output signals to a multi-dimensional thinned transmit antenna network  601  for electromagnetic transmission, and accepts v input signals from a multi-dimensional thinned receive antenna network  621 , where u and v are each integers greater than or equal to 1. The same components are denoted by the same reference numerals, and will not be explained again. A typical frequency of the transmitted signal from the multi-dimensional thinned transmit antenna network  601  can be within, but is not limited to, the frequency range of 1 GHz-1 THz, and can be a fixed frequency or be frequency modulated. The imaging sensor&#39;s total occupied transmit spectral bandwidth is dependent on the frequency modulation bandwidth, and can be wideband (WB) or ultra-wideband (UWB) in order to achieve adequate range resolution for some applications. In addition, this arrangement can be mechanically scanned or dithered in one or more directions for the purpose of sampling different spatial positions for the elements along the mechanically scanned or dithered direction.  
         [0050]     An antenna arrangement with mechanical movement capability is presented in  FIG. 1D  as an embodiment of aspects of the present invention. The example of an antenna arrangement with mechanical movement capability shown in  FIG. 1D  is for illustration purposes and is not considered a limitation. In this arrangement, a mechanical actuator  601   d  provides mechanical movement of a multi-dimensional antenna array  601   c  in one or more directions. The arrangement shown in  FIG. 1D  can be utilized to provide mechanical movement for a transmit array, a receive array, or both transmit and receive arrays in one or more directions. In addition, the arrangement shown in  FIG. 1D  can be utilized to provide mechanical dithering for a transmit array, a receive array, or both transmit and receive arrays in one or more directions. Furthermore, the arrangement shown in  FIG. 1D  can be utilized to provide mechanical scanning for a transmit array, a receive array, or both transmit and receive arrays in one or more directions.  
         [0051]      FIG. 2A  illustrates the phase shift in received signals from an object  22  for spatially separated antennas  157 ,  158 ,  159 ,  160  across an array, according to aspects of the present invention. The example of antenna spatial separation shown in  FIG. 2A  is for illustration purposes and is not considered a limitation. In this arrangement, k antennas  157 ,  158 ,  159 ,  160  are separated from one another in the axis of object direction (θ determination as illustrated in  FIG. 2A . The axis of object direction determination can be, but is not limited to, the azimuth or the elevation axis. As can be seen, the received reflected signals from object  22  at angle θ from boresight will generate phase shifts ΔΨ 1,2 , ΔΨ 1,k-1 , ΔΨ 1,k  between ANT  1  and the other antenna elements due to the angle of the reflected RF wavefronts as illustrated. For an antenna array, these received phase shifts can be utilized to determine the direction of an object, and it is the unique spatial position of the elements in the array that allows unique phase sampling of the received signals across the array. The concept of building an array from a set of unique phase length combinations between transmit and receive elements makes it possible for a thinned transmit and thinned receive array to synthesize an array having a larger number of elements than the sum of the transmit and receive array elements, which is termed a “virtual array” in the present invention.  
         [0052]     Through selection of various combinations of transmit and receive antenna pairs, a receive antenna array, or virtual array, is synthesized with the number of elements and spacing of elements based upon the number of unique transmit and receive pairs selected and the physical spacing between the elements of these pairs. Let the physical transmit antenna elements  140   a ,  140   b  and receive antenna elements  145   a ,  145   b ,  145   c ,  145   d  be spaced in the axis of target direction determination as illustrated in  FIG. 2B . Let transmit antenna TX 1  be selected and receive antenna RX 1  be selected simultaneously. During the radar dwell time let the down-converted signals be digitized and stored. Then let the receive element RX 2  be selected for the next radar dwell time during which the down-converted signals be digitized and stored. Perform the same operations for the elements RX 3  and RX 4 . Repeat the above receive antenna selection settings for the next four radar dwell times but with the transmit antenna TX 2  selected instead of the transmit antenna TX 1 . When completed, digitized down-converted signals corresponding to 8 combinations of transmit and receive antenna selections will be stored and can be used for image processing. The 8 combinations of transmit and receive antenna selections can be used to synthesize a receive virtual array  150  of 8 elements with each element having a center-to-center spacing of D as illustrated in  FIG. 2C . As an example, not meant as a limitation, let the antenna combination of TX 1  RX 1  be utilized for the received signal reference. Then the next antenna combination in the virtual array, which is TX 1  RX 2  in  FIG. 2C , will have a relative amplitude and phase of the received signal with respect to the received signal reference that is equivalent to that of an antenna element being offset by distance D from the reference element as shown. Continuing the example, the third element in the virtual array, which is TX 1  RX 3  in  FIG. 2C , will have a relative amplitude and phase of the received signal with respect to the received signal reference that is equivalent to that of an antenna element being offset by distance 2*D from the reference element as shown. This can be repeated for all the elements in the virtual array. One advantage of using the thinned transmit and thinned receive arrays illustrated in FIG.  2 B is that only 6 antenna elements were needed to synthesize an 8-element virtual array as shown in  FIG. 2C  resulting in a reduction in hardware. For larger one-dimensional or two-dimensional thinned arrays, the hardware savings can be much greater. The example illustrated in  FIG. 2B  is for a one-dimensional array where the spacing distance between the transmit and receive elements is utilized in synthesizing a one-dimensional virtual array. For multi-dimensional arrays, the spatial displacement between selected transit and receive element pairs must be used in synthesizing the virtual array element spatial positions rather than the spacing between them as for the one-dimensional array. The spatial displacement is a vector quantity which is composed of the scalar displacement values in each of the dimensions of the multi-dimensional arrays. For example, for two-dimensional transmit and receive arrays, the spatial displacement between a selected pair of transmit and receive elements would include a scalar value for the difference in x coordinates between the elements, and a scalar value for the difference in y coordinates between the elements. It is the set of unique spatial displacements between transmit and receive element pairs that is utilized to synthesize a multi-dimensional virtual array.  
         [0053]     The thinned array arrangement shown in  FIG. 2B  can be modified according to aspects of the present invention. One example of such a modification, not meant as a limitation, can be to utilize a spacing between receive antenna elements that is greater than a spacing between transmit antenna elements. As an example, not meant as a limitation, the antenna elements  140   a ,  140   b  in  FIG. 2B  can be utilized for a receive function, and the antenna elements  145   a ,  145   b ,  145   c ,  145   d  can be utilized for a transmit function as part of a thinned array configuration. Another example of such a modification, not meant as a limitation, can be to utilize a non-uniform spacing between elements.  
         [0054]     A two-dimensional, bi-static thinned-array arrangement is presented in  FIG. 2D  as one embodiment of aspects of the present invention. In this arrangement, a k by p RX antenna array  168  is illustrated with an element-to-element spacing of D in each axis, and an m by n TX antenna array  165  is illustrated with an element-to-element spacing of k*D in the y-axis and p*D in the x-axis, where m and n are non-zero integers whose sum is greater than or equal to 3, and k and p are non-zero integers whose sum is greater than or equal to 3. In this arrangement, the TX antenna array  165  and RX antenna array  168  are illustrated to be oriented diagonally with respect to each another, where the rows of the TX antenna array  165  span a range in the x-axis that is non-overlapping with the span of the rows of the RX antenna array  168  in the x-axis, and the columns of the TX antenna array  165  span a range in the y-axis that is non-overlapping with the span of the columns of the RX antenna array  168  in the y-axis. Whether the arrays are one-dimensional or multi-dimensional, utilizing non-overlapping arrays allows synthesis of a virtual array having an order equal to the multiplication of the orders of the smaller arrays. As an example, using this arrangement, an (m*k) by (n*p) array having m*n*k*p elements can be synthesized from the unique combinations of transmit and receive elements, resulting in a reduction in sensor hardware. As an example, not meant as a limitation, let m=n=k=p=3. For this exemplary arrangement, the synthesized 9-by-9 virtual array  210  is illustrated in  FIG. 2E  according to aspects of the present invention. In the virtual array  210 , let the antenna combination T 1 , 1  R 1 , 1  be defined as the reference element in the virtual array  210 , and let the received signal for that reference element be defined as the reference signal for the virtual array  210 . The remaining elements of the virtual array  210  have relative spatial displacements from the reference element that correspond to the sum of the relative spatial displacements of the physical transmit and receive element pair with respect to the physical T 1 , 1  R 1 , 1  element pair that represents the reference element in the virtual array  210 . Since all the sums of the relative spatial displacements of the physical transmit and receive element pairs with respect to the physical T 1 , 1  R 1 , 1  element pair are unique, the corresponding relative spatial positions in the virtual array  210  with respect to the reference element are unique, resulting in a fully populated virtual array having a number of virtual elements that is far greater than the sum of the physical transmit and receive elements that was used to synthesize it. Using that definition of reference element in the virtual array  210 , the antenna combinations indicated in the virtual array  210  will have a relative amplitude and phase of the corresponding received signal with respect to the defined reference signal that is equivalent to that of an antenna element having a physical position relative to the reference element as shown in  FIG. 2E . Since many image processing techniques, such as, but not limited to, digital beam-forming processing, utilize the relative phase of measurements made between elements in a two-dimensional array, the absolute phase resulting from the positional offset of the RX antenna array  168  relative to the TX antenna array  165  can be non-critical, since it is the relative distances between elements within each array that affects the synthesized virtual array configuration. However, it may be advantageous to have the TX and RX arrays close to one another to avoid other issues that may cause performance degradation, such as, but not limited to, the difference in transmit illumination angles versus reception angles, or performance of the virtual array for imaging objects that are closer than the far-field. The digitized, down-converted signals corresponding to the transmit and receive antenna combinations illustrated in the virtual array in  FIG. 2E  can be utilized for object imaging, through the use of image processing techniques well known in the art, such as, but not limited to, digital beam-forming (DBF) processing, super-resolution processing, such as, but not limited to, the use of the multiple signal classification algorithm (MUSIC), or the estimation of signal parameters via rotational invariance techniques (ESPRIT) algorithm, spatial Fourier transform processing, two-dimensional image processing, three-dimensional image processing, two or three-dimensional image reconstruction processing, microwave or millimeter-wave holography processing, backward-wave reconstruction processing, wavefront reconstruction processing, synthetic aperture radar (SAR) processing, or Kirchoff diffraction integral processing. The examples shown are meant as an illustration of virtual array synthesis techniques, not as a limitation. For example, not meant as a limitation, the distance between elements in each array need not be constant, but can be varied or be given multiple different values by one skilled in the art for advantage. In addition, not meant as a limitation, the spacing between receive array elements can be greater than the spacing between transmit array elements. As an example, not meant as a limitation, the antenna array  168  can be utilized for a transmit function and the antenna array  165  can be utilized for a receive function as part of a thinned array configuration. Another example, not meant as a limitation, can be for a transmit array to be a one-dimensional array positioned at an angle or orthogonal to a one-dimensional receive array for the purpose of synthesizing a virtual array without departing from the spirit of the present invention. Furthermore, overlapping or intertwined transmit and receive arrays may be utilized to synthesize a virtual array without departing from the spirit of the present invention. Other array sizes and configurations can be implemented by one of ordinary skill in the art without departing from the spirit of the present invention.  
         [0055]     An antenna arrangement is illustrated in  FIG. 3A  as one embodiment of the multi-dimensional thinned transmit antenna network  601 , as one embodiment of the multi-dimensional thinned receive antenna network  621 , as one embodiment of the mechanically scanned thinned transmit antenna network  601   b , and as one embodiment of the mechanically scanned thinned receive antenna network  621   b  according to aspects of the present invention. In this arrangement, a plurality of antennas  178 ,  179  are connected to the u transmit signals and/or v receive signals as defined in FIGS.  1 A-B. The antennas can be arranged in a one-dimensional array, two-dimensional array, a conformal array, or a multi-dimensional array according to aspects of the present invention. The antennas can each have similar characteristics to one another, or can have different characteristics from one another depending on the requirements of the application. In addition, the antennas can have a polarization such as, but not limited to, linear polarization, circular polarization, or dual polarization according to aspects of the present invention.  
         [0056]     An antenna arrangement is illustrated in  FIG. 3B  as another embodiment of the multi-dimensional thinned transmit antenna network  601 , as another embodiment of the multi-dimensional thinned receive antenna network  621 , as another embodiment of the mechanically scanned thinned transmit antenna network  601   b , and as another embodiment of the mechanically scanned thinned receive antenna network  621   b  according to aspects of the present invention. In this arrangement, a selector  112  selectively establishes a connection between each of a plurality of antennas  180 ,  181  and a common input or output connection depending on whether the selector is used for a transmit or receive application respectively. In this way, this arrangement can be used to sequentially select between a number of antenna elements, and can be utilized to enable electrical sequencing or scanning of antenna arrays. A selector  112  can be used with each or any of the u transmit signals and/or v receive signals as defined in FIGS.  1 A-B. Selector  112  can be implemented by, but is not limited to, a switch or a combination of switches, variable attenuators, or a combination of switched amplifiers and signal combiners/splitters wherein switching the gain/loss of said amplifiers is used for the selection function and said signal combiners/splitters can be implemented by, but are not limited to, Wilkinson combiners/splitters. One advantage of using switched amplifiers and signal combiners/splitters as a selection means is the elimination of the signal loss associated with series selection switches. The antennas can each have similar characteristics to one another, or can have different characteristics from one another depending on the requirements of the application. The antennas can be arranged in a one-dimensional array, two-dimensional array, a conformal array, or a multi-dimensional array according to aspects of the present invention. In addition, the antennas can have a polarization such as, but not limited to, linear polarization, circular polarization, or dual polarization according to aspects of the present invention.  
         [0057]     An antenna arrangement is illustrated in  FIG. 3C  as a further embodiment of the multi-dimensional thinned transmit antenna network  601 , as a further embodiment of the multi-dimensional thinned receive antenna network  621 , as a further embodiment of the mechanically scanned thinned transmit antenna network  601   b , and as a further embodiment of the mechanically scanned thinned receive antenna network  621   b  according to aspects of the present invention. In this arrangement, a plurality of selectors  114 ,  116  are used to select between antennas in a plurality of antenna groups. Selector  114  selectively establishes a connection between each of the plurality of antennas  183 ,  185  in one antenna group and a common input or output connection depending on whether the selector is used for a transmit or receive application respectively. Similarly, selector  116  selectively establishes a connection between each of the plurality of antennas  187 ,  189  in another antenna group and a common input or output connection depending on whether the selector is used for a transmit or receive application respectively. In this way, this arrangement can be used to sequentially select between a number of antenna elements, and can be utilized to enable electrical sequencing or scanning of antenna arrays. Selectors  114 ,  116  can be used with each or any of the u transmit signals and/or v receive signals as defined in FIGS.  1 A-B. Selectors  114 ,  116  can be implemented by, but are not limited to, switches or a combination of switches, variable attenuators, or combinations of switched amplifiers and signal combiners/splitters. The antennas can each have similar characteristics to one another, or can have different characteristics from one another depending on the requirements of the application. The antennas can be arranged in a one-dimensional array, two-dimensional array, a conformal array, or a multi-dimensional array according to aspects of the present invention. In addition, the antennas can have a polarization such as, but not limited to, linear polarization, circular polarization, or dual polarization according to aspects of the present invention.  
         [0058]     In addition, the multi-dimensional thinned transmit antenna network  601  and the multi-dimensional thinned receive antenna network  621  can share one or a plurality of antennas according to aspects of the present invention. Furthermore, the mechanically scanned thinned transmit antenna network  601   b  and the mechanically scanned thinned receive antenna network  621   b  can share one or a plurality of antennas according to aspects of the present invention.  
         [0059]     An imaging sensor arrangement is presented in  FIG. 4A  as one embodiment of aspects of the present invention. In this arrangement, a signal generated by the signal generator  405  is split by a signal splitter  27 , where one portion of the signal proceeds to an amplifier  30  where it is amplified prior to proceeding to a selector  501 . The selector  501  is used to selectively connect the signal to one of a plurality of an antennas  101   a ,  101   b , designated by TX  1 , 1 , TX m,n, where m and n are non-zero integers whose sum is greater than or equal to 3, for transmission in a sequential manner. A signal designated as TX_SEL controls which antenna  101   a ,  101   b  is selected by selector  501 . A typical frequency of the transmission signal can be within, but is not limited to, the frequency range of 1 GHz-1 THz, and can be a fixed frequency or be frequency modulated. The imaging sensor&#39;s total occupied transmit spectral bandwidth is dependent on the frequency modulation bandwidth, and can be wideband (WB) or ultra-wideband (UWB) in order to achieve adequate range resolution for some applications. A typical WB bandwidth value can be, but is not limited to, a value greater than 100 MHz. A typical UWB bandwidth value can be, but is not limited to, a value greater than 1 GHz. The arrangement of the antennas  101   a ,  101   b  can be, but is not limited to, a one-dimensional array, a two-dimensional array, a three-dimensional array, multiple one-dimensional arrays arranged in multiple axes, or a conformal array. The reflected signal from an object is received by a plurality of receive antennas  102   a ,  102   b , designated by RX  1 , 1 , RX k,p, where k and p are non-zero integers whose sum is greater than or equal to 3. The arrangement of the receive antennas  102   a ,  102   b  can be, but is not limited to, a one-dimensional array, a two-dimensional array, a three-dimensional array, multiple one-dimensional arrays arranged in multiple axes, or a conformal array. A selector  502  is used to selectively connect one receive antenna at a time with the low noise amplifier  62  where the received signal is amplified prior to being split by splitter  28 . A signal designated as RX_SEL controls which antenna  102   a ,  102   b  is selected by selector,  502 . One of the outputs from splitter  28  is input to mixer  55 , which mixes the signal with the 0-degree phase output signal from the 90-degree splitter  77   a , and the other output from splitter  28  is input to mixer  56 , which mixes the signal with the 90-degree phase output signal from the 90-degree splitter  77   a , creating in-phase (I) and quadrature (Q) down-converted signals. The I and Q down-converted signals are then amplified by amplifiers  65 ,  66  and filtered by filters  45 ,  46  prior to sampling by A/D converters  340 ,  341 . The resulting sampled I and Q signals are then input to signal processor  300  for signal processing.  
         [0060]     The block diagram shown in  FIG. 4A  can be modified according to aspects of the present invention. One example of such a modification, not meant as a limitation, can be to not perform complex (I and Q) signal down-conversion or to perform it digitally in the signal processor, only having one down-converting mixer path to a single A/D converter, and to modify the block diagram accordingly. Another example of such a modification, not meant as a limitation, can be for the sensor architecture to use remote signal processing, remote analog-to-digital (A/D) conversion, or shared processing and/or A/D conversion with another sensor or system. A further example of such a modification, not meant as a limitation, can be for the sensor architecture to replace one or both of the selectors  501 ,  502  with a plurality of switched amplifiers and signal combiners, utilizing the gain/loss of the switched amplifiers to realize an antenna selection and routing function. A yet further example of such a modification, not meant as a limitation, can be for the sensor architecture to utilize any of the antenna networks illustrated in FIGS.  3 A-C for any or both of the transmit or receive selectors and antenna functions. Another example of such a modification, not meant as a limitation, can be for the sensor architecture to use a plurality of simultaneously selected transmit signals and/or a plurality of simultaneously selected receive signals connected to a plurality of receiver/down-converter circuits. Mixers  55 ,  56  can be implemented by, but are not limited to, mixers, multipliers, or switches without changing the basic functionality of the arrangement. Filters  45 ,  46  can be implemented by, but are not limited to, low-pass filters or band-pass filters. Signal splitters  27 ,  28  can be implemented by, but are not limited to, Wilkinson power dividers, passive splitters, active splitters, or microwave couplers. A variety of amplifiers, filters, or other system elements known to those skilled in the art, such as low-noise amplifiers, power amplifiers, drivers, buffers, gain blocks, gain equalizers, logarithmic amplifiers, equalizing amplifiers, switches, and the like, can be added to or deleted from the described arrangement, or the position of existing elements may be modified, without changing the basic form or spirit of the invention.  
         [0061]     Signal processor  300  shown in  FIG. 4A  may comprise a single or plurality of individual processors. Signal processor  300  may perform, but is not limited to, any single or combination of the functions of signal or image processing, real or complex DFT or FFT signal processing, CFAR threshold detection, spectral peak detection, target peak association, frequency measurement, magnitude measurement, phase measurement, magnitude scaling, phase shifting, spatial FFT processing, digital beam-forming (DBF) processing, super-resolution processing such as, but not limited to, the use of the MUSIC or ESPRIT algorithms, neural network processing, two-dimensional image processing, three-dimensional image processing, two or three-dimensional image reconstruction processing, microwave or millimeter-wave holography processing, backward-wave reconstruction processing, wavefront reconstruction processing, synthetic aperture radar (SAR) processing, or Kirchoff diffraction integral processing. Additional processing techniques that can be used with the abovementioned methods may include, but are not limited to, windowing, digital filtering, Hilbert transform, least squares algorithms, or non-linear least squares algorithms. Furthermore, one or a combination of object signature methods can be used to determine the presence or identification of potential threats, weapons or contraband such as, but not limited to, radial cross-section characteristics, angular cross-section characteristics, strength of returns, wideband or ultra-wideband frequency response characteristics, wideband or ultra-wideband frequency resonance characteristics, polarization response characteristics, or image shape characteristics, and such signatures may be determined for the entire object or for one or more regions of an object. In addition, the object signature methods can utilize complex signal attributes such as amplitude and/or phase. The signal processor may include, but is not limited to, one or more digital signal processors (DSPs), microprocessors, micro-controllers, electrical control units, or other suitable processor blocks.  
         [0062]     An imaging sensor arrangement is presented in  FIG. 4B  as another embodiment of the present invention. The arrangement in  FIG. 4B  is similar to the arrangement in  FIG. 4A , except for the addition of a transmission pulsing switch  8 , a receiver gating switch  9 , and the omission of amplifiers  62 ,  30  for clarity. The same components are denoted by the same reference numerals, and will not be explained again. In this configuration, the TX PULSE CONTROL signal is used to control the operation of a transmission pulsing switch  8 , pulse modulating the output signal. The RX GATE CONTROL signal is used to control the operation of the receiver gating switch  9 , which only allows received signals to pass through for down-conversion during specified time periods dictated by the RX GATE CONTROL signal. Through the use of this arrangement of transmit pulsing and receive signal gating, the performance of the sensor can be improved as illustrated in the signal timing example in  FIG. 5 .  
         [0063]     One example of pulsed transmit and gated receiver signal timing for an imaging sensor is shown in  FIG. 5  in accordance with aspects of the present invention. The timing diagram shown in  FIG. 5  is meant as an example to illustrate the operation and potential benefits of pulsed transmission and gated reception, and is not meant as a limitation. In this example, during the time period τ 1 , the antenna pair consisting of transmit antenna TX  1 , 1  and receive antenna RX  1 , 1  is selected by use of the signals TX_SEL and RX_SEL, followed by a pulse of the transmit signal by use of the TX PULSE CONTROL signal, and a subsequent gating of the receiver after some time delay by use of the RX GATE CONTROL signal. The gating “on” time of the receiver corresponding to the “on” state of the RX GATE CONTROL signal as shown in  FIG. 5  can be matched to the transmit pulse “on” time corresponding to the “on” state of the TX PULSE CONTROL signal as shown in  FIG. 5 , and is configured that way for this example. Also shown in  FIG. 5  is an example of the output envelope of a typical matched filter that could be utilized for filters  45 ,  46  in  FIG. 4B  in the receiver, and I and Q A/D sampling at the peak of the matched filter output envelope that could be utilized by A/D converters  340 ,  341  in  FIG. 4B  for optimal signal-to-noise-ratio performance. The pulsing of the transmit signal and gating of the received signal allows the sensor to selectively receive object returns in a range zone between a specific minimum range (Rmin) and maximum range (Rmax), related to the time delay between transmit pulse and receive gate and the time durations of each, and to reject object returns that occur at ranges less than Rmin and ranges greater than Rmax. This operation allows rejection of signals such as, but not limited to, signals coupling directly from the transmitter to the receiver, radome returns, near-field clutter, and far-field clutter. In addition, this operation can give the ability to design spatial selectivity to the range of detection for a particular application or scenario, and can be used to eliminate multi-path reflections from near-field objects. A variety of modifications can be made to the sensor timing shown in  FIG. 5  by those skilled in the art, such as, but not limited to, the order of antenna pair selection or the number of transmit pulses and receive gates per antenna pair dwell time without changing the basic form or spirit of the invention.  
         [0064]     An imaging sensor arrangement is presented in  FIG. 6A  as a further embodiment of aspects of the present invention. The arrangement in  FIG. 6A  is similar to the arrangement in  FIG. 4A , except for the replacement of signal generator  405  with TX &amp; LO signal generator  407 , the addition of an IF frequency reference  70 , and modification of the down-conversion circuitry used to create in-phase (I) and quadrature (Q) signals prior to signal A/D conversion. The same components are denoted by the same reference numerals, and will not be explained again. In this arrangement, one signal generated by the TX &amp; LO signal generator  407  designated by TX is fed to an amplifier  30  where it is amplified prior to transmission. The other signal generated by the TX &amp; LO signal generator  407 , designated by LO, has a frequency which is offset from the frequency of the TX signal by an amount equal to the frequency of IF frequency reference  70 , and is fed to the mixer  55  where it is mixed with the received signal output from amplifier  62 ′. A typical frequency used for the IF frequency reference  70  can be within, but is not limited to, the frequency range of 1 MHz-500 MHz. The output signal from mixer  55  is then input to filter  39 , and the output signal from filter  39  is split and input to mixers  85  and  86 . One of the outputs from filter  39  is input to mixer  85 , which mixes the signal with the 90-degree phase output signal from 90-degree splitter  77   b , and the other output from filter  39  is input to mixer  86 , which mixes the signal with the 0-degree phase output signal from 90-degree splitter  77   b , creating in-phase (I) and quadrature (Q) down-converted signals. The I and Q down-converted signals are then filtered by filters  36 ,  35 , respectively, prior to sampling by A/D converters  340 ,  350 . The resulting sampled I and Q signals are then input to signal processor  300 . Through the use of this arrangement of intermediate frequency conversion, the noise associated with the down-conversion process can be improved.  
         [0065]     An imaging sensor arrangement is presented in  FIG. 6B  as a yet further embodiment of aspects of the present invention. The arrangement in  FIG. 6B  is similar to the arrangement in  FIG. 4A , except for the replacement of selectors  501 ,  502  and associated antennas with polarization selectors  510 ,  520 , antenna selectors  503 ,  504 ,  505 ,  506  and associated antennas  103   a ,  103   b ,  104   a ,  104   b ,  105   a ,  105   b ,  106   a ,  106   b , and the omission of amplifiers  62 , for clarity. The same components are denoted by the same reference numerals, and will not be explained again. In this arrangement, one signal from splitter  27  is input to a polarization selector  510 , which outputs the signal to either selector  503  or  504  according to a control signal designated as TX_POL_SEL. The selector  503  is used to selectively connect a transmission signal to one of a plurality of antennas  103   a ,  103   b  which have a certain polarization, designated by TX-P 1   1 , 1 , TX-P 1  m,n, where m and n are non-zero integers whose sum is greater than or equal to 3, for transmission in a sequential manner. The selector  504  is used to selectively connect a transmission signal to one of a plurality of an antennas  104   a ,  104   b  which have a polarization different than that of antennas  103   a ,  103   b , designated by TX-P 2   1 , 1 , TX-P 2  m,n, where m and n are non-zero integers whose sum is greater than or equal to 3, for transmission in a sequential manner. A typical frequency of the transmission signal can be within, but is not limited to, the frequency range of 1 GHz-1 THz, and can be a fixed frequency or be frequency modulated. The imaging sensor&#39;s total occupied transmit spectral bandwidth is dependent on the frequency modulation bandwidth, and can be wideband (WB) or ultra-wideband (UWB) in order to achieve adequate range resolution for some applications. A typical WB bandwidth value can be, but is not limited to, a value greater than 100 MHz. A typical UWB bandwidth value can be, but is not limited to, a value greater than 1 GHz. The arrangement of the antennas  103   a ,  103   b  can be, but is not limited to, a one-dimensional array, a two-dimensional array, a three-dimensional array, multiple one-dimensional arrays arranged in multiple axes, or a conformal array, and can have a polarization that is, but not limited to, vertical, horizontal, or circular. The arrangement of the antennas  104   a ,  104   b  can be, but is not limited to, a one-dimensional array, a two-dimensional array, a three-dimensional array, multiple one-dimensional arrays arranged in multiple axes, or a conformal array, and can have a polarization that is, but not limited to, linear, vertical, horizontal, or circular. The reflected signal from an object is received by a plurality of receive antennas  105   a ,  105   b , designated by RX-P 1   1 , 1 , RX-P 1  k,p, where k and p are non-zero integers whose sum is greater than or equal to 3, and a plurality of receive antennas  106   a ,  106   b , designated by RX-P 2   1 , 1 , RX-P 2  k,p, where k and p are non-zero integers whose sum is greater than or equal to 3. Antennas  105   a ,  105   b  have the same polarization as antennas  103   a ,  103   b , and antennas  106   a ,  106   b  have the same polarization as antennas  104   a ,  104   b . The arrangement of the antennas  105   a ,  105   b  can be, but is not limited to, a one-dimensional array, a two-dimensional array, a three-dimensional array, multiple one-dimensional arrays arranged in multiple axes, or a conformal array, and can have a polarization that is, but not limited to, linear, vertical, horizontal, or circular. The arrangement of the antennas  106   a ,  106   b  can be, but is not limited to, a one-dimensional array, a two-dimensional array, a three-dimensional array, multiple one-dimensional arrays arranged in multiple axes, or a conformal array, and can have a polarization that is, but not limited to, vertical, horizontal, or circular. A selector  505  is used to selectively connect one receive antenna  105   a ,  105   b  at a time with one input of polarization selector  520 . A selector  506  is used to selectively connect one receive antenna  106   a ,  106   b  at a time with the other input of polarization selector  520 . The polarization selector  520  is used, to selectively connect one receiver antenna of a certain polarization and a certain spatial position at a time with the receiver/down-converter circuitry for the sensor in a sequential manner. A signal designated as RX_POL_SEL controls which selector  505 ,  506  is selected by polarization selector  520 . Through the use of this arrangement, the response of objects to signals having multiple polarizations can be sampled and utilized for image processing and/or object identification.  
         [0066]     The block diagram shown in  FIG. 6B  can be modified according to aspects of the present invention. One example of such a modification, not meant as a limitation, can be for the sensor architecture to replace any or all of the selectors  503 ,  504 ,  505 ,  506 ,  510 ,  520  with a plurality of switched amplifiers and signal combiners, utilizing the gain/loss of the switched amplifiers to realize an antenna selection and routing function. Another example of such a modification, not meant as a limitation, can be for the sensor architecture to utilize any of the antenna networks illustrated in FIGS.  3 A-C for any or both of the transmit or receive selectors and antenna functions. A further example of such a modification, not meant as a limitation, can be for the sensor architecture to use a plurality of simultaneously selected transmit signals and/or a plurality of simultaneously selected receive signals connected to a plurality of receiver/down-converters. A yet further example of such a modification, not meant as a limitation, can be for the sensor architecture to utilize antenna elements that are dual-polarized, such that selectors  503 ,  504  feed only one set of dual-polarized antenna elements, and selectors  505 ,  506  feed only one set of dual-polarized antenna elements. Another example of such a modification, not meant as a limitation, can be for the sensor architecture to share one or a plurality of antennas between transmit and receive functions. A further example of such a modification, not meant as a limitation, can be for the sensor architecture to utilize a two-stage down-conversion structure such as illustrated in  FIG. 6A . A variety of amplifiers, filters, or other system elements known to those skilled in the art, such as low-noise amplifiers, power amplifiers, drivers, buffers, gain blocks, gain equalizers, logarithmic amplifiers, equalizing amplifiers, switches, and the like, can be added to or deleted from the described arrangement, or the position of existing elements may be modified, without changing the basic form or spirit of the invention.  
         [0067]     One example of a two-dimensional, dual-polarized thinned-array is illustrated in  FIG. 6C  according to aspects of the present invention. The configuration shown is meant as an illustration of a dual-polarized thinned-array, not as a limitation. In this configuration, a TX arrangement  193 , containing a transmit antenna array having a polarization P 1  and a transmit antenna array having a polarization P 2 , and an RX arrangement  197 , containing a receive antenna array having a polarization. P 1  and a receive antenna array having a polarization P 2 , are positioned diagonally. The polarization P 1  can be, but is not limited to, linear, vertical, horizontal, or circular. The polarization P 2  can be, but is not limited to, linear, vertical, horizontal, or circular. In this arrangement, the TX arrangement  193  and RX arrangement  197  are illustrated to be diagonal to one another, where the rows of the TX arrangement  193  span a range in the x-axis that is non-overlapping with the span of the rows of the RX arrangement  197  in the x-axis, and the columns of the TX arrangement  193  span a range in the y-axis that is non-overlapping with the span of the columns of the RX arrangement  197  in the y-axis. The 3 by 3 element P 1  polarized transmit and receive arrays can synthesize a 9 by 9 P 1  polarized virtual array using the method described in  FIGS. 2D &amp; 2E . Similarly, the 3 by 3 element P 2  polarized transmit and receive arrays can synthesize a 9 by 9 P 2  polarized virtual array using the method described in  FIGS. 2D &amp; 2E . Utilizing this configuration, each virtual array can be processed separately to generate images of object responses to each of the polarizations. The digitized, down-converted signals can be utilized for object imaging, through the use of image processing techniques well known in the art such as, but not limited to, digital beam-forming (DBF) processing, super-resolution processing such as, but not limited to, the use of the multiple signal classification algorithm (MUSIC) or the estimation of signal parameters via rotational invariance techniques (ESPRIT) algorithm, spatial Fourier transform processing, two-dimensional image processing, three-dimensional image processing, two or three-dimensional image reconstruction processing, microwave or millimeter-wave holography processing, backward-wave reconstruction processing, wavefront reconstruction processing, synthetic aperture radar (SAR) processing, or Kirchoff diffraction integral processing. The example shown is meant as an illustration of a dual-polarized virtual array synthesis technique, not as a limitation. For example, not meant as a limitation, the distance between elements in each array need not be constant, but can be varied or be given multiple different values by one skilled in the art for advantage. Other array sizes and configurations can be implemented by one of ordinary skill in the art without departing from the spirit of the present invention.  
         [0068]     According to one aspect of the present invention, the use of multiple selectable polarizations can be used for generation of object polarization signatures and utilized for object detection and/or identification purposes. According to another aspect of the present invention, the angular resolution provided by imaging techniques such as, but not limited to, digital beam-forming can provide spatial selectivity for object signatures as well as spatial rejection of other object signatures or clutter signals for improved performance and object identification capability. The object signature methods that can be used with the spatial selectivity methods described to determine the presence or identification of potential threats, weapons or contraband can include, but are not limited to, strength of returns, wideband or ultra-wideband frequency response characteristics, wideband or ultra-wideband frequency resonance characteristics, polarization response characteristics, spectral absorption characteristics, or image shape characteristics. In addition, a combination of object imaging and spatially isolated regional scanning for weapons signatures can be utilized in order to provide additional capability or performance. Furthermore, the beam-width or area of the spatially isolated regions utilized for detection of weapons signatures can be different than the resolution utilized for object imaging, and the techniques utilized for object imaging and scanning of spatially isolated regions need not be the same. For example, not meant as a limitation, a high resolution object image can be generated utilizing a two-dimensional image reconstruction technique for the purpose of providing image characteristics for image processing, while a lower resolution spatially isolated beam could be generated by a digital beam-forming process and scanned across areas of the object in order to utilize weapons signature techniques for detection and/or identification of concealed weapons. Additionally, the resolution of the image and/or the size of the spatially isolated region can be varied adaptively. Furthermore, the area of the spatially isolated region can encompass a part of an object in order to isolate weapons signatures from other parts of the object, or can encompass the entire object in order to isolate weapons signatures from the surroundings of the object.  
         [0069]     In accordance with one aspect of the present invention, the millimeter-wave imaging techniques and/or weapons signature techniques can be combined with an image generated by another sensor such as, but not limited to, an optical wavelength camera. For example, not meant as a limitation, an optical wavelength image of an object can be enhanced by the addition of indicators added to the optical image at locations where threats or contraband is suspected to be. The indicators can include, but are not limited to, colored shapes where the color indicates threat or confidence level and/or the shape indicates type of threat, text indicating a threat type with an arrow pointing to a location on the object in the optical image, or any combination of these. One benefit of this arrangement is that the optical image can be utilized additionally for identification of the object such as, but not limited to, the identification of a person carrying the concealed threat. Another benefit of this arrangement is that if the millimeter-wave image is not shown to the operator, then privacy concerns for the individual being scanned may be avoided. Indicator types other than the ones presented can be utilized without departing from the spirit of the present invention.  
         [0070]     Another aspect of the present invention is the utilization of the electrically sequenced or scanned virtual array arrangement for through-wall imaging. For example, not meant as a limitation, the electrically sequenced or scanned virtual array can be utilized to provide a 2D or 3D image of the interior of a room from behind a door or wall of the room. The digital lensing and image reconstruction methods can be adapted to additionally compensate for the characteristics of the medium of the wall or door though which the electro-magnetic waves propagate.  
         [0071]     One embodiment of signal generator  405  is shown in  FIG. 7A . In this configuration, a frequency controller  410  controls the frequency of a transmit voltage-controlled-oscillator  90 . The embodiment shown in  FIG. 7A  represents an open-loop transmit signal generator configuration. The configuration shown is meant as an illustration of a transmit signal generation technique, not as a limitation. Other open-loop signal generation techniques can be implemented by one of ordinary skill in the art without departing from the spirit of the present invention.  
         [0072]     Another embodiment of signal generator  405  is shown in  FIG. 7B . In this configuration, the output of a frequency controller  430  controls the frequency of a transmit voltage-controlled-oscillator  90 . The output signal from the transmit voltage-controlled-oscillator  90  is split by signal splitter  411 , where one portion of the signal is output, and the other portion of the signal is fed back to the frequency controller  430 , where it is used to monitor and adjust the frequency of the transmit voltage-controlled-oscillator  90 , forming a closed-loop transmit signal generator.  
         [0073]     A further embodiment of signal generator  405  is shown in  FIG. 7C . In this configuration, the output of a phase-locked loop (PLL)  465  is filtered by loop filter  421  and used to control the frequency of a transmit voltage-controlled-oscillator (TX VCO)  90 . The PLL  465  can be implemented by, but is not limited to, a phase detector, phase-frequency detector, integer-N PLL, or fractional-N PLL. The output from TX VCO  90  is split by splitter  411 , where one portion of the signal is output and the other portion of the signal is frequency divided by N by divider  417 , where N is an integer greater than 1, and fed back to the PLL  465  forming a closed-loop transmit signal generator. A frequency reference  444  is input to the PLL  465 , and the PLL  465  can be controlled by an external control signal if required.  
         [0074]     A yet further embodiment of signal generator  405  is shown in  FIG. 7D . In this configuration, the output of a PLL  465  is filtered by loop filter  421  and used to control the frequency of a transmit voltage-controlled-oscillator (TX VCO)  90 . The output from TX VCO  90  is split by splitter  411 , where one portion of the signal is output and the other portion of the signal is frequency divided by N by divider  417 , where N is an integer greater than 1, and fed back to the PLL  465  forming a closed-loop transmit signal generator. A direct-digital-synthesizer (DDS)  482  is input as a frequency reference to the PLL  465 . Through the control of the output frequency of the DDS  482 , the frequency of the TX VCO  90  can be controlled.  
         [0075]     Another embodiment of signal generator  405  is shown in  FIG. 7E . The arrangement in  FIG. 7E  is similar to the arrangement in  FIG. 7C , except for the use of a frequency multiplier  573  at the output of transmit voltage-controlled-oscillator (TX VCO)  90 . The same components are denoted by the same reference numerals, and will not be explained again. The use of a frequency multiplier  573  allows the frequency of TX VCO  90  to be lower than the output transmit frequency of the signal generator  405 .  
         [0076]     The arrangement shown in  FIG. 7E  can be modified according to aspects of the present invention. One example of such a modification, not meant as a limitation, can be for the frequency reference  444  to be replaced by a DDS  482 , such as described in the arrangement of  FIG. 7D . Other modifications can be implemented by one of ordinary skill in the art without departing from the spirit of the present invention.  
         [0077]     One embodiment of TX &amp; LO signal generator  407  is shown in  FIG. 7F . In this configuration, the output of a PLL  465  is filtered by loop filter  421  and used to control the frequency of a transmit voltage-controlled-oscillator (TX VCO)  90 . The output from TX VCO  90  is split by splitter  411 , where one portion of the signal is fed to splitter  412 , while the other portion of the signal is frequency divided by N by divider  417 , where N is an integer greater than 1, and fed back to the PLL  465  forming a closed-loop transmit signal generator. A direct-digital-synthesizer (DDS)  482  is input as a frequency reference to the PLL  465 . Through the control of the output frequency of the DDS  482 , the frequency of the TX VCO  90  frequency can be controlled. The output from splitter  411  is split by splitter  412 , where one portion of the signal is output as the signal designated by TX, while the other portion of the signal is fed to mixer  59 , where it is mixed with an IF frequency reference signal. The output from mixer  59  is filtered by filter  426  and output as the signal designated by LO.  
         [0078]     Another embodiment of TX &amp; LO signal generator  407  is shown in  FIG. 7G . In this configuration, the output of a PLL  465  is filtered by loop filter  421  and used to control the frequency of a transmit voltage-controlled-oscillator (TX VCO)  90 . The output from TX VCO  90  is split by splitter  411 , where one portion of the signal is output as the signal designated by TX, and the other portion of the signal is frequency divided by N by divider  417 , where N is an integer greater than 1, and fed back to the PLL  465  forming a closed-loop transmit signal generator. A direct-digital-synthesizer (DDS)  482  is input as a frequency reference to the PLL  465 . An IF frequency reference is input to the DDS  482  as a frequency reference for the DDS. A second PLL  465   b  is filtered by loop filter  421   b  and used to control the frequency of a local oscillator voltage-controlled-oscillator (LO VCO)  90   b . The output from LO VCO  90   b  is split by splitter  411   b , where one portion of the signal is output as the signal designated by LO, while the other portion of the signal is frequency divided by N by divider  417   b , where N is an integer greater than 1, and fed back to the PLL  465   b  forming a closed-loop local oscillator signal generator. A direct-digital-synthesizer (DDS)  482   b  is input as a frequency reference to the PLL  465   b . An IF frequency reference is input to the DDS  482   b  as a frequency reference for the DDS. The DDS  482   b  is programmed to have a frequency offset from the DDS  482  such that the TX output signal and LO output signal are offset in frequency an amount equal to the IF frequency reference frequency.  
         [0079]     The embodiments shown in FIGS.  7 A-G represent examples of signal generation configurations. The configurations shown are meant as an illustration of signal generation techniques, not as a limitation. Other signal generation techniques can be implemented by one of ordinary skill in the art without departing from the spirit of the present invention.  
         [0080]      FIG. 8A  illustrates a linearly frequency-modulated waveform for use in the transmit signal generator  650 , signal generator  405  or TX &amp; LO signal generator  407  according to aspects of the present invention. This waveform shows a linearly modulated frequency with a period equal to Tp. This waveform shown is an example of linear frequency modulation and is not meant as a restriction. The waveform can also comprise, but is not limited to, a repeating pattern of linearly increasing frequency ramps, a repeating pattern of linearly decreasing frequency ramps, or alternating periods of linearly increasing and decreasing frequency ramps. Also, periods where the frequency modulation is stopped may be inserted into the abovementioned patterns. Furthermore, in order to achieve adequate range resolution for some applications, the total frequency modulation bandwidth, defined as |f 2 −f 1 | in  FIG. 8A , can be wideband (WB) or ultra-wideband (UWB).  
         [0081]     Using the frequency modulation waveform described in  FIG. 8A , object range information may be calculated from the down-converted signals of the architectures shown in FIGS.  1 A-C, FIGS.  4 A-B and FIGS.  6 A-B in the following way. Peaks in the down-converted signal spectrum represent returns from objects within the field of view. The frequency of the peaks is proportional to object range and is used to calculate object range. As an example, not meant as a limitation, let the arrangement of  FIG. 4A  utilize a linearly increasing frequency modulation as shown in  FIG. 8A . Let the down-converted signal be sampled &amp; measured during each coherent measurement interval T P . Under these conditions, object range can be calculated by the following equation:  
             R   =         c   ·     T   P         2   ·     (       f   2     -     f   1       )         ·     (     f   B     )               (   1   )             
 
 where R is the calculated object range, c is the speed of light in a vacuum, f 2  is the maximum frequency of the linear modulation, f 1  is the minimum frequency of the linear modulation, and f B  is the beat frequency in the down-converted signal corresponding to measurements during the coherent measurement interval T P . The object range data calculated using this method can be utilized to generate three-dimensional object images through use with methods well known in the art, such as, but not limited to, digital beam-forming angular processing or super-resolution angular processing. 
 
         [0082]     Another approach to calculating object range data is to use an inverse fast Fourier transform (IFFT) or inverse discrete Fourier transform (IDFT), after sampling the down-converted signal, to build an object return range profile. The peaks in the IFFT or IDFT profile represent object returns with range proportional to the peak&#39;s associated time bin. The object range data calculated using this method can be utilized to generate three-dimensional object images through use with methods well known in the art, such as, but not limited to, digital beam-forming angular processing or super-resolution angular processing, which will be described in more detail in the following text.  
         [0083]      FIG. 8B  illustrates a stepped frequency modulation waveform for use in the transmit signal generator  650 , signal generator  405  or TX &amp; LO signal generator  407  according to aspects of the present invention. This waveform shows a linearly stepped frequency pattern with a frequency increasing step sequence period equal to T P . This waveform shown is an example of linearly stepped frequency modulation and is not meant as a restriction. A typical value of Δf S  can be within, but is not limited to, the range of 100 KHz-100 MHz. A typical value of T S  can be within, but is not limited to, the range of 500 nanoseconds (ns)-20 microseconds (μs). The waveform can also comprise, but is not limited to, a repeating pattern of linearly increasing frequency steps, a repeating pattern of linearly decreasing frequency steps, or alternating periods of linearly increasing and decreasing frequency step patterns. Also, periods where the stepped frequency modulation pattern is stopped may be inserted into the abovementioned patterns. In addition, the value of T S  may be varied or dithered, or the linearity of the frequency steps with respect to time may be varied by one skilled in the art without departing from the spirit of the present invention. Furthermore, in order to achieve adequate range resolution for some applications, the total frequency modulation bandwidth, defined as |f 2 −f 1 | in  FIG. 8B  can be wideband (WB) or ultra-wideband (UWB).  
         [0084]     Using the frequency modulation waveform described in  FIG. 8B , object range information may be calculated from the down-converted signals of the architectures shown in FIGS.  1 A-C, FIGS.  4 A-B and FIGS.  6 A-B in the following way. Peaks in the down-converted signal spectrum represent returns from objects within the field of view. The frequency of the peaks is proportional to object range and is used to calculate object range. As an example, not meant as a limitation, let the arrangement of  FIG. 4A  utilize a linearly increasing frequency step sequence as shown in  FIG. 8B . Let the down-converted signal be sampled &amp; measured during each coherent measurement interval T P , which for this example also corresponds to the frequency-modulated step sequence period. Under these conditions, object range can be calculated by the following equation:  
             R   =         c   ·     T   S           2   ·   Δ     ⁢           ⁢     f   S         ·     (     f   B     )               (   2   )             
 
 where R is the calculated object range, c is the speed of light in a vacuum, T S  is dwell time of each frequency step, Δf S  is the difference between adjacent frequency step values in the linear step sequence, and f B  is the beat frequency in the down-converted signal corresponding to measurements during the frequency-stepped sequence period T P . The object range data calculated using this method can be utilized to generate three-dimensional object images through use with methods well known in the art, such as, but not limited to, digital beam-forming angular processing or super-resolution angular processing. 
 
         [0085]     Another approach to calculating object range data is to use an inverse fast Fourier transform (IFFT) or inverse discrete Fourier transform (IDFT), after sampling the down-converted signal, to build an object return range profile. The peaks in the IFFT or IDFT profile represent object returns with range proportional to the peak&#39;s associated time bin. The object range data calculated using this method can be utilized to generate three-dimensional object images through use with methods well known in the art, such as, but not limited to, digital beam-forming angular processing or super-resolution angular processing which will be described in more detail in the following text.  
         [0086]      FIG. 8C  illustrates a multiple-slope, linearly frequency-modulated waveform for use in the transmit signal generator  650 , signal generator  405  or TX &amp; LO signal generator  407  according to aspects of the present invention. This waveform shows a linear up-slope frequency modulation during a first time period Tp, and a linear down-slope frequency modulation during a second time period Tp. This waveform shown is an example of frequency modulation, and is not meant as a restriction. A typical value of Tp can be within, but is not limited to, the range of 100 microseconds (μs)-100 milliseconds (ms). The frequency modulation can also consist of, but is not limited to, a repeating pattern of linear up-slope modulation, a repeating pattern of linear down-slope modulation, an alternating pattern of up- and down-slope modulation, a monotonically increasing frequency over a time period, a monotonically decreasing frequency over a time period, or an alternating pattern of monotonically increasing and decreasing frequency modulation. In addition, one or more blanking periods where the frequency is constant may be inserted within or between the up or down slope periods. Furthermore, in order to achieve adequate range resolution for some applications, the total frequency modulation bandwidth, defined as |f 2 −f 1 | in  FIG. 8C  can be wideband (WB) or ultra-wideband (UWB).  
         [0087]     Using the frequency modulation waveform described in  FIG. 8C , object information may be calculated from the down-converted signals of the architectures shown in FIGS.  1 A-C, FIGS.  4 A-B and FIGS.  6 A-B in the following way. Peaks in the down-converted signal spectrum represent object returns. The frequency of the peaks is proportional to object range, and is used to calculate object range. As an example, not meant as a limitation, let the sensor arrangement of  FIG. 4A  utilize a frequency modulation according to  FIG. 8C . Let the down-converted signal be sampled &amp; measured during each coherent measurement interval T P , which also corresponds in this example to the frequency up-ramp and down-ramp periods. Under these conditions, object range can be calculated by the following equation:  
             R   =         c   ·     T   P           4   ·   Δ     ⁢           ⁢     f   BW         ·     (       f   U     +     f   D       )               (   3   )             
 
 where R is the calculated object range, c is the speed of light in a vacuum, T P  is the period of the up-ramp or down-ramp of the frequency modulation, Δf BW  is the total frequency excursion during the coherent measurement interval T P  which is equal to |f 2 −f 1 | in  FIG. 8C , and f U  and f D  are the beat frequencies in the down-converted signal corresponding to measurements during the frequency up-ramp and frequency down-ramp periods Tp respectively. 
 
         [0088]     The Doppler frequency shift of the frequency peaks measured across the down-converted signal spectrum is used to calculate object relative velocity. As an example, not meant as a limitation, let the sensor arrangement of  FIG. 4A  utilize a frequency modulation according to  FIG. 8C . Let the down-converted signal be sampled and measured during each coherent measurement interval T P , which also corresponds in this example to the frequency up-ramp and down-ramp periods. Under these conditions, object relative velocity can be calculated by the following equation:  
             V   =       c   ·     (       f   D     -     f   U       )         4   ·     f   0                 (   4   )             
 
 where V is the calculated object relative velocity defined as positive for an approaching target, c is the speed of light in a vacuum, f U  and f D  are the beat frequencies in the down-converted signal corresponding to measurements during the frequency up-ramp and frequency down-ramp modulation intervals T P  respectively, and f 0  is the average frequency of the transmitted signal during a coherent measurement period T P . 
 
         [0089]      FIG. 8D  illustrates a stepped frequency modulation waveform for use in the transmit signal generator  650 , signal generator  405  or TX &amp; LO signal generator  407  according to aspects of the present invention. This waveform shows a linearly stepped frequency pattern with a frequency increasing step sequence period and decreasing step sequence period each equal to Tp. This waveform shown is an example of linearly stepped frequency modulation and is not meant as a restriction. A typical value of Δf S  can be within, but is not limited to, the range of 100 KHz-100 MHz. A typical value of T S  can be within, but is not limited to, the range of 500 nanoseconds (ns)-20 microseconds (μs). The waveform can also comprise, but is not limited to, a repeating pattern of linearly increasing frequency steps, a repeating pattern of linearly decreasing frequency steps, or alternating periods of linearly increasing and decreasing frequency step patterns. Also, periods where the stepped frequency modulation pattern is stopped may be inserted into the abovementioned patterns. In addition, the value of T S  may be varied or dithered, or the linearity of the frequency steps with respect to time may be varied by one skilled in the art without departing from the spirit of the present invention. Furthermore, in order to achieve adequate range resolution for some applications, the total frequency modulation bandwidth, defined as |f 2 −f 1 | in  FIG. 8D  can be wideband (WB) or ultra-wideband (UWB).  
         [0090]     Using the frequency modulation waveform described in  FIG. 8D , object information may be calculated from the down-converted signals of the architectures shown in FIGS.  1 A-C, FIGS.  4 A-B and FIGS.  6 A-B in the following way. Peaks in the down-converted signal spectrum represent object returns. The frequency of the peaks is proportional to object range and is used to calculate object range. As an example, not meant as a limitation, let the sensor arrangement of  FIG. 4A  utilize a linearly increasing frequency step sequence and linearly decreasing frequency step sequence as shown in  FIG. 8D . Let the down-converted signal be sampled and measured during each coherent measurement interval T P , which for this example also corresponds to the frequency increasing step sequence period and decreasing step sequence period. Under these conditions, object range can be calculated by the following equation:  
             R   =         c   ·     T   S           4   ·   Δ     ⁢           ⁢     f   S         ·     (       f   U     +     f   D       )               (   5   )             
 
 where R is the calculated object range, c is the speed of light in a vacuum, T S  is dwell time of each frequency step, Δf S  is the difference between adjacent frequency step values in the linear step sequence, and f U  and f D  are the beat frequencies in the down-converted signal corresponding to measurements during the frequency increasing sequence and frequency decreasing sequence periods T P , respectively. 
 
         [0091]     The Doppler frequency shift of the frequency peaks measured across the down-converted signal spectrum is used to calculate object relative velocity. As an example, not meant as a limitation, let the sensor arrangement of  FIG. 4A  utilize a linearly increasing frequency step sequence and linearly decreasing frequency step sequence as shown in  FIG. 8D . Let the down-converted signal be sampled once per frequency step in each sequence, and measured during each coherent measurement interval T P , which for this example also corresponds to the frequency increasing step sequence period and decreasing step sequence period. Under these conditions, object relative velocity can be calculated by the following equation:  
             V   =       c     2   ·     (       f   1     +     f   2       )         ·     (       f   D     -     f   U       )               (   6   )             
 
 where V is the calculated object relative velocity defined as positive for an approaching target, c is the speed of light in a vacuum, f 1  and f 2  are the minimum and maximum frequency steps in the linear sequence during a coherent measurement period T P , and f U  and f D  are the beat frequencies in the digitized down-converted signal corresponding to the measurements during the frequency up-step sequence and down-step sequence periods T P , respectively. 
 
         [0092]     Object velocity information can be utilized in a variety of applications according to aspects of the present invention. One application, not meant as a limitation, is to utilize the velocity information of an object in conjunction with its positional information to determine the threat potential for purposes such as, but not limited to, deployment of countermeasures. Another application, not meant as a limitation, is to determine if there is object motion as part of a security system.  
         [0093]     An alternate way to utilize the frequency-modulated data is with three-dimensional image reconstruction techniques well known in the art. According to these techniques, the data sampled at different frequencies is utilized to reconstruct a three-dimensional rendered image using an algorithm such as, but not limited to, a backward-wave reconstruction technique.  
         [0094]     Another way to utilize the frequency-modulated data is with two-dimensional image reconstruction techniques well known in the art for each frequency step in the sequence, then average or combine the two-dimensional rendered images to improve the image characteristics such as, but not limited to, reduction of speckle or noise in the image.  
         [0095]      FIG. 9A  illustrates an example of timing of thinned-array antenna selection for use with a fixed transmission frequency according to aspects of the present invention. According to this example, unique combinations of transmit and receive antennas in the thinned-array architecture are each selected for a period of time designated by T DW , during which the down-converted signal is sampled and stored. In this example, not meant as a limitation, the transmit array consists of m by n elements, and the receive array consists of k by p elements, where m and n are non-zero integers whose sum is greater than or equal to 3, and k and p are non-zero integers whose sum is greater than or equal to 3. A typical value of T DW  can be within, but is not limited to, the range of 100 nanoseconds (ns)100 microseconds (μs). After all unique combinations of transmit and receive elements are sequenced through, the sequence is repeated for the duration of the coherent processing time period T P . The stored digital samples of the down-converted signals during this period Tp are grouped separately for each unique combination of transmit and receive antennas to create a sequence of time-ordered samples of the down-converted signals for each synthesized array element spatial position, and will be utilized for image processing. Alternately, a sequence of samples can be taken for each unique antenna combination period of time T DW  before switching to the next unique antenna combination without departing from the present invention.  
         [0096]      FIG. 9B  illustrates another example of timing of thinned-array antenna selection for use with a linearly frequency-modulated waveform according to aspects of the present invention. According to this example, unique combinations of transmit and receive antennas in the thinned array architecture are each selected for a period of time denoted T DW , during which the down-converted signal is sampled and stored. In this example, not meant as a limitation, the transmit array consists of m by n elements, and the receive array consists of k by p elements, where m and n are non-zero integers whose sum is greater than or equal to 3, and k and p are non-zero integers whose sum is greater than or equal to 3. A typical value of T DW  can be within, but is not limited to, the range of 100 nanoseconds (ns)-100 microseconds (is). After all unique combinations of transmit and receive elements are sequenced through, the sequence is repeated for the duration of the coherent processing time period T P  of the linearly frequency-modulated waveform. The stored digital samples of the down-converted signals during this period T P  are grouped separately for each unique combination of transmit and receive antennas to create a sequence of time ordered samples of the down-converted signals for each synthesized array element spatial position, and will be utilized for image processing. Alternately, the entire linear frequency modulation can be performed and a sequence of samples can be taken for each unique antenna combination period of time T DW  and the linear frequency modulation repeated for the next unique antenna combination without departing from the present invention.  
         [0097]      FIG. 9C  illustrates a further example of timing of thinned-array antenna selection for use with a linearly frequency stepped modulation waveform according to aspects of the present invention. According to this example, unique combinations of transmit and receive antennas in the thinned array architecture are each selected for a period of time denoted T DW , during which the down-converted signal is sampled and stored. In this example, not meant as a limitation, the transmit array consists of m by n elements, and the receive array consists of k by p elements, where m and n are non-zero integers whose sum is greater than or equal to 3, and k and p are non-zero integers whose sum is greater than or equal to 3. A typical value of T DW  can be within, but is not limited to, the range of 100 nanoseconds (ns)-100 microseconds (μs). After all unique combinations of transmit and receive elements are sequenced through, the sequence is repeated for the duration of the coherent processing time period T P  of the stepped frequency modulation waveform. The stored digital samples of the down-converted signals during this period T P  are grouped separately for each unique combination of transmit and receive antennas to create a sequence of time ordered samples of the down-converted signals for each synthesized array element spatial position, and will be utilized for image processing.  
         [0098]      FIG. 9D  illustrates a yet further example of timing of antenna selection for use with a linearly frequency stepped modulation waveform, compatible with image processing methods according to aspects of the present invention. This example is similar to that illustrated in  FIG. 9C , with the exception that the entire set of unique combinations of transmit and receive antennas in the thinned array architecture is sequentially selected and corresponding down-converted signals sampled during each step of the frequency stepped waveform.  
         [0099]      FIG. 9E  illustrates another example of timing of antenna selection for use with a linearly frequency stepped modulation waveform, compatible with image processing methods according to aspects of the present invention. This example is similar to that illustrated in  FIG. 9C , with the exception that the entire stepped-frequency waveform is repeated for each time period T DW  for each unique combination of transmit and receive antennas in the thinned array architecture.  
         [0100]      FIG. 9F  illustrates an example of a down-converted object return signal and A/D sample timing consistent with the stepped frequency modulation waveform and receiver antenna sequencing method described in  FIG. 9C . The A/D sample values of the down-converted signal are illustrated by the black dots superimposed on the signal, and are labeled Aj m,n,p,k , where j is an integer representing the sample number for each of the unique transmit and receive antenna combinations, m and n represent the transmit antenna element index m,n, and p and k represent the receive antenna element index p,k. As can be seen, each successive A/D sample is delayed in time with respect to the preceding A/D sample by a time equal to T DW , and occurs at a different phase on the down-converted object return signal. For image processing methods that utilize complex signal phase, it is advantageous to utilize digitized down-converted signals which have the difference in A/D sample timing between them compensated. The difference in sample timing can be compensated for in the complex frequency domain as a frequency-dependent phase shift. As an example, let each digitized sample sequence Ai m,n,p,k  of the down-converted signals during the period T P  be grouped separately for each corresponding unique transmit and receive antenna combination and ordered in time. Let each separate N-sample sequence be processed separately by an N-point complex FFT. The difference in sample timing between each antenna combination&#39;s FFT sequence can be compensated by applying the phase shift in the following equation to the complex frequency points in the FFT sequence: 
 ΔΨ j =2 π·f   j   ·ΔT   k   (7)  
 where ΔΨ j  is the complex phase shift to be applied the jth complex FFT point, f j  is the frequency of the jth position in the FFT sequence, j is an integer between 1 and N−1 for an N-point FFT sequence, and ΔT k  is the difference in time between the A/D samples in the N-sample sequence. 
 
         [0101]     According to one aspect of the present invention, the digital beam-forming (DBF) method is presented as one method of image processing. The digital beam-forming method is adapted for use with the architectures illustrated in FIGS.  1 A-C, FIGS.  4 A-B and FIGS.  6 A-B utilizing the digitized fast Fourier transformed (FFT) phase-corrected sequences for each unique combination of transmit and receive antennas in the thinned array architecture, which represent the spatial positions in the synthesized array. Once an FFT sequence is obtained for each element in this synthesized array, a multitude of array gain patterns can be generated from this set of data, and target range can be determined from the Fourier transform profiles calculated for each. One method of digital beam-forming signal processing is to generate array gain beam-patterns through combining of digitally phase shifted or digitally phase shifted and amplitude scaled complex FFT data from each synthesized array spatial position. One method of imaging an object is through scanning of these generated beam patterns across the field of view for each range bin, creating a three-dimensional rendering of the object or objects in the field of view.  
         [0102]     According to another aspect of the present invention, a super-resolution processing method is presented as another method of image processing. The super-resolution processing method is adapted for use with the architectures illustrated in FIGS.  1 A-C, FIGS.  4 A-B and FIGS.  6 A-B utilizing the digitized fast Fourier transformed (FFT) phase-corrected sequences for each of the synthesized antenna positions. In this method, a super-resolution algorithm is used to process the phase of the complex sampled signals. As an example, for a synthesized line-array of k antenna elements, the relation of the phase difference between antenna elements and angular direction of object returns can be expressed by the following equation: 
 
θ j =arcsin [ΔΨ j,b,g ·λ(2 ·π˜D   b,g )]  (8) 
 
 where θ j  is the direction from boresight in the axis of the array elements of the j th  object return, ΔΨ j,b,g  is the phase difference corresponding to the j th  object return between synthesized antenna spatial positions b and g, D b,g  is the distance separating synthesized receive antenna positions b and g in the axis in which target direction θ is to be determined, λ is the average wavelength of the transmitted waveform during a coherent measurement interval, k is an integer greater than or equal to 3, b is an integer greater than 1 and less than or equal to k+1, and g is an integer greater than 0 and less than or equal to k. Since phase differences between receive antenna positions are preserved after down-conversion, the phase differences between the down-converted difference signals corresponding to the synthesized receive antenna positions can be used for ΔΨ. The set of phase measurements between a plurality of synthesized antenna spatial positions can be used as inputs to a super-resolution algorithm, which outputs the maximum likelihood of object return angular positions based upon the set of input data. Furthermore, a super-resolution algorithm has the ability to provide angular resolution of object returns within the field of view. One super-resolution algorithm well known in the art is the multiple signal classification algorithm (MUSIC). Another super-resolution algorithm well known in the art is the estimation of signal parameters via rotational invariance techniques (ESPRIT). 
 
         [0103]     Although the preceding examples have illustrated one-dimensional and two-dimensional antenna array arrangements, the concepts and methods described can be extended to multi-dimensional arrays such as, but not limited to, multiple one-dimensional arrays arranged in multiple axes, orthogonal line-arrays, conformal arrays or three-dimensional arrays by one skilled in the art without departing from the spirit of the present invention. Also, although the preceding examples illustrate the use of switching elements to sequentially switch between antenna elements in an array to minimize hardware and cost, multiple parallel receive down-conversion channels can be utilized, as well as combinations of parallel and sequential operation as part of the present invention.  
         [0104]     Additionally, according to aspects of the present invention, a method can be utilized whereby a coarse, lower-resolution imaging mode is used to determine the location of an object rapidly, and a higher-resolution imaging mode is utilized to analyze the object. One way this can be achieved is to utilize a lower number of antenna array elements, or a sub-array of elements, for the lower-resolution imaging to determine the location of objects, and to utilize a higher number of array elements for the higher-resolution imaging where objects are determined to be located. One benefit of such a method can be to reduce the time and processing required to scan an area or volume of space.  
         [0105]     Additionally, according to aspects of the present invention, a method can be utilized whereby two or more sensors are utilized to image a common area or volume, and the sensors are synchronized such that only one sensor transmits at a time. Utilizing this method, the images from each sensor can be integrated into a common multi-dimensional view of the common area or volume.  
         [0106]     Furthermore, according to aspects of the present invention, a method is presented whereby the imaging sensor can be utilized to determine if a further action by another sensor or system is deployed. One example, not meant as a limitation, utilizes the imaging sensor to determine the location where a second type of sensor such as, but not limited to, an optical imager or camera should focus. One way the sensor can be utilized is for, but not limited to, detection of movement of one or more objects within the field of view. Another example, not meant as a limitation, utilizes the imaging sensor to determine if an object is a threat whereby a countermeasures system can be deployed.  
         [0107]     The preceding concepts, methods, and architectural elements described are meant as illustrative examples of aspects of the present invention, not as a limitation. Different combinations of these concepts, methods, and architectural elements than that described in the preceding figures can be utilized by one of ordinary skill in the art without departing from the spirit of the present invention.  
         [0108]     While certain exemplary embodiments have been described and shown in the accompanying drawings, it is to be understood that such embodiments are merely illustrative of and not restrictive on the broad invention, and that this invention not be limited to the specific constructions and arrangements shown and described, since various other modifications may occur to those ordinarily skilled in the art.

Technology Category: h