Patent Document

This application claims the benefit of Taiwan Patent Application No. 92130231 filed on Oct. 30, 2003, and No. 93113700 filed on May 14, 2004, which are hereby incorporated by reference for all purposes as if fully set forth herein. 
   BACKGROUND OF THE INVENTION 
   1. Field of Invention 
   The invention relates to a high voltage charge circuit, and particularly to a high voltage charge circuit with the ability of rapid charging. 
   2. Related Art 
   When taking a picture with a camera, a user usually uses a flash to provide sufficient light to the external environment. However, the working voltage for a flash, e.g. 300V, is much higher than the DC voltage provided by the camera, e.g. 5V. To solve this problem, a camera is set up with a high voltage charge circuit. The DC voltage (low voltage) is raised to a high voltage using a transformer having a high winding ratio charging a high voltage capacitor. When the high voltage capacitor is charged to match the working voltage of the flash, the high voltage capacitor is used as a source to provide the desired working voltage. 
   The prior high voltage charge circuit  10  is also referred to as a ring chock converter (RCC). As shown in  FIG. 1 , the RCC  10  comprises a DC source  12 , e.g. 5V, resistors  14 ,  16  and  30 , a power transistor  18 , a transformer  20 , a diode  22 , a high voltage capacitor  24 , e.g., 300V, a Zener diode  26  with a break down voltage of 300V, a capacitor  28  and a standby control circuit  32 . The transformer  20  has primary side windings N 1 , secondary side windings N 2  and auxiliary side windings N 3 , where the primary side windings and the auxiliary side windings N 3  may induct with the secondary side windings N 2 . The primary side windings N 1  have opposite polarity with the secondary side windings N 2  and the secondary side windings N 2  have a winding number N times the winding number of the primary side windings N 1 , e.g., 60 times. 
   When the DC source  12  provides a current to the transformer  20 , the resistor  14  and the primary side windings N 1  are turned on and the power transistor  18  is operated in a saturation region. Next, the auxiliary windings N 3  and resistor  14  are turned on. At this time, the current flowing through the primary windings N 1  is a magnetic current whose energy is stored in the transformer and does not charge the high voltage capacitor  24 . 
   When the current flowing through the resistor  14  gradually increases, the power transistor  18  is operated from the saturation region to the active region to decrease the current flowing through the primary side windings and invert the polarities of the primary and secondary windings N 1  and N 3 . At that time, the power transistor  18  is cut off and the secondary windings N 1  and the diode  22  turn on. After the secondary side windings N 2  transfer the energy stored in the transformer  20  to the high voltage capacitor  24 , the primary side windings N 1  go back to their initial state. Then the loop of the resistor  14  becomes conductive again and the entire process repeats. 
   When the high voltage  24  reaches a predetermined voltage level, e.g. 300V, it enables the Zener diode  26  to break down and leads to a short circuit. At that time, the standby control circuit  32  is triggered to stop the operation of the transformer  20 , and the high voltage capacitor  24  is no longer charged. 
     FIG. 2  is a diagram depicting the relation between the charging current and time. It can be seen from the drawing that when the charging current becomes zero the transformer  20  begins to work, providing a charging current to charge the high voltage capacitor  24 . 
   From the above, it may be known that the prior high voltage charging circuit  10  has the following disadvantages: 
   1. In the prior art the power transistor  18  is a bi-polar junction transistor (BJT) that when turned on requires an additional driven base current. Since the power transistor has a saturated voltage VCE of about 300 m and consumes a great deal of power, the charging effect may not be satisfactory. 
   2. Since the transformer  20  requires the winding N 3  and the power transistor  18  generally has a switch frequency of 10 kHz, the transformer is not easily miniaturized. 
   3. Since the prior high voltage charging circuit is not operated in a continuous conduction mode, it may not be efficient enough. 
   SUMMARY OF THE INVENTION 
   In view of the foregoing problem, an object of the invention is to provide a high voltage charging circuit with rapid charging ability. 
   In view of the foregoing problems, another object of the invention is to provide a high voltage charging circuit to reduce the volume of the transformer. 
   To achieve the above objects, the invention discloses a high voltage circuit that charges a high voltage capacitor and comprises a DC source outputting a large current with a low logic level and outputting a small current by performing energy transformation; a diode with a small current and a high logic level, outputting a current to the high voltage capacitor to charge the high voltage capacitor; a power transistor enabling a transformer with a large current and a low logic current, performing energy transformation while not outputting a small current with a high logic level when the power transistor is turned on, disenabling the transformer with a large current and a low logic level, performing energy transformation and outputting a small current with a high logic level when the power transistor is not turned on; a turn-on control circuit controlling the turn-on time of the power transistor; and a turn-off control circuit controlling the turn-off time of the power transistor, wherein the turn-on control circuit and the turn-off control circuit control the turn-on time or turn-off time by controlling a positive output signal of the positive output of a flip-flop. 
   According to the principle of the invention, since an extreme value of the primary side current of the transformer may setup the turn-on time of the power transistor to be shorter, the invention may use a miniature transformer that will not become saturated, achieving the object of reducing the volume of the high voltage charging circuit. 
   The detailed description of the features and advantages of the invention will be given in the following, which may enable a person skilled in the art to realize and implement the invention. 
   It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are intended to provide further explanation of the invention as claimed 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this specification, illustrate embodiments of the invention and together with the description serve to explain the principles of the invention. In the drawings: 
       FIG. 1  is a block diagram of a high voltage charging circuit in the prior art; 
       FIG. 2  depicts the relation between charging and the corresponding time in a high voltage charging circuit in the prior art; 
       FIG. 3  is a block diagram of a high voltage charging circuit according to the invention; 
       FIG. 4  is a diagram illustrating the relation of the voltage level of a primary side current, a secondary side current, a charging current and a high voltage capacitor under a continuous charging mode; 
       FIG. 5  is a diagram illustrating the relation of the voltage level of a primary side current, a secondary side current, a charging current and a high voltage capacitor under a boundary charging mode; 
       FIG. 6  is a block diagram of a second embodiment according to the invention; 
       FIG. 7  is a diagram illustrating the relation of the voltage level of a primary side current, a secondary side current, a charging current and a high voltage capacitor under a boundary charging mode according to the second embodiment of the invention; 
       FIG. 8  is a block diagram of a third embodiment of the invention; and 
       FIG. 9  is a diagram illustrating the relation of the voltage level of a primary side current, a secondary side current, a charging current and a high voltage capacitor under a boundary charging mode according to the third embodiment of the invention. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   The features and implementation of the preferred embodiment of the invention will be described in detail with the accompanying drawings. 
     FIG. 3  illustrates a first embodiment of a high voltage charging circuit according to the invention. Referring to  FIG. 3 , the circuit  40  includes three units, which are a turn-off control circuit  50 , turn-on control circuit  60  and a standby detection circuit  70 . The turn-on control circuit  60  is used to enable a positive output signal Q of the positive output of a flip-flop  46  to output a high logic level signal to control the turn-on time of a power transistor  44  by controlling an output with a low logic level to a reset terminal of the flip-flop  46 . The turn-off control circuit  50  enables the positive output signal Q of the positive output of the flip-flop  46  to have a low logic level to control the turn-off time of the power transistor by outputting a low logic level signal to the set terminal of the flip-flop. The standby detection circuit  70  is used to detect the voltage level of the charging capacitor  28 . If the charging capacitor is charged to a predetermined voltage level, e.g., 300V, the standby detection circuit  70  outputs a detection signal to enable an operation of the standby control circuit  32 , which stops the charging capacitor of the invention. As such, the high voltage charging circuit  40  does not charge the high voltage charging capacitor  24  any more. In the above, the primary side windings N 1  of the transformer  42  have an inductance of Lp, and a secondary side windings N 2  have an inductance of LS. The secondary side windings N 2  have N times windings as compared to the primary side windings N 1 . 
   When the high voltage charging circuit  40  is initialized, the secondary side current is zero, the voltage across a resistor  52  is zero and the comparator  54  outputs a high logic level signal. At this time, the flip-flop  46  has a high logic level at its set terminal, enabling the positive output signal Q of the positive output of the flip-flop  46  to have a high logic level, causing the power transistor  44  to turn on and enter into a saturation state. Because of the current mirror  62 , a current begins to charge an internal capacitor  66  of the turn-on control circuit  60  from a zero voltage and the current is as large as the current flowing through a resistor  64 . When the voltage across the capacitor  66  is less than V 3 , the power transistor  44  continues to be turned on. When the capacitor  66  is charged to a voltage greater than V 3 , the comparator  69  outputs a high logic level signal to the reset terminal of the flip-flop  46 , enabling the positive signal Q of the positive output of the flip-flop  46  to be a low logic level signal of flip-flop  46 . The low logic level output signal is driven by a driver  48  and then turns off the power transistor  44 , i.e. the power transistor  44  enters into a cut-off state. When the power transistor  44  is turned on by the turn-on control circuit  60 , this is referred to as the on time. When the power transistor  44  is in the on state, the DC source  12  provides a primary side current Ip to the transformer  42 . Since the polarity of the secondary side of the transformer may not turn on the diode, the energy generated by the primary side current Ip is stored in the transformer  42  in the form of magnetic energy. The primary side current IP has its maximum IP,max, presented in the following equation: 
                   I     P   ,   max       =       (       (       V     i   ⁢           ⁢   n       -     V   ds       )     /     L   P       )     ×     t   ON               (   1   )               
wherein Vin is the input voltage provided by the DC source  12 , Vds is the voltage across the power transistor  44 . When the power transistor  44  turns on, Vds is considerably low and thus neglected, so equation (2) may be simplified as:
 
                   I     P   ,   max       =       (       V     i   ⁢           ⁢   n       /     L   P       )     ×     t   ON               (   2   )               
wherein Vin/LP is the slope of the increase of the primary side current IP and tON is the turn-on time of the power transistor  44 , which is identical to the turn-on time of the primary side windings N 1  and is also called a turn-on time. The tON may be determined by the resistance of the resistor  64  and the capacitance of the capacitor  66 . Since the charging current for the capacitor  66  is a ratio of the voltage V 1  to the resistance of the resistor  64 , simply adjusting the resistance of the resistor  64  may change the value of tON and consequently change the maximum IP,max of the primary side current IP. Also, a change of the charging current may lead to a change of tON. When a user selects the resistance of the resistor  64 , tON is fixed. It may be seen from equation (2) that the less Vin is, the less the maximum IP,max of the primary side current IP is. When Vin falls, the charging current also decreases, so the high voltage charging circuit  40  according to the invention may provide a charging mode with a varied current, which may lengthen the lifetime of the DC source  12  when Vin is low. In addition, it may be seen through equation (2) that a user may obtain the same maximum IP,max of the primary side current IP by use of a lesser Lp. In this case, tON may be lessened, which prevents the smaller transformer  14  from being saturated. Hence, the object of reducing the volume of the high voltage charging circuit  40  may be achieved.
 
   When the flip-flop  46  outputs a low logic level signal at its positive output terminal, the flip-flop  46  simultaneously outputs a high logic level signal −Q at its negative output terminal. The high logic level output signal turns the transistor  68  on, and the energy stored in the capacitor  66  is removed. 
   In the above, the capacitor  41  is a regulated capacitor, and the power transistor  44  is preferably an NMOS transistor with its gate connected to the driver  48  to maintain its operation. The preferred power transistor has the advantages of fast response speed and reduced turn-on resistance (RDS,ON). Certainly, the power transistor  44  may also be a PMOS transistor and its gate has to be connected to a negative driver. 
   The counter  67  adds 1 to its value every 1 μs. When the value of the counter  67  exceeds a threshold, e.g. 10, the counter  67  outputs a high logic level to the reset terminal of the flip-flop  46  and the value of the counter  46  is reset to 0. Therefore, the turn-on time controlled by the turn-on control circuit  60  may be controlled by the counter and thus the primary side current IP may be limited as well. In this case, the primary side current IP may be prevented from continuous increase due to the open state of the resistor  64 . 
   During the turn-on time, the polarity at the secondary side of the transformer  42  may not turn on the diode  22 . At this time, the secondary side current Is is zero, and the voltage across the resistor  52  is zero. When the voltage across the capacitor  66  is greater than the voltage V 3 , the flip-flop  46  closes the power transistor  44 , i.e., the transistor  44  enters into a cut-off state. When the power transistor  44  is cut off, the diode  22  is turned on because the magnetic energy has to be continuous, enabling the transformer  42  to charge the high voltage capacitor  24  with its previously stored magnetic energy. At this time, the transformer  42  has current Is at its secondary side and the current Is reduces. The current Is decreases with the rate of the ratio between Vout, the voltage level of the high voltage capacitor, Ls, and the inductance of the secondary side of the transformer  42 . Since Vout increases at a slow rate in the whole charging process, the secondary side current IS has a varied decrease rate. When the voltage across the resistor  52  is greater than the voltage V 2 , the comparator  54  outputs a low logic signal to a set terminal of the flip-flop  46  and the power transistor  44  continues to be in a cut-off state. The turn-off control circuit  50  continues to cut off the power transistor  44 . When the voltage across the resistor  52  is less than V 2 , the comparator  54  outputs a high logic level signal to the set terminal of the flip-flop  46 , which turns on the power transistor  44  after the operation of the driver  48 . When the turn-off control circuit  50  turns off the power transistor  44 , this is termed OFF time. 
   The maximum IS,max of the secondary side current IS may be presented in the following equation: 
   
     
       
         
           
             
               
                 
                   I 
                   
                     S 
                     , 
                     max 
                   
                 
                 = 
                 
                   
                     
                       N 
                       1 
                     
                     
                       N 
                       2 
                     
                   
                   × 
                   
                     I 
                     
                       P 
                       , 
                       max 
                     
                   
                 
               
             
             
               
                 ( 
                 3 
                 ) 
               
             
           
         
       
     
   
   When the secondary side current IS decreases to the ratio between the voltage V 2  and the resistance of the resistor  52  (the current of the secondary side current IS is defined as a minimum Imin/N), the comparator  54  outputs a high logic level signal to the set terminal of the flip-flop, enabling the power transistor  44  to enter into a saturation state. The primary side windings N 1  then begin to conduct and the primary side current IP begins to flow from the minimum current Imin. When the above steps are repeated, the transformer  42  may continuously charge the high voltage capacitor  24  until the high voltage capacitor  24  reaches a voltage level of 300V. 
   Now the standby detection circuit  70  is described. When the high voltage capacitor  24  reaches a voltage level of 300 V, the Zener diode  26  breaks down and charges the capacitor  28 . Until the capacitor  28  has a greater voltage difference than the voltage V 3 , the comparator  72  outputs a detection signal to the standby control circuit  32 , enabling the high voltage charging circuit  40  to stop. 
   The minimum current Imin may be set by a user. For example, when the minimum current Imin is greater than zero, the high voltage charging circuit  40  in the invention continues to charge the high voltage capacitor  24 , which is referred to as in a continuous charging mode. The variations of the primary side current Ip, the secondary side current Is and Vout may be readily known through  FIG. 4  during the period the power transistor  44  turns on and off. Since the high voltage charging circuit  40  charges the high voltage capacitor  24  in a continuous charging mode, the charging efficacy is better than the continuous/non-continuous charging efficacy used in the prior art. In addition, the minimum current Imin may be set as zero. The high voltage charging circuit  40  then charges the high voltage capacitor  24  in a continuous/non-continuous mode, also termed the boundary charging mode. Referring  FIG. 5 , the variations of the primary side current Ip, the secondary side current Is and Vout when the power transistor  44  is on and off may be clearly known. 
   In the embodiment illustrated in  FIG. 3 , when V 1 =Vin, the higher the input voltage Vin, the faster the charging speed of the capacitor  66  and tON, wherein the value of tON may be determined by the resistor  64  and the capacitor  66  since the charging current for the capacitor  66  is equal to the value of the ratio of the input voltage Vin and the resistance of the resistor  64 . On the other hand, the less input voltage Vin, the slower the charging of the capacitor  66  and the greater tON. That is, the value of Vin×Ton may be kept constant. 
   Therefore, a user may change the value of Vin×tON by adjusting the resistance of the resistor  64 , and the value of the maximum current IP,max of the primary side current IP may also be changed. When the resistance of the resistor  64  is fixed, the value of Vin*tON is also fixed, and the maximum current IP,max of the primary side current IP may not change with Vin, so the invention provides a constant-current charging mode. 
     FIG. 6  illustrates a second embodiment of the high voltage charging circuit  40  according to the invention. 
   Referring to  FIG. 6 , the high voltage charging circuit  40  also comprises the turn-off control circuit  50  and the turn-on control circuit  60 . The turn-on control circuit  60  is used to receive the input voltage Vin and is connected to the negative output (−Q) and the reset terminal (R) of the flip-flop  46 . The turn-off control circuit  50  is connected to the set terminal (S) of the flip-flop  46 . In this embodiment, the turn-off control circuit  50  operates according to the voltage Vin provided by the DC source  12  and the voltage difference of the power transistor  44  Vds. The transformer  42  is connected at one terminal of its secondary side to the ground. 
   In the second embodiment, the turn-on control circuit  60  comprises a first current source  63 , a capacitor  66 , a comparator  69  and a transistor  68 . The first current source  63  and the capacitor  66  are coupled together, providing a first charging current to the capacitor  66  and charging the capacitor  66 , wherein the first charging current Ip=(Vin×Ton)/Lp. When the voltage across the capacitor  66  is the first reference voltage  3 , the power transistor  44  is always turned on. When the capacitor  66  is charged to a voltage level greater than the first reference voltage V 3 , the comparator  69  outputs a reset signal to the reset terminal (R) of the flip-flop  46  to control a positive output signal at the positive output +Q of the flip-flop  46 , and cuts off the power transistor  44  through the driver  46 . At this time, the negative signal of the negative output −Q of the flip-flop  46  turns on the transistor  68 , enabling the transistor  68  to provide a discharging path for the capacitor  66 . When the capacitor  66  is charged, the power transistor  44  is in an on state. 
   In this embodiment, the value of the current provided by the first current source  63  is related with the voltage Vin provided by the DC source  64  and the resistor  65 , and is a function of the input voltage Vin or the resistor  65 . The outputted current charges the capacitor  66 , and the turn-on time of the power transistor  44  is determined by the charging period of the capacitor  66 . Referring  FIG. 7 , the turn-on period of the power transistor  44  is kept constant at all times and may be determined by changing the first reference voltage V 3  or the resistance of the resistor  64 . 
   In the second embodiment, during a turn-on period, the polarity of the transformer  42  at its secondary side results in the un-conductivity of the diode  28 . The secondary side current Is at this time is zero. When the capacitor  66  has a voltage difference between its two terminals greater than the voltage V 3 , the flip-flop  46  turns off the power transistor  44 , i.e., the transistor  44  enters into a cut-off state. After the power transistor  44  is cut off, the diode  28  is turned on, enabling the previously stored magnetic energy in the transformer  42  to charge the high voltage capacitor  24 . At this time, the transformer  42  has a current is flowing through in its secondary side, and the current is reduces. When the voltage Vds across the power transistor  44  is greater than the sum of Vin and the voltage V 2 , the comparator  54  outputs a low level signal to the set terminal of the flip-flop  46  and the power transistor  44  is kept in a cut off state. The turn-off control circuit  50  maintains the power transistor  44  in the cut-off state. Until the power transistor  44  has the voltage drop Vds smaller than the sum of Vin and the voltage V 2 , the comparator  54  outputs a high level signal to the set terminal of the flip-flop  46 , and the power transistor  44  turns on after the operation of the driver  48 . In other words, the power transistor is turned on by the turn-off control circuit  50  when the energy of the transformer is released to a threshold defined by the summation of Vin and V 2 . The time that the turn-off control circuit  50  turns off the power transistor  44  is termed an off time. 
   In the second embodiment, the turn-off time of the power transistor  44  is determined by the current Is flowing through the secondary side of the transformer  42 . When the current Is in the secondary side falls to zero, the voltage on the primary side of the transformer falls to 0V at a rapid speed and the voltage drop Vds of the power transistor  44  rapidly falls to Vin. Since the energy stored in the secondary side of the transformer has been totally released, the power transistor  44  again turns on when the power transistor  44  has the Vds falling to the sum of Vin and the voltage V 2 . For example, when Vds is 50 mV, the flip-flop  46  has a set terminal (S) output of 1. The time period that the secondary side current Is falls from Ipeak/N to 0 is determined by the turn-off time of the power transistor  44 . With the circuit in the second embodiment, a maximum turn-on time is set so as to avoid transformer saturation. 
   Turning to  FIG. 7 , when the power transistor  44  turns on and off in the second embodiment, the variations of the primary side current Ip, secondary side current Is and the output voltage Vout may be clearly seen. 
   The transformer  42  at its secondary side has the Is of the ramp-down slope of Vout/Ls and the turn-off time of the power transistor  44  becomes shorter and shorter. Since the turn-on time is fixed, the turn-off time reduces and the switching frequency of the power transistor  44  is frequency variable. 
   In the second embodiment, the standby detection circuit  70  further comprises a third comparator  74  comparing the voltage across the resistor  30  and the second reference voltage V 4 , and a fourth comparator  76  comparing the voltage across the resistor  30  and the third reference voltage V 5 . The second and third reference voltages V 4  and V 5  have a particular relationship. For example, the third reference voltage V 5  is 0.9 times the fourth reference voltage V 4 . When the voltage of the resistor  30 , equal to the voltage across the capacitor  28 , reaches the second reference voltage V 4 , the third comparator  74  outputs a turn-off signal to turn off the turn-off control circuit  50  and the turn-on control circuit  60 . When the voltage of the resistor  30  reaches the third reference voltage V 5 , the fourth comparator  76  outputs a signal to turn on the turn-off control circuit  50  and the turn-on control circuit  60 . Through the operation of the comparator  74  and the fourth comparator  76 , the high voltage charging circuit  40  is equipped with an automatic recharging function, i.e., when the voltage drop of the capacitor  28  falls to the third reference voltage V 5 , the high voltage charging circuit  40  automatically restarts to charge the capacitor  28 . The time of automatic recharging Trefresh=−In(V 5 /V 4 )×resistor  30 ×capacitor  28 . 
     FIG. 8  illustrates a third embodiment according to the invention. In this embodiment, the configuration and operation of the circuit is the same as in the second embodiment except for the turn-off control. 
   In the third embodiment, the turn-off control circuit  50  is similar to the turn-on control circuit  60 . The turn-off control circuit  50  comprises a second current source  53 , a leading edge blanking circuit (hereinafter LEB circuit)  55 , a comparator  54 , a capacitor  56  and a transistor  58 . In addition, the turn-off control circuit comprises a single shot circuit  80  coupled to the set terminal S of the flip-flop  46  to trigger the turn-on control circuit  60 . The LEB circuit  55  leaves blank on the rising edge of the pulse for the voltage difference of Vds and Vin. For example, by setting the LEB time to about 200 ns, the generated pulse may be neglected. The second current source  53  is coupled to the LEB circuit  55  and outputs a current as a function of the voltage difference Vds−Vin, i.e., I=f(Vds−Vin). The capacitor  56  is coupled to the second current source  53 . The transistor  58  is coupled to the positive input of the comparator  54  and to the positive output of the flip-flop  46  with the gate thereof. 
   The single shot circuit  80  is used to trigger the turn-on control circuit  60 . The first current source  63  has the linear output current I 1 =f(Vin) as described above to charge the capacitor  66 . The time during which the capacitor  68  is charged to the first reference voltage V 3  is used to determine the turn-on time of the power transistor  44 . 
   The second current source  53  has a current of I 2 =f(Vds−Vin), which is consistent with the first current source  63  and is used to charge the capacitor  56 . The time during which the capacitor  68  is charged to the first reference voltage V 3  determines the turn-off time of the power transistor  44 . 
   In the third embodiment, the first current source  63  and the second current source  53  charge the capacitor  68  so that the averaged voltage of the inductor is zero when the magnetic element is in a stable state, i.e., the relation Vin×Ton=(Vds−Vin)×Toff is satisfied. 
   Referring to  FIG. 9 , during the turn-on and turn-off time of the power transistor  44 , the variations of the primary side current Ip, secondary side current Is and the output voltage Vout in the third embodiment may be clearly known. 
   According to the principle of the invention, the high voltage charging circuit  40  has the following advantages: 
   1. Auxiliary windings N 3  are not necessary. A smaller Lp may be used to set a smaller value of the turn-on time tON so that a small-sized transformer  14  may be used. The transformer will not become saturated, thus allowing reduction in volume of the high voltage charging circuit. 
   2. The high voltage charging circuit may operate in a continuous turn-on mode and thus has a higher power conversion efficiency and a shorter charging time. 
   3. A constant current (the charging current does not vary with Vin) or a variable current (the charging current decreases as Vin falls) may be used in the charging mode, depending upon the user. 
   4. An automatic recharging function may be set by the user. 
   The invention being thus described, it will be obvious that the same may be varied in many ways. Such variations are not to be regarded as a departure from the spirit and scope of the invention, and all such modifications as would be obvious to one skilled in the art are intended to be included within the scope of the following claims.

Technology Category: h