Patent Document

BACKGROUND 
     1. Field of the Invention 
     The present invention relates to signal demodulation circuits, and in particular, to quadrature signal demodulators for use in phased array applications, including ultrasound. 
     2. Related Art 
     Ultrasound or SONAR pressure waves are generated by feeding a radio frequency (RF) transmit signal voltage to groups of piezoelectric elements in an array. These elements convert the electrical energy to acoustic energy. Such energy conversion is reciprocal in nature, since the same elements, as well as additional elements, convert reflected acoustic energy into electrical energy. Generally, this converted reflected energy is then amplified by a sensitive signal receiver. If the timing or phase of the individual elements is somehow adjusted to form a transmit and/or receive “lens”, the process is called a “phased array” and is used in phased array ultrasound, phased array sonar and phased array radar applications. When circuitry, e.g., digital signal processors (DSPs) create, transmit, focus, steer and detect with the reflected energy beams, the process is called “digital beamforming”. 
     Doppler frequency shift in ultrasound or sonar is used to determine the velocity of a target relative to the observer. In medical ultrasound applications, the target is usually a moving bolus or volume of blood. If blood flows exactly toward the transmitted beam, the receive signal is shifted up in frequency (positive Doppler shift) by an amount proportional to the product of twice the directed target velocity and the transmitted signal frequency, divided by the velocity of sound in blood. If the directed or effective target velocity is not exactly toward the transmitted signal beam, the frequency shift is multiplied by the cosine of the angle between the instantaneous target velocity vector and that of the transmitted signal beam. Similarly, Doppler shift is negative (frequency decrease) if the directed target velocity is away from the transmitted signal beam. This can be expressed as follows:
 
Δ f =(2 Vt  cos Φ)( f   TX )/ c  
 
     Δf=Doppler frequency shift 
     V t  cos Φ=directed target velocity including the angular component 
     f TX =transmit frequency 
     c=velocity of sound in blood 
     Blood is predominantly water, and the velocity of sound in water is approximately 1,580 meters per second (m/sec.) or 1.58 millimeters per microsecond (mm/usec). Assuming the operating frequency, i.e., the transmitted signal frequency, is three megahertz (3 MHz) and normal blood flow in the carotid arteries of the neck varies between 30 and 80 centimeters per second (cm/sec.) throughout the cardiac cycle, and assuming the blood flow is directed 45 degrees from the transmitted signal beam, the resultant Doppler shift, based upon the equation above, will vary between 800 and 2140 Hertz, both of which are well within the audible signal range. When the carotid cross-section is partially occluded by plaque, the blood velocity increases to maintain flow and Doppler shift increases. An unusual shift will be audible to the sonographer and visible when displayed on a color monitor. If the obstruction creates turbulence or cavitation, the effect will be even more noticeable. 
     SUMMARY 
     In accordance with the presently claimed invention, quadrature signal demodulator circuitry is provided for demodulating multiple related input signals into respective pairs of quadrature signals for selective combining to provide a composite pair of quadrature signals with a maximized signal-to-noise ratio (SNR). 
     In accordance with one embodiment of the presently claimed invention, quadrature signal demodulator circuitry includes: 
     a plurality of quadrature signal demodulator circuits each of which includes
         signal routing circuitry responsive to a respective one of a plurality of received signals by providing corresponding first and second input signals, and   signal mixing circuitry coupled to the signal routing circuitry and responsive to the first and second input signals, one or more clock signals and respective one or more phase control signals by providing respective first and second output signals which are related to the first and second input signals, have substantially mutually quadrature signal phases, and are respective ones of first and second pluralities of output signals, respectively, wherein first and second ones of the first plurality of output signals have a first mutual signal phase difference, first and second ones of the second plurality of output signals corresponding to the first and second ones of the first plurality of output signals have a second mutual signal phase difference, and the first and second mutual signal phase differences are substantially equal; and       

     output signal combining circuitry coupled to the plurality of quadrature signal demodulator circuits and responsive to the first and second pluralities of output signals by providing first and second resultant signals, respectively. 
     In accordance with another embodiment of the presently claimed invention, quadrature signal demodulator circuitry includes: 
     a plurality of quadrature signal demodulator means each of which includes
         signal router means for routing a respective one of a plurality of received signals to provide corresponding first and second input signals, and   signal mixer means for receiving the first and second input signals, one or more clock signals and respective one or more phase control signals and in response thereto providing respective first and second output signals which are related to the first and second input signals, have substantially mutually quadrature signal phases, and are respective ones of first and second pluralities of output signals, respectively, wherein first and second ones of the first plurality of output signals have a first mutual signal phase difference, first and second ones of the second plurality of output signals corresponding to the first and second ones of the first plurality of output signals have a second mutual signal phase difference, and the first and second mutual signal phase differences are substantially equal; and       

     output signal combiner means for combining the first and second pluralities of output signals to provide first and second resultant signals, respectively. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  illustrates a use of a piezoelectric array to detect and track a moving target in accordance with the presently claimed invention. 
         FIG. 2  is a functional block diagram of quadrature signal demodulator circuitry in accordance with various embodiments of the presently claimed invention. 
         FIGS. 3A-3D  are functional block diagrams of portions of the circuitry of  FIG. 2  in which signal phases can be controlled in accordance with various embodiments of the presently claimed invention. 
         FIG. 4  is a functional block diagram of a portion of the circuitry of  FIG. 2  for dividing the incoming signal to perform a quadrature signal demodulation. 
         FIGS. 5A and 5B  are functional block diagrams of current mirror circuitry used as part of the circuitry of  FIG. 2  for controlling the phases of the quadrature demodulated signals in accordance with one embodiment of the presently claimed invention. 
         FIG. 6  is a functional block diagram of signal scaling circuitry for scaling the output signals of  FIGS. 5A and 5B  in accordance with another embodiment of the presently claimed invention. 
         FIG. 7  illustrates phasor diagrams depicting signal phase control in accordance with the presently claimed invention. 
         FIG. 8  is a functional block diagram of the clock circuitry of  FIG. 2  in accordance with another embodiment of the presently claimed invention. 
         FIG. 9  is a logic diagram of an exemplary embodiment of the multi-phase frequency divider of  FIG. 8 . 
     
    
    
     DETAILED DESCRIPTION 
     The following detailed description is of example embodiments of the presently claimed invention with references to the accompanying drawings. Such description is intended to be illustrative and not limiting with respect to the scope of the present invention. Such embodiments are described in sufficient detail to enable one of ordinary skill in the art to practice the subject invention, and it will be understood that other embodiments may be practiced with some variations without departing from the spirit or scope of the subject invention. 
     Throughout the present disclosure, absent a clear indication to the contrary from the context, it will be understood that individual circuit elements as described may be singular or plural in number. For example, the terms “circuit” and “circuitry” may include either a single component or a plurality of components, which are either active and/or passive and are connected or otherwise coupled together (e.g., as one or more integrated circuit chips) to provide the described function. Additionally, the term “signal” may refer to one or more currents, one or more voltages, or a data signal. Within the drawings, like or related elements will have like or related alpha, numeric or alphanumeric designators. Further, while the present invention has been discussed in the context of implementations using discrete electronic circuitry (preferably in the form of one or more integrated circuit chips), the functions of any part of such circuitry may alternatively be implemented using one or more appropriately programmed processors, depending upon the signal frequencies or data rates to be processed. 
     Maximizing transmit signal-to-noise ratio (SNR) in phased array ultrasound or SONAR requires focusing the transmitted signal beam within an intended volume of interest. This is done by phasing or time delaying the signal to individual array elements, thereby creating an acoustical lens. During reception, the SNR becomes maximized by aligning the phase of each individual receive channel signal prior to summation of the signal amplitudes for each channel. Ideally, the noise in each received channel is uncorrelated with noise in other channels. Amplitude summation improves the SNR by the square root of the number of added channels, e.g., ideally producing a three decibel (3 dB) improvement each time the channel count is doubled. 
     Referring to  FIG. 1 , in accordance with one embodiment of the presently claimed invention, a piezoelectric array  10  is driven with one or more transmit signals  11   t  which are typically continuous wave (CW) signals. Typically, one end portion  12   t  of the array  10  is driven by the transmit signals  11   t , with each individual piezoelectric element  100  emitting a respective sound wave  13   t . Another end portion  12   r  of the array  10  a number of elements  100  are used for receiving reflected sound waves  13   r  from the target  14  moving along a velocity vector  15 . These elements provide one or more receive signals  11   r  for processing (discussed in more detail below). An exemplary array  10  includes 64 elements  100 , of which 25 elements will be transmit elements  12   t  and another 25 elements will be receive elements  12   r , with 14 unused elements  12   n  separating them to minimize direct transmit-to-receive crosstalk. 
     In CW Doppler there is no range information. If range information is required, pulsed Doppler can be used, with the elements driven by narrow transmit signal pulses. Most or all of these same elements  100  can be used to both transmit the original signal pulses as well as receive the reflected pulses, since “separation” for minimizing direct transmit-to-receive crosstalk is achieved temporally instead of spatially. The receive signals are generally range-gated and the receive signal returns through either a B-mode path or through a CW Doppler path with sample-and-hold signal processing. 
     To detect a CW Doppler frequency shift, the receive signal  11   r  is compared to a sample of the CW carrier used to create the transmit signal  11   t . One common way is to multiply the two signals together in a double balanced mixer. In accordance with well known techniques, proper signal mixing produces sum and difference signal frequencies, with the magnitudes of the original two signals being highly attenuated. The sum frequencies, all other harmonic frequencies or high frequency cross-products are removed downstream by lowpass filtering, with the low frequency difference or baseband Doppler signal being left substantially intact. To further suppress spurious frequencies, receive signals are sometimes first mixed down to a convenient intermediate frequency (IF), narrow-band filtered with a bandpass filter (BPF), and then mixed again down to baseband frequency. 
     As discussed in more detail below, a Doppler demodulator in accordance with the presently claimed invention rotates or aligns the phase of each individual receive channel signal, performs baseband demodulation by heterodyning the signal with a system clock signal, and coherently sums a selected number of properly phased baseband signals, thereby maximizing the SNR as discussed above. Based on factors, such a position of a transmit and receive element position within the array  10  and the depth of focus (i.e., the distance to the target  14 ), the system estimates the required phase shift for each channel element  100  needed to align the baseband in-phase I and quadrature Q signals with the baseband I and Q signals of the other channels prior to their summations (magnitude). If the RF signal is shifted in phase, the I and Q baseband signals will exhibit the same nominal phase shift. Baseband in-phase signal summation (magnitude) adds the signals directly, while uncorrelated noise adds as the square root of the number of added channels. Accordingly, as channels are correctly aligned and added, the SNR increases as the square root of the number of channels. 
     Referring to  FIG. 2 , an exemplary embodiment of a receive channel  100  in accordance with the presently claimed invention includes a low noise amplifier (LNA)  102 , a signal divider (magnitude)  104  and quadrature signal demodulators circuitry  106 , which includes I signal circuitry  106   i , Q signal circuitry  106   q  and clock circuitry  106   c  (discussed in more detail below). The clock circuitry  106   c  provides quadrature clock signals  101   i ,  101   q  to the I signal  106   i  and Q signal  106   q  circuitry. The resulting I signal  107   i  and Q signal  107   q  are then summed in magnitude with I signals  107   in  and Q signals  107   qn  from other elements  100  in the receiver portion  12   r  of the array  10  in respective signal summing (magnitude) circuits  108   i ,  108   q . The resulting summation signals  109   i ,  109   q  can, if desired, be scaled in magnitude by respective scaling circuits  110   i ,  110   q  in accordance with one or more magnitude scaling control signals  101   m . Internal scaling allows summation of multiple IC output signals without changing the gain of external amplifiers. The resulting scaled signals  111   i ,  111   q  are filtered by respective lowpass filters  112   i ,  112   q  to produce the baseband I signal  11   ri  and Q signal  11   rq . These signals  11   ri ,  11   rq  can be processed using the well known Hilbert transform into “left” and “right” audio signals for use in a stereo audio circuit (not shown) to be monitored by the user of the system. One ear will hear proper cardiac blood flow while the other ear will hear a reverse flow echo from a leaky or regurgitating heart valve. 
     Referring to  FIG. 3A , the estimated phase shift (discussed above) can be applied to the output signal  103   a  of the LNA  102  using controllable phase shift circuitry  200  in accordance with one or more phase shift control signals  201 . The resulting phase-shifted input signal  103   b  is then distributed for quadrature demodulation using the quadrature clock signals  101   i ,  101   q . As discussed above, the phase shift introduced via the phase shifter  200  may be similar or different among the various receive channels  100  in the receiver portion  12   r  of the array  10 , and is intended to be dynamic with the capability of changing in microseconds. 
     Referring to  FIG. 3B , the I signal  105   i  and Q signal  105   q  are mixed (e.g., multiplied with respect to frequency) with the I clock signal  101   i  and Q clock signal  101   q  in respective signal mixing circuits  206   i ,  206   q . The Q clock signal  101   q  is provided by phase shifting one of the signals  203  provided by the signal divider (magnitude)  202  via a fixed phase shift circuit  204 . In accordance with one embodiment of the presently claimed invention, the estimated phase shift is applied further downstream individually to the I signal  207   ia  and Q signal  207   qa  via respective phase shifters  208   i ,  208   q  in accordance with one or more phase shift control signals  209  (discussed in more detail below). The resulting phase-shifted signals  207   ib ,  207   qb  retain their mutual quadrature phase relationship while both being rotated in phase in accordance with the estimated phase shift. Additionally, these signals  207   ib ,  207   qb  can be scaled in magnitude using scaling circuits  210   i ,  210   q  controlled in accordance with one or more magnitude scaling control signals  211  (discussed in more detail below). 
     Referring to  FIG. 3C , in accordance with another embodiment of the presently claimed invention, the estimated phase shift is introduced via the input clock signal  101  ca (which, as discussed above, has the same frequency as the transmit signal  11   t  carrier, and is shared by the two channels I, Q since it is the source of the quadrature clock signals  101   i ,  101   q ). This phase shift is introduced via a phase shifter  212   a  controlled by one or more phase shift control signals  213 . The resulting phase-shifted clock signal  101   cb  is divided (magnitude) into two signals, one serving as the I clock signal  101   i , and the other signal  203  being further phase shifted via a fixed phase shift circuit  214  to produce the Q clock signal  101   q . In accordance with a preferred embodiment, this phase shifter  214  is a four-stage delay shift register which is clocked by a clock signal  217  having a frequency four times that of the original clock signal  101   c . This can be done with a frequency multiplier  216  (e.g., a phase-locked loop) that multiplies the frequency of the clock signal  101   c  by the factor of four. 
     While the incoming clock signal  101  ca, which is the source of the local oscillator signal, can be analog in form, thereby requiring an analog phase shifter  212   a , it is preferable to use a digital clock signal  101  ca. This allows the phase shifter  212   a  to be implemented using shift registers (discussed in more detail below). This allows variable phase shifting to be performed easily using digital shift registers, with the time delay or angular resolution depending simply upon the granularity (frequency or fineness) of the master clock  101   ca  cycles and the number of divider delay stages. Additionally, the downstream signal divider  202  can be implemented using logic gates. Accordingly, with the quadrature clock signals  101   i ,  101   q  being digital, the demodulated signals  207   ia ,  207   qa  produced by the mixers  206   i ,  206   q  ( FIG. 3B ) will also be digital, i.e., square waves, consisting of the fundamental frequency and its odd harmonics. This significantly simplifies the requirements for the downstream lowpass filters  112   i ,  112   q  ( FIG. 2 ). 
     Referring to  FIG. 3D , as discussed in more detail below, if the incoming clock signal  101   ca  is digital, it is possible for the phase shifter  212   b  to shift phase, in accordance with its one or more phase shift control signals  213 , as well as mix signals simultaneously to produce the I and Q clock signals  101   i ,  101   q  (discussed in more detail below). 
     As discussed in more detail below, in accordance with the presently claimed invention, the respective channel signals  101   r  ( FIG. 2 ) are individually rotated or aligned in one of 16 angles as determined by a multi-bit binary code provided via the appropriate control signals  209  ( FIG. 3B ),  213  ( FIGS. 3C and 3D ). Accordingly, the selected angles include 0, 22.5, 45.0, . . . , and 337.5 degrees. Each channel receives its own set of control bits. These 16 angles provide for a maximum ideal magnitude error of 1−cos(11.25 degrees), which equates to 1.92%. 
     Referring to  FIG. 4 , in accordance with an exemplary embodiment, the input LNA  102   a  provides a differential signal having positive  103   p  and negative  103   n  signal phases. These signals  103   p ,  103   n  drive the signal divider (magnitude)  104   a  which is implemented as two sets of unity gain sinking current mirror circuits  104   p ,  104   n . The positive signal phase  103   p  is mirrored as two sinking output currents  105   pi ,  105   pq . Similarly, the negative signal phase  103   n  is mirrored as two sinking output currents  105   ni ,  105   nq . Current signals  105   pi ,  105   ni  from each of these current mirrors  104   p ,  104   n  are provided to the I signal mixer  206   i . Similarly, additional current signals  105   pq ,  105   nq  from each of these current mirrors  104   p ,  104   n  are provided to the Q signal mixer  206   q . These signals  105   pi ,  105   ni ,  105   pq ,  105   nq  are mixed with the clock signals  101   i ,  101   q  to produce a differential I signal  207   i  having positive  207   ip  and negative  207  in signal phases, and a differential Q signal  207   q  having positive  207   qp  and negative  207   qn  signal phases. 
     Referring to  FIGS. 5A and 5B , in accordance with one embodiment of the presently claimed invention, the positive  207   ip ,  207   qp  and negative  207   in ,  207   qn  signal phases of the I signal  207   i  and Q signal  207   q  drive the phase shifters  208   i ,  208   q  ( FIG. 3B ). Referring to  FIG. 5A , the I signal phase shifter  208   i  is implemented with four stages  208   iun ,  208   iup ,  208   idn ,  208   idp , each of which includes a respective current source mirror circuit  220   iun ,  220   iup  or current sink mirror circuit  220   idn ,  220   idp , and a respective output switching matrix  222   iun ,  222   iup ,  222   idn ,  222   idp . These current source  220   iun ,  220   iup  and current sink  220   idn ,  220   idp  mirror circuits provide their respective output current signals  221   i  each of which is trigonometrically weighted relative to their respective input signals  207   ip ,  207  in,  221   iunf ,  221   iupf  (each of the current source mirror circuits  220   iun ,  220   iup  also provides a unity gain output current  221   iunf ,  221   iupf  mirrored from its respective input current  207   ip ,  207   in ). In accordance with well known techniques, the relative dimensions of the transistors, values of their emitter resistances, or both, are used to implement the current mirror circuits  220   iun ,  220   iup ,  220   idn ,  220   idp  are selected so as to provide the desired trigonometric weighting of the mirrored currents. 
     Accordingly, the first source current mirror circuit  220   iun  provides five current signals  221   iuna ,  221   iunb ,  221   iunc ,  221   iund ,  221   iune , each of which has a magnitude trigonometrically weighted relative to the input current  207   ip  with a respective one of the following values: −0.3827, −0.9239, −0.7071 (2 each), −0.9999. Similarly, the second current source mirror circuit  220   iup  provides five currents  221   iupa ,  221   iupb ,  221   iupc ,  221   iupd ,  221   iupe  with respective magnitudes trigonometrically weighted relative to the input signal  207  in: +0.3827, +0.9239, +0.7071 (2 each), and +0.9999. Further similarly, the first current sink mirror circuit  220   idn  provides five currents  221   idna ,  221   idnb ,  221   idnc ,  221   idnd ,  221   idne  having magnitudes trigonometrically weighted relative to its input signal  221   iunf : −0.3827, −0.9239, −0.7071 (2 each), −0.9999. Further similarly, the second current sink mirror circuit  220   idp  provides five currents  221   idpa ,  221   idpb ,  221   idpc ,  221   idpd ,  221   idpe  having magnitudes trigonometrically weighted relative to its input signal  221   iupf +0.3827, +0.9239, +0.7071 (2 each), and +0.9999. 
     In accordance with their respective control signals  209   iun ,  209   iup ,  209   idn ,  209   idp , each of the switch matrixes  222   iun ,  222   iup ,  222   idn ,  222   idp  selects each of its respective input current signals  221   iun ,  221   iup ,  221   idn ,  221   idp  as its corresponding respective output current signal  223   iun ,  223   iup ,  223   idn ,  223   idp , and directs unselected source currents to the circuit ground reference and unselected sink currents to the positive power supply node (thereby preventing base currents of the inactive, or unselected, devices from diverting base drive currents from the active, or selected, devices). 
     Referring to  FIG. 5B , the Q signal phase shifter  208   q  is implemented with four stages  208   qun ,  208   qup ,  208   qdn ,  208   qdp , each of which includes a respective current source mirror circuit  220   qun ,  220   qup  or current sink mirror circuit  220   qdn ,  220   qdp , and a respective output switching matrix  222   qun ,  222   qup ,  222   qdn ,  222   qdp . These current source  220   qun ,  220   qup  and current sink  220   qdn ,  220   qdp  mirror circuits provide their respective output current signals  221   q  each of which is trigonometrically weighted relative to their respective input signals  207   qn ,  207   qp ,  221   qunf ,  221   qupf  (each of the current source mirror circuits  220   qun ,  220   qup  also provides a unity gain output current  221   qunf ,  221   qupf  mirrored from its respective input current  207   qn ,  207   qp ). In accordance with well known techniques, the relative dimensions of the transistors, values of their emitter resistances, or both, are used to implement the current mirror circuits  220   qun ,  220   qup ,  220   qdn ,  220   qdp  are selected so as to provide the desired trigonometric weighting of the mirrored currents. 
     Accordingly, the first source current mirror circuit  220   qun  provides five current signals  221   quna ,  221   qunb ,  221   qunc ,  221   qund ,  221   qune , each of which has a magnitude trigonometrically weighted relative to the input current  207   qn  with a respective one of the following values: −0.3827, −0.9239, −0.7071 (2 each), −0.9999. Similarly, the second current source mirror circuit  220   qup  provides five currents  221   qupa ,  221   qupb ,  221   qupc ,  221   qupd ,  221   qupe  with respective magnitudes trigonometrically weighted relative to the input signal  207   qp : +0.3827, +0.9239, +0.7071 (2 each), and +0.9999. Further similarly, the first current sink mirror circuit  220   qdn  provides five currents  221   qdna ,  221   qdnb ,  221   qdnc ,  221   qdnd ,  221   qdne  having magnitudes trigonometrically weighted relative to its input signal  221   qunf : −0.3827, −0.9239, −0.7071 (2 each), −0.9999. Further similarly, the second current sink mirror circuit  220   qdp  provides five currents  221   qdpa ,  221   qdpb ,  221   qdpc ,  221   qdpd ,  221   qdpe  having magnitudes trigonometrically weighted relative to its input signal  221   qupf : +0.3827, +0.9239, +0.7071 (2 each), and +0.9999. 
     In accordance with their respective control signals  209   qun ,  209   qup ,  209   qdn ,  209   qdp , each of the switch matrixes  222   qun ,  222   qup ,  222   qdn ,  222   qdp  selects each of its respective input current signals  221   qun ,  221   qup ,  221   qdn ,  221   qdp  as its corresponding respective output current signal  223   qun ,  223   qup ,  223   qdn ,  223   qdp , and directs unselected source currents to the circuit ground reference and unselected sink currents to the positive power supply node (thereby preventing base currents of the inactive, or unselected, devices from diverting base drive currents from the active, or selected, devices). 
     The current mirrors providing a weighting of 0.9999 have a gain of unity, a weighting of 0.38271 corresponds to the nominal sine of 22.5 degrees and cosine of 67.5 degrees, a weighting of 0.7071 corresponds to the nominal sine and cosine of 45 degrees, and a weighting of 0.9239 corresponds to the nominal cosine of 22.5 degrees and sine of 67.5 degrees. The current source mirror circuits  220   iun ,  220   iup ,  220   qun ,  220   qup  provide pull-up currents mirrored from the positive power supply electrode. The current sink mirror circuits  220   idn ,  220   idp ,  220   qdn ,  220   qdp  provide pull-down currents mirrored from the negative power supply electrode, or circuit reference ground in the case of a single power supply voltage. 
     Referring to  FIG. 6 , the selected trigonometrically weighted source currents  223   iun ,  223   iup ,  223   qun ,  223   qup  and sink currents  223   idn ,  223   idp ,  223   qdn ,  223   qdp  are summed (magnitude) together in respective current summing circuitry  224   iu ,  224   qu ,  224   id ,  224   qd  (e.g., respective shared circuit electrodes via which hard-wired current summing occurs). The resulting sum currents  225   iu ,  225   qu ,  225   id ,  225   qd  are scaled by respective thermometer-weighted signal scalers  210   iu ,  210   qu ,  210   id ,  210   qd . Each input current  225   iu ,  225   qu ,  225   id ,  225   qd  is replicated by its respective current mirror circuit  226   iu ,  226   qu ,  226   id ,  226   qd  in accordance with one or more gain control signals  211   a . The resulting scaled currents  227   iu ,  227   qu ,  227   id ,  227   qd  are individually selected or diverted to circuit ground reference or the positive power supply node (as discussed above for  FIGS. 5A and 5B ) by a respective switch matrix  228   iu ,  228   qu ,  228   id ,  228   qd  in accordance with one or more control signals  211   b . The selected I signals  229   iu ,  229   id  are summed (magnitude) in a current summing circuit  230   i  (e.g., a common circuit electrode via which hard-wired current summing occurs) to produce the demodulated I signal  107   i . Similarly, the selected Q signals  229   qu ,  229   qd  are summed (magnitude) in a current summing circuit  230   q  (e.g., a common circuit electrode via which hard-wired current summing occurs) to produce the demodulated Q signal  107   q.    
     Referring to  FIG. 7 , the phasor rotation discussed above can be better understood. At zero degrees, the preferred notation has the I phasor pointing toward 3:00 and the Q phasor pointing toward 12:00, each with a unit length. Initially, the I phasor at zero degrees is composed of +0.99991 and 0.0Q, while the Q phasor at 90 degrees is composed of 0.0I and +0.9999Q. When rotated to 135 degrees, the I phasor is composed of −0.7071I and +0.7071Q, while the Q phasor, now at 225 degrees, is composed of −0.7071I and −0.7071Q. Rotated to 225 degrees, the I phasor is composed of −0.7071I and −0.7071Q, while the Q phasor, now at 315 degrees, is composed of +0.7071I and −0.7071Q. 
     Referring to  FIG. 8 , in accordance with another embodiment of the presently claimed invention, the phase shifter  212   a  for the clock signal  101  ca ( FIG. 3C ) includes frequency dividers  240 ,  242 , multiplexors  244   i ,  244   q  and registers  246   i ,  246   q , substantially as shown. The input clock signal  101   ca  is divided in frequency by a factor of two by the input frequency divider  240 . This ensures that the resulting frequency-divided signals  241   t ,  241   f  have good signal symmetry regardless of the symmetry of the input signal  101   ca . The “true” (i.e., non-inverted) output signal  241   t  is further divided in frequency by a factor of 16 by the second frequency divider  242  (discussed in more detail below), which produces 16 output signals  243 , each of which has a respective one of 16 signal phases adjacent ones of which are mutually equidistant. (Alternatively, a half-frequency incoming clock can be frequency-divided by a factor of eight with 16 individual phases generated by logically ANDing both “true” and “false” register output signals.) Each of the signal multiplexors  244   i ,  244   q , in accordance with its respective one or more control signals  213   i ,  213   q , selects one of the 16 frequency-divided phase signals  243 . Each of the selected signals  245   i ,  245   q , which have mutually quadrature signal phases, are captured by the output registers  246   i ,  246   q  (e.g., D-type flip-flops) in accordance with the “false” (i.e., inverted) frequency-divided clock signal  241   f  to produce “true”  101  it,  101   qt  and “false”  101  if,  101   qf  versions of the I clock signal  101   i  and Q clock signal  101   q.    
     Referring to  FIG. 9 , in accordance with a preferred embodiment, the divide-by-16 frequency divider  242  is implemented as a series  242   a  of registers  250   a ,  250   b ,  250   c ,  250   d ,  250   e ,  250   f ,  250   g ,  250   h  (e.g., each as a D-type flip-flop) interconnected as a ring counter, and clocked by the divide-by-to counter  240   a  (e.g., a D-type flip-flop). The 16 output signals  243  include output signals Ø00, Ø01, Ø02, . . . , Ø15, having signal phases 0.0 degrees, 22.5 degrees, 45.0 degrees, . . . , 337.5 degrees, respectively. Respective pairs of these signals  243  are selected by the multiplexors  244   i ,  244   q , in accordance with the control signals  213   i ,  213   q  such that the selected signals  245   i ,  245   q  have mutually quadrature signal phases. For example, such signal pairs would include Ø00 and Ø12, Ø01 and Ø13, Ø02 and Ø14, Ø03 and Ø15, Ø04 and Ø 0 , Ø 05  and Ø1, Ø06 and Ø2, Ø07 and Ø3, Ø08 and Ø4, Ø09 and Ø5, Ø10 and Ø6, Ø11 and Ø7, Ø12 and Ø8, Ø13 and Ø9, Ø14 and Ø10, or Ø15 and Ø11 for use as the Q clock signal  101   q  and I clock signal  101   i , respectively. 
     With the use of respective pairs of these register output signals  243 , as discussed above, for the I and Q clock signals  101   i ,  101   q , circuit implementation is simplified significantly, since the phase shifters  208   i ,  208   q  ( FIGS. 2 ,  5 A and  5 B) are unnecessary. 
     The signal scalers  110   i ,  110   q  ( FIG. 2 ),  210   i ,  210   q  ( FIGS. 3B and 6 ) can be used with any embodiment of the presently claimed invention to minimize external amplifier gain changes that may be needed as more circuit outputs are summed. 
     Various other modifications and alternations in the structure and method of operation of this invention will be apparent to those skilled in the art without departing from the scope and the spirit of the invention. Although the invention has been described in connection with specific preferred embodiments, it should be understood that the invention as claimed should not be unduly limited to such specific embodiments. It is intended that the following claims define the scope of the present invention and that structures and methods within the scope of these claims and their equivalents be covered thereby.

Technology Category: 5