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Us 4958638
P. 1Us 4958638Us 4958638|Views: 4|Likes: 0Published by Christian UFO IlluminatiMore info:Categories:Types, ResearchPublished by: Christian UFO Illuminati on Jun 16, 2011Copyright:Attribution Non-commercialAvailability:Read on Scribd mobile: iPhone, iPad and Android.download as PDF, TXT or read online from ScribdFlag for inappropriate content|Add to collectionSee moreSee lesshttps://www.scribd.com/doc/58052155/Us-495863807/15/2013pdftextoriginalUnited States PatentSharpe et ale
[54] NON.cONTACf
[llJ [45J
128/653 R 128/653 R
SteTen M. Sharpe, Atlanta; Joseph Seals, Stone Mountain; Anita H. MacDonald, Tucker; Scott R. Crowgey, Avondale Estates, all of Ga.
4,488,559 12/1984 Iskander '4,638,808 1/1987 Mawhinney
Primary Examiner-Kyle L. Howell Assistant Examiner-K. M. Pfaffle Attorney, Agent, or Firm-Hurt, Richardson, Garner, Todd & Cadenhead [57] ABSTRACf
An apparatus for measuring simultaneous physiological parameters such as heart rate and respiration without physically connecting electrodes or other sensors to the body. A beam of frequency modulated continuous wave radio frequency energy is directed towards the body of a subject. The reflected signal contains phase information representing the movement of the surface of the body, from which respiration and heartbeat information can be obtained. The reflected phase modulated energy is received and demodulated by the apparatus using synchronous quadrature detection. The quadrature signals so obtained are then signal processed to obtain the heartbeat and respiratory information of interest. 21 Claims. 6 Drawing Sheets
[73] [21] [22] [51] [52J [58] [56]
Georgia Tech Research Corporation, Atlanta, Ga. Jun. 30, 1988
A61B 5/02; A61B 5/08 128/653 R; 128/671; 128/721 128/653, 716, 721, 670, 128/671, 782
AppI. No.: 213,783 Filed: Int. CLs
References Cited U.S. PATENT DOCUMENTS 3,483,860 12/1969 Namerow
Malech Allen, Jr 128/653 R 128/653 R 128/653 R
3,951,134 4/1976 4,085,740 4/1978
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of an electromagnetically-based approach is that the NON-CONTACf VITAL SIGNS MONITOR system could be designed to simultaneously interrogate the entire chest surface and provide information perGOVERNMENT INTEREST· taining to any respiratory or cardiac function maniThis invention was made with Government support 5 fested as chest wall motions. Conversely, by modifying the antenna design, a localized region of the chest surunder contract No. N 00014-82C-0930 awarded by the face could be interrogated to obtain information about Department of the Navy and under Contract No. F some specific aspect of respiratory or cardiac function. . 33615-83D-0601 awarded by the Department of the Air Such versatility would be difficult to achieve with other Force. The Government has certain rights in the inven10 motion detection techniques. tion. In the prior art the patent to Allen, U.S. Pat. No. BACKGROUND OF THE INVENTION 4,085,740 discloses a method for measuring physiologThis invention relates in general to the 'use of radar ical parameters such as pulse rate and respiration withtechniques to detect minute body movements which are out electrodes or other sensors being connected to the associated with cardiac and respiratory activity. The 15 body. A beam of electromagnetic energy is directed at invention is based on the principle that breathing and the region of interest which undergoes physical disheartbeat produce measurable phase changes in electroplacement representing variations in the parameter to magnetic waves as they reflect off of a living person. be measured. The phase of the reflected energy when The invention offers significant advantages over other compared with the transmitted energy indicates the similar and earlier approaches, including greater sensi- 20 amount of actual physical movement of the body region tivity, lower radiated power, improved reliability and concerned. The method does disclose simultaneous lower cost detection and processing of respiration and heart beat; Functionally, the non-invasive, electromagneticallyhowever, frequency modulation is not used, therefore based Vital Signs Monitor (VsM) is an extremely sensiand the subject must be reasonably still. The receiver tive motion detection system capable of detecting small 25 includes two channels and in one of them the received body jnotions produced by respiratory and cardiac signal is mixed with a signal substantially in quadrature functioning. Motion detection is achieved by transmitwith the transmitted signal to maximize amplitude outting an interrogating electromagnetic field at the target put in those cases in which the received signal is 180 of interest, and then measuring the time-delay-of the return signal reflected back from the surface of the 30 out of phase with the transmitted signal. The patent to Kaplan, et al., U.S. Pat. No. 3,993,995 target. When the target surface is moving, as does the discloses an apparatus for monitoring the respiration of surface of the chest in conjunction with respiratory and a patient without making physical contact. A portion of cardiac activities, corresponding variations will be obthe patient's body is illuminated by a transmitted probe served in the measured time delay. The observed variations can be used to determine motion-related target 35 signal with the reflected echo signal detected by a moniparameters such as displacement and velocity. tor. The phase difference between the transmitted and In the medical field, it is essential that a subject's reflected signals is determined in a quadrature mixer respiration and heartbeat be capable of being measured. which generates outputs indicative of the sine and coThe medical profession is accustomed to voltagesine of the difference signal. These two outputs are derived electrocardiogram waveforms for monitoring 40 coupled to differentiators and when both time derivaheartbeat. Most respiration monitors also require physitives are substantially zero an x-ray unit is triggered cal connection to the subject's body. Many commercialsince it represents an instant of respiration extrema (aply-available devices are available for measuring heart nea). The outputs of the quadrature mixer are also couand respiration rates, but most of them are electrodepled to a direction of motion detector which indicates based requiring physical contact with the subject. De- 45 inhalation or exhalation. vices requiring physical contact, however, are difficult The patent to Kearns, U.S. Pat. No. 4,289,142 disto use on children susceptible to sudden infant death closes a respiration monitor and x-ray triggering appasyndrome (SIDS) or burn patients who cannot tolerate ratus in which a carrier signal is injected into the pathe touch of electrodes. Many infants wear sensors tient's thorax which is indicative of the transthoracic while they sleep that trigger an alarm if their breathing 50 impedance of the patient. This impedance changes as a stops, but electrodes attached to the child can be jarred function of the respiration cycle. The carrier signal is loose as the infant tosses and turns. injected through electrodes coupled to the patient's The invention has similarities with motion-detection thorax. The transthoracic impedance has an alternating systems based on ultrasonic or optical techniques. However, an electromagnetically-based approach offers sev- 55 current component having a respiratory component between 0.2 to 5 ohms and a cardiac component varying eral advantages for monitoring of vital signs-related between 0.02 to 0.2 ohms. motions. For example, with proper antenna design, an The patent to Robertson et al., U.S. Pat. No. interrogating electromagnetic field will suffer minimal 3,524,058 discloses a respiration monitor which uses attenuation while propagating in air (unlike ultrasonic signals which propagate poorly in air). Thus, the elec- 60 body electrodes to direct an electric current to a particular part of the patient's body where changes in electritromagnetically-based Vital Signs Monitor can easily be. cal impedance provide output signals that vary with used in a completely non-contacting mode and can, in respiration. fact, be placed an appreciable distance from the test The patent to Bloice, U.S. Pat. No.3, 796,208 dissubject if required. Electromagnetic signals in the microwave band are also capable of penetrating through 65 closes an apparatus for monitoring movements of a patient including a microwave scanner (doppler radar) heavy clothing. This offers advantages over optical which creates a movement sensitive field surrounding techniques which would have a difficult time of detecting motion through even thin clothing. Another feature part of the patient. Movements of the patient create
disturbances in the field which are monitored and which trigger alarm circuitry. Also in the prior art, apexcardiograms (ACG), which represent a contact technique for measuring small chest surface motions overlying the cardiac apex, have been used to estimate cardiac contractility, left ventricular end-diastolic pressure, pressure changes during atrial systole and cardiac ejection fraction, in addition to diagnosing myocardial wall abnormalities and dysfunction. One of the problems associated with the use of an ACG for the estimation of cardiac function is that the motions recorded are indicative only of activity at the apex of the heart and not of the heart as a whole. Analysis of the VSM waveform is potentially a better choice for estimation of cardiac function since the larger beamwidth of the VSM antenna actually integrates motion over a certain area of the chest. In addition, since the VSM waveform appears to contain information related to aortic and other vascular pulses, it can be. used to measure pulse transit times directly out of the heart into the aorta. This measurement can potentially be used as a non-invasive, non-contact means of estimating blood pressure as discussed by L. A. Geddes, M. Voelz, C. F. Babbs, J. D. Bourland and W. A. Tucker in "Pulse Transit Time as Indicator of Arterial Blood Pressure," Psychophysiology, Vol. 18, No.1, pp. 71-74, 1981. This paper showed that the pulse-wave velocity in the dog aorta increased linearly with increasing diastolic pressure. Similarly, pulse pressures may be related to either the magnitude of the aortic peak in the VSM waveform, or possibly to the rate of rise of this peak. SUMMARY OF THE PRESENT INVENTION Provide an electromagnetic vital signs monitor that can reliably measure simultaneously both heart and respiration rates. It is a further object of this invention to provide a device for measuring physiological parameters without physically contacting the subject with sensors or like attachments. It is a further object of this invention to provide a device for measuring physiological parameters of subjects remotely at distances up to approximately 20 feet. It is a further object of this invention to provide a device for non-contact and non-invasive diagnosis and monitoring capabilities of cardiac, pulmonary, and thoracic mechanical functions resulting from normal or induced physiological responses, trauma, disease or response to therapy. .. It is a further object of this invention to provide a device for measuring remotely the physiological parameters of subjects that are fully clothed and that can be either stationary or moving while sitting or standing. It is a still further object of this invention to provide a device which can be used as an apnea monitor for patients in hospital or clinic intensive care units, or as a patient monitor in bum or trauma clinics or in nursing homes. It is a still further object of this invention to provide a portable device that can be taken into patient areas for the purpose of measuring heart beat and respiration rates. It is a still further object of this invention to detect the presence of persons in visually obstructed areas or under debris resulting from certain disasters. The non-contact electromagnetic vital signs monitor is comprised of a coherent, linear, frequency modulated
continuous wave radar with refmements to optimize the detection of small body movements. The transmitter of the device is frequency modulated by a linear ramp derived from a master clock. The transmitted signal is 5 fed to the radio frequency (RF) network which routes a portion ofthe energy to the antenna which then interrogates the subject. Signals reflected by the subject containing motion-related phase modulation are intercepted by the antenna and applied to the RF network 10 where they are mixed with a portion of the original signal. . The mixing process produces a difference signal which contains harmonics of the original modulating ramp, Each harmonic line is surrounded by sidebands 15 which are related to the body movements. The relative levels of these sidebands are a function of target range, transmitter frequency deviation, and harmonic number. By properly choosing these latter two parameters, signals from a desired range can be detected while others 20 are suppressed. The process is further refmed to result in more ideal range discrimination by multiplication by a weighting function synchronized to the ramp which reduces the range sidelobes. The final synchronous demodulation is accomplished 25 by mixing the received signal (after weighting) with both the in-phase, and quadrature components of the desired harmonic of the modulating ramp which is generated by a synthesizer. After recovery of the in-phase 30 and quadrature components of the received signal, sophisticated digital signal processing can be economically applied since the bandwidths are relatively low. In the preferred embodiment, a high order linear phase fmite impulse response digital filter is used on each 35 channel to reduce the dominance of the strong respiratory signal. A complex autocorrelation is performed from which the rates of interest may be calculated. Still other objects, features and attendant advantages of the present invention will become apparent to those 40 skilled in the art from a reading of the following detailed description of the preferred embodiment, taken in conjunction with the accompanying drawings. BRIEF DESCRIPTION
FIG. 1is a block diagram of the VSM. FIG. 2 is a functional diagram of the modulatorreceiver/demodulator without a weighting circuit. FIG. 3 is a functional diagram of the modulatorreceiver/demodulator with a weighting circuit. FIG. 4 is a schematic diagram of the weighting circuit. FIG. 5 is a schematic diagram of the chopper-based synchronous detectors used in a demodulator. FIG. 6A is a power spectrum for a composite VSM output displaying respiratory and cardiac spectral components. FIG. 6B is a power spectrum for a composite VSM output using logarithmic amplitude and linear frequency scale to demonstrate harmonic structure of cardiac information. FIG. 7A is a VSM return signal from a stationary subject immediately after sampling. FIG. 7B is a result of applying a Fast Fourier Transform to the sampled data. FIG. 7C is a result of applying a digital high pass filter to the VSM return signal. FIG. 7D is a complex autocorrelation of the filter signals.
DETAILED DESCRIPTION INVENTION OF THE
This result provides insight into the operation and potential problems of electromagnetically-based motion The apparatus which is used to carry out the method detection. One useful observation is that only the cosine of the present invention is referred to hereinafter as the 5 of the angular (phase) information of interest is accessiVITAL SIGNS MONITOR (VSM). ble. Thus, for accurate recovery of the desired motion At the electromagnetic frequencies of 3 and 10 GHz information (i.e., R(t», both the nominal distance to the that have been used in the VSM, the surface of the body target and the magnitude of the target motion must is highly reflective to incident electromagnetic fields. In fulfill certain requirements. The nominal target distance addition, biological tissue is very lossy at these frequen- 10 should be set to insure that the average value of the cies and there is minimal penetration of radiated electrophase term in Equation (4) is an odd integer multiple of magnetic energy into the body. Therefore, a return one-half wavelength. This is equivalent to requiring signal from a radiated electromagnetic field incident on that vi(t) and vr(t) be in phase. the body will primarily contain information associated The motion magnitude should also be small enough with events occurring at the body surface. 15 to insure that the motion-related variations in the phase Motion of a target with an electromagneticallyterm do not exceed approximately ±45°. This limits reflective surface can be detected by transmitting an operation to portions of the cosine function that approxinterrogating signal at the target surface, and then meaimate a straight line and insures that Equation (4) prosuring the motion related time-delay of the return signal duces a nearly linear approximation of the motion R(t). that reflects back from the target. The interrogating 20 This requirement is fulfilled if the target motion is small signal travels at the speed of light and the time delay in comparison to the wavelength. of the interrogating experienced by the return signal is equal to the roundelectromagnetic field. From Equation (4), it can be trip distance to the target surface, divided by the speed deduced that the magnitude of the target motion should of light. Thus, the time delay of the return, signal is not exceed approximately ± one-eighth wavelength. proportional to the range or distance to the target sur- 25 The requirements posed by Equation (4) result from the face. If the target is moving in a manner that varies the fact that detection systems based on the preceding target range, variations in the measured time delay can model are not true coherent systems capable of providbe used as a measure of target motion. ing both phase and amplitude information. This probA simple mathematical model can be used to describe 30 lem can be eliminated by splitting vr(t) into two signals. this motion detection phenomenon. Assume that a sinuOne of these signals is demodulated against vi(t) to soidal interrogating signal, vi(t), of the following form is produce the result obtained in Equation (4). The second transmitted at the target of interest, signal is demodulated against a signal in phase quadrature with vi(t) to produce a result of the form vi(t)=A sin(wt). (I) 35 !k'sin(w1). From trigonometric relationships, the sine and cosine terms can then be used to directly determine The return signal, vr(t), reflected from the target surthe desired phase information. face can be represented as, The use of quadrature channels eliminates the previously discussed limitations on the target motion R(t) vr(t)=kA sin w(t+ 1). (2) 40 and development of such a capability has been a focal point of this invention. The parameter k represents losses due to propagation Turning now to the figures in which like numerals attenuation and imPerfect reflection from the target denote like parts, a specific embodiment of the basic surface. The parameter T represents the time delay design for the VSM is shown in FIG. 1. Although there information of interest. To extract this information, a portion of the transmitted signal vi(t) can be used as a 45 are no specific operating frequency limitations on the VSM, systems operating at frequencies of 3 GHz and 10 reference signal to demodulate the return signal vr(t). GHz have been implemented so far. Thus for these By combining vi(t) and vr(t) and passing them through specific implementations each RF section of VSM 10 a nonlinear device such as a mixer or square-law detecincludes: (1) a voltage controllable microwave oscillator, an output signal vo(t) of the following form can be produced, 50 tor 12 to produce a frequency modulated RF signal, (2) a directional coupler 16 to split the voltage controlled microwave oscillators output, (3) fixed attenuators 14 to vo(t)= l k cos(w1). (3) control the radiated power and local oscillator level 15, (4) an antenna 20 for transmitting the interrogating field The parameter k' includes the previously defined parameter k as well as conversion losses associated with 55 and receiving the target return signal, (5) a circulator 18 to recover the return signal from the antenna, (6) a devices (splitters, couplers, mixers, etc.) involved in the double balanced mixer 24 for demodulating the RF demodulation process. return signal to obtain an IF signal (the receiver/Assuming that k' is relatively constant for small tardemodulator performs another demodulation to reget motions, the motion-related information is contained within the argument (i.e., wT) of the cosine func- 60 trieve the phase information from the IF signal); (7) an isolator 22 to prevent local oscillator (LO) levelS to RF tion. R(t) is defmed as the range to the target surface 17 leakage through the double balanced mixer 24 from (the range is time-varying because of the target motion). reaching the antenna 20 and (8) a preamplifier 26 to As previously noted, T is then equal to 2R(t)/c, where minimize noise problems. In addition, a coaxial low-pass c is the speed of light. Using this relationship and noting that 2 pi c/w equals the wavelength (denoted as L), 65 filter is placed on the mixer IF in the 3 GHz system to block LO to IF leakage. Other configurations of hardEquation (3) can be expressed in the form, ware components can be used to mix the transmitted vo(t)=l k cos(4 pi R(t)IL). (4) and return signals to produce the IF result; however,
the embodiment described here is the best mode currently known to the inventors. Frequency modulating the voltage controlled microwave oscillator 12 is an effective and convenient way to reduce the effects of low frequency semiconductor noise. With frequency modulation, the motion-related information of interest that is output by the double balanced mixer 24 appears as sidebands centered at the modulating frequency. By using a modulating frequency of several kHz, the largest portion of the lowfrequency flicker noise is avoided and greater receiver sensitivity is achieved. The modulator 28 provides a time-varying waveform that can be used to frequency modulate the voltage controlled microwave oscillator 12 in the RF section of the VSM. The main advantage of frequency modulating the voltage controlled microwave oscillator 12 is that it makes it possible to achieve greater receiver sensitivity, which in turn, enables the use of lower and thus safer, radiated power levels. The modulator 28 can be conveniently divided into two sections: a digital section 32 which is used to provide all timing information, and an analog section 34 which is used to form the actual ramp waveform used as the modulating signal. A functional diagram is shown in FIGS. 2 and 3. An operational amplifier-based Howland integrator in ramp circuit 36 is used to create the basic ramp waveform from inputs from a oneshot 55 and a voltage divider 38. This particular integrator is used to provide a simple means of generating a nonlinear ramp to compensate for the nonlinear tuning curve of the voltage controlled microwave oscillator 12, thereby producing a relatively linear frequency ramp. Alternative ramp generator circuits could be used as well. A transistor in ramp circuit 36 controls the charging of the integrator. When the input to the transistor is zero volts, the transistor cuts off and a capacitor in ramp circuit 36 charges to 4.5 V (the reference voltage which is the output 39 of the voltage divider 38). The current that charges the capacitor consists of two parts: a constant current produced by the reference input voltage and input resistor, and a variable current which is the ratio of the voltage being amplified by the operational amplifier and a resistance R which is a function of both the input resistor and a variable resistor. The variable resistor is adjusted so that R may be either positive or negative. The result is that the ramp may be expressed in the form
r{t)=Vo(l-exp( -tIRC)
4.958,638
where Vo is a constant. When R is positive, the ramp bows upward; and when R is negative the ramp bows downward. A high input ( + 5 V) to the transistor saturates the transistor and rapidly discharged the capacitor, resetting the ramp. The combination of the transistor and the Howland integrator in this particular embodiment produces a basic modulating ramp waveform. However, additional circuitry controls the amplitude and offset level of the modulating waveform and compensates for nonlinearities of the voltage controlled microwave oscillator 12. The ramp amplitude controls the total frequency range over which the voltage controlled microwave oscillator 12 is modulated. The offset level controls the center frequency. At the output of the Howland integrator, a potentiometer acting as a voltage divider controls the amplitude of the ramp. The offset level of the ramp waveform is controlled by a summing net-
work 50 that can be used to add or subtract a DC level to the output of the integrator as can be seen in FIG. 2. The ability to control the ramp amplitude and offset level is important "toinsure that the 5 modulating frequency is centered at an appropriate value and to insure that operation is performed over a suitable linear portion of the voltage controlled microwave oscillator's 12 tuning response. The digital timing circuit 32 generates the signal re10 quired to control charging and discharging of the integrator as can be seen in FIG. 2. In the preferred embodiment, a 1 MHz crystal oscillator 40 is used as a stable timing reference. The output of this crystal oscillator 40 is input to a binary ripple counter 42 that divides the 1 15 MHz crystal frequency by a factor of 2n, where n is an integer between 6 and 9. The resultant signal from the ripple counter 42 is then fed into a two-stage walking ring counter 44 where two significant operations occur. The walking ring counter 44 provides an additional 20 division off our in frequency. The output of the walking ring counter 44 is at the modulating frequency, which is one of the following: 4 kHz, 2 kHz, 1 kHz, or 500 Hz. In addition, the walking ring counter 44 outputs two peri25 odic square waves on lines 46, 48 that are exactly in phase quadrature (relative phase difference of 90°). These quadrature signals are essential to the synchronous detectors 58 in the receiver/demodulator 30 which provide the VSM 10 with a quadrature channel 30 capability. The VSM 10 employs a receiver consisting of a frontend double-balanced mixer 24 and a receiver/demodulator that serves as a narrow band detector by demodulating the IF signal to obtain the phase informa35 tion; this is the second demodulation performed in the VSM. With the FM-CWapproach used in the VSM 10, motion-related information output from the mixer appears as sidebands centered around a carrier frequency of 4 kHz. The carrier frequency is an integral multiple 40 of the modulating frequency. An appropriate narrow band detection scheme is required to extract these motion-related sidebands. The receiver/demodulator 30 represents the required detection scheme and consists of a low noise preamplifier 26 that provides needed gain and filtering at the 45 carrier frequency, an optional weighting circuit 52 to attenuate extraneous returns, an inverting amplifier 54, a band pass filter 56 to remove unwanted frequency components (especially third order harmonics) prior to 50 synchronous detection, a pair of synchronous detectors, shown within block 58, that enable coherent detection of both amplitude and phase information, and a pair of band pass filters 60 to remove unwanted frequency components in the outputs of the synchronous detectors 55 (DC components and high frequency mixing products). A block diagram of the receiver/demodulator 30 is contained in the lower half of the VSM block diagrams in FIGS. 2 and 3. Use of the weighting circuit 52 improves the ability of 60 the ramp-based frequency modulation technique used in the VSM 10 to reject extraneous return signals. With this frequency modulation technique, the demodulation outPut of the VSM 10 contains sinusoidal bursts due to returns from the subject being evaluated. Thesesinusoi65 dal bursts repeat at a rate equal to the modulating frequency and the frequency of these bursts is a function of the subject's distance from the VSM 10 (range). However, discontinuities exist at end points of each sinusoi-
dal burst due to recycling of the modulating ramp. pass filter 56 Provides two signals identical in amplitude These discontinuities cause the demodulated return but opposite in phase. These signals each have a fresignal to have a spectrum containing harmonics of the quency of 4 kHz and contain low frequency' side bands modulating frequency. The undesired harmonic compocorresponding to the motion-related information of nents, also know as range sidelobes, enable return sig- 5 interest. The availability of the out-of-phase signals naIs from other subjects to generate frequency compoenable synchronous detection using a conveniently imnents at the receiver frequency. plemented chopper approach. Weighting functions substantially reduce the effects A model of a chopper-based detector 58 is shown in of these discontinuities. To achieve range sidelobe supFIG. 5. Because of its out of phase outputs, the bi-quad pression in this manner, a demodulated return signal is 10 bandpass filter 56 is represented as a center tapped multiplied by a weighting function which is synchrotransformer 68. The chopper (electronic switch) 70 is nized with a sinusoidal burst in the demodulated return switched between the two outputs of the transformer 68 signal. Weighting functions are usually bell-shaped and at a frequency of 4 kHz. Since the chopper frequency is have a value of unity at their center with ends that taper identical to that of the information from the transformer 15 68, the sum and difference frequency terms outputted by the chopper 70 contain a DC term (difference frequency component) and a 8 kHz term (sum frequency component). The low pass filter 72 following the chop20 per is used to remove the undesired sum frequency component. The remaining DC term is dependent on the phase difference between the information signal (55 or 57) into the transformer 68 and the reference signal (46 or 48) controlling switching of the chopper. 25 For the demodulator 30 in the VSM 10, a synchronous detector 58 is employed. A model of the detector 58 is synchronous to the model shown in, FIG. 5. The reference signals 46 and 48 into the synchronous detector 58 have a phase difference of 90° (the quadrature30 phase reference signals are provided by the walking ring counter 44 in the, modulator subsystem). Thus, the one of the outputs 71 from the synchronous detector 58 can be considered an in-phase or I channel term that is (6) w(t)=0.50-0.50 cos(2 pi t). O<t< 1 equivalent. to the cosine of the phase of the information 35 signal. The other output 73, from the synchronous deHamming: tector 58 can be considered a quadrature or Q channel w(t)=0.54-0.46 cos(2 pi t), O<t< 1 (7) term equivalent to the, sine of the information signal. The I and Q signals, 71 and 73, can be evaluated jointly Pulses at the modulating frequency are input into the, or independently to extract the motion-related informa- operational amplifier-based bi-quad low pass filter 62. tion of interest. Alternatively, many other low pass filter designs can be 40 The phase of the demodulated return signal varies as implemented to achieve the same result. The low pass a function of target motion. Therefore, the I and Q filter 62 attenuates the higher ordered harmonics of the signals, 71 and 73, that are output by the synchronous pulses, generating. a sinusoidal signal. The potentiomedetector 58 are not true DC terms. Instead, these signals ter 90 at the input varies the input signal level so that the amplitude of the sinusoidal term in the weighting func- 45 occupy a frequency band related to that of the motion being detected. Since the VSM 10 was designed to tion can be varied depending on the choice of weighting provide an almost linear estimate of target motion, the function. A summing junction 91 allows a DC offset frequency band of the I and Q channels is the same as determined by another potentiometer 92 to be added to that of the respiratory and cardiac motions, being, dethe sinusoidal term and generates the desired weighting function. The remaining potentiometer 92 involved 50 tected. Thus, filters 60 used on the outputs, 71 and 73 of the synchronous detector must have sufficient bandwith the generation of the weighting function sets the width to pass information in the respiratory and cardiac cutoff frequency of the low-pass bi-quad bandpass filter bands but must be narrow enough to reject unwanted 56 precisely. The transistor acts as a voltage-to-current noise and mixing products. converter, supplying a current proportional to the weighting function to the CA3080 transconductance 55 Lowpass filters 72 and bandpass filters 60 with passbands of approximately 0.1-75 Hz are used to filter the amplifier 64. The CA3080 64 essentially multiplies the outputs 71 and 73 of the synchronous detector 58. The weighting function by the signal output 27 from the 0.1 Hz lower frequency cutoff of filter 60 effectively preamplifier. The output of the weighting circuit 52 is blocks DC terms but is low enough to pass slow respirathen input to the receiver 30. The bi-quad bandpass filter 56 provides additional 60 tory information. The 75 Hz low-pass cutoff frequency filter 72 blocks the 8 kHz sum frequency term generated rejection of DC and low frequency components and by the synchronous detectors 58 but is high enough to also suppresses any harmonics of the modulating frepass fast cardiac motion. In addition, the 75 Hz cutoff quency that are generated by the double-balanced mixer should make it possible to determine if vibrations associ24. The bi-quad design permits the cutoff frequency of the filter 56 to be tuned by adjustment of a single feed- 65 ated with respiratory and cardiac sounds can be detected. Cascaded together, the filters 72, 60 have such a back transistor and has the unique characteristic of wide passband that for all practical purposes, the data maintaining a constant absolute bandwidth as it is tuned. The inverting amplifier stage used in the bi-quad bandoutput may be considered unprocessed. Therefore, suit!~rar~= ::~ Y:~;:~~:rer~~~~f ~~e:~=~ discontinuities, but the weighting procedure reduces their significance and results in a more ideal spectrum. The VSM 10 may be configured with or without weighting. As shown in FIG. 2, when weighting is not used, the ramp frequency equals the, receiver frequency, 4 kHz. FIG. 3 shows that, when weighting is used, the ramp frequency equals 500 Hz, 1 kHz or 2 kHz. The weighting circuit 52 is shown in FIG. 4. The weighting circuit 52 has been designed so that the detected signal 27 from thepreamplifier 26 may be multiplied by one of two weighting functions, the Hamming window or the Hanning window. Other windows could be chosen and the two that have been implemented are not a limitation of the VSM 10. The equations for each of the windows are given below: Hanning:
of the respiratory-related signal is a factor of 5-20 times able signal processing techniques must be used to exgreater than that of the corresponding cardiac-related tract useful information from the output data. signal. Thus, filters used to recover cardiac information In order to digitally process the output data, the I and in the presence of a strong respiratory signal must Q channels are sampled at a rate of 100 Hz by a sampler included in the Digital Sampling and Processing block 5 greatly attenuate the low-frequency band occupied by the respiratory information. This requires filters with a SO, shown in FIG. 1. Prior to sampling, the I and Q sharp-rolloff (i.e., a large number of poles). Because of channels are input to a pair of 34 Hz second order low the potentially small separation between the respiratory pass anti-aliasing filters, which are included in digital and cardiac information bands, it is necessary that a sampling block SO in FIG. 1. Without these anti-aliasing filters, it would be possible for signal components over 10 filter used to recover cardiac information have a precise cutoff frequency. In addition, since respiratory and 50 Hz, half the sampling rate, to take on the identity of cardiac rates will vary significantly, it is necessary that lower frequencies, resulting in signal distortion. the filter's cutoff frequency be adaptive or at least adTypically, respiratory rates of normal subjects correjustable. The limited capabilities of simple analog filters spond to frequencies of approximately 0.12-0.30 Hz (7-18 breaths per minute) while cardiac rates corre- 15 appear inadequate for this task. Thus, a digital signal processing capability is required for the VSM to spond to approximately 0.8-1.5 Hz (48-90 beats per achieve reliable rate determination and to produce diagminute). Since there is more than an octave difference nostically useful respiratory and cardiac waveforms. between the highest respiratory frequency and the lowWhat follows is a description of the signal processing est cardiac frequency, it is possible to examine the individual respiratory and cardiac components. 20 methods used in the VSM, but those skilled in the art will understand alternative implementations are possiFIG. 6A shows a power spectrum for a composite ble. VSM return signal obtained with a laboratory signal In order to determine the cardiac rates, the analog I analyzer. The linear amplitude and logarithmic freand Q signals are first sampled at a 100 Hz rate and then quency scales permit detailed evaluation of the respiratory spectrum. The fundamental respiratory component 25 high pass ftltered digitally with a cutoff frequency behas an RMS amplitude of280 millivolts while the fundatween 0.75 and 1.0 Hz. These filters, though not shown mental cardiac component has an RMS amplitude of individually, are included with the Digital Sampling only 17.9 millivolts. The respiratory component occurs and Processing block of FIG. 1. The complex autocorat 0.21 Hz corresponding to a respiratory rate of aprelation to the two filtered signals is then computed, as proximately 12 to 13 breaths per minute. The smaller 30 shown in FIG. 1 by the Autocorrelation block S2. Periodicities in the signal due to cardiac related motions cardiac spectral component occurs at 1.18 Hz correappear as relative maxima in the real part of the autosponding to the subject's heart rate of approximately 70-71 beats per minute. Harmonics of the respiratory correlation function. The signal processing is not perfundamental can also be observed at 0.42 and 0.63 Hz. formed in real time. The second harmonic of the respiratory motion is ap- 35 Digital signal processing techniques can be divided proximately 9 dB lower than the fundamental respirainto two types: time domain techniques and frequency tory component. The third harmonic is approximately domain techniques, both of which are well known in the 21 dB below the fundamental respiratory component. art of signal processing. With frequency domain techKnowledge of the relative strengths of the individual niques, the digitized I and Q channel signals are transharmonic components can be used to identify specific 40 formed into the frequency domain, either with a Fast features in the corresponding time based signal. Such Fourier Transform (FFT) or some non Fourier spectral information can be useful for characterizing or classifyestimation algorithm, and a peak detection algorithm ing the breathing pattern. By changing to a more convethen estimates the frequency of the spectral line which nient logarithmic amplitude scale and a linear frequency corresponds to the cardiac rate. Time domain techscale, it is possible to better evaluate the cardiac spec- 45 niques require the processing of time domain signals trum as shown in FIG. 6B. In this figure, wider frewith filtering and autocorrelation algorithms to obtain quency resolution has resulted in a smearing of the other time domain signals in which the time period respiratory portion of the spectrum so that only a single between relative maxima can be more easily detected respiratory component appears at a frequency of apthan in the original signal. Time domain techniques proximately 0.15 Hz. It can be seen that the fundamen- 50 require less processing time to obtain an initial estimate tal, second, third, fifth and possibly sixth harmonics are of heart rate and, therefore, are preferred to frequency strong while the fourth harmonic is partially supdomain techniques. FIGS. 7A-D illustrate the process pressed. Again, knowledge of such spectral information of heart rate termination. The I and Q channels from the is valuable for evaluating aspects of cardiac function. output of the VSM are shown in FIG. 7A. These two One of the difficulties with spectral analysis is that the 55 signals are transformed into the frequency domain using rates determined actually represent the average rate of a Fast Fourier Transform (FFT) algorithm. The resulseveral events. With average but not instantaneous rate tant signals are depicted in FIG. 7B. FIG. 7C shows the values available, it is difficult to compute useful parameI and Q channel signals after they are digitally high pass ters such as heart rate variability. filtered, The cutoff frequency is 1 Hz. The complex If cardiac and respiratory functions are to be evalu- 60 autocorrelation of the filter signals is shown in FIG. 70. ated from familiar time-based waveforms, filtering techThe real part of the complex autocorrelation is used to niques capable of separating the composite signal from calculate the rates displayed at the bottom of the figure. the VSM .10 into its individual respiratory and cardiac The digital filtering algorithm employed is the window components are required. In practice, it is relatively technique. More complex filter design procedures can simple to build filters to extract the desired respiratory 65 achieve better filters for a given filter length. More component while several factors combine to make respecifically, these more complex filter designs can yield covery of cardiac information difficult to achieve. For filters with similar performance to the filters used to example, for most test subjects the peak-to-peak levels obtain the data in FIG. 7C, but the filter length could be
receiving means for receiving said frequency modureduced substantially, therefore decreasing computalated beam as a motion-related, phase modulated tion time. reflected signal from said body portion; and Comparison of FIGS. 7C and 7D reveals that the signal processing means for extracting the heart and autocorrelation process yields a signal more suitable for respiration rates from said phase modulated rea peak detection algorithm. The autocorrelation algo- 5 rithm used is based on a method described by G. Hoflected signal. . 2. The apparatus of claim 1 wherein the transmitting shal, M. Siegel and R. Zapp in "A Microwave Heart means comprises: Monitor and Life Detection System," published in 1984 voltage controlled oscillator means to produce a conin IEEE Frontiers of Engineering and Computing in tinuous wave radio frequency signal output; Health Care on pp. 331-333. The method described by 10 modulator means for providing a time-varying ramp Hoshal et al. is incorporated by reference herein. Each waveform. to frequency modulate said continuous of the eight autocorrelation segments shown in FIG. 7D wave radio frequency signal output of said voltage was computed for lags between 60 and 120. Since the controllable oscillator means; sampling frequency is 100 Hz, the sampling period is 10 milliseconds, and an autocorrelation lag of 60 corre- 15 directional coupler means to split the oscillator's output into a first signal for transmission and a sponds to a time delay of 60 (10 ms)=0.6 seconds, or to second signal for mixing with said phase modulated a heart rate of 1.67 Hz or 100 beats per minute. Simireflected signal; larly, an autocorrelation lag of 120 corresponds to a attenuator means to control the radiated power of heart rate of 50 beats per minute. said first signal output and said second signal outFour hundred samples of the filtered VSM output are 20 put; and antenna means for transmitting said first used to compute each of the 8 autocorrelation segments. signal output toward said body portion. As a result, four seconds of data are required to com3. The apparatus of claim 2 wherein said modulator pute each heart estimate shown in FIG. 7D. If the nummeans comprises a digital timing circuit means for prober of samples is too large, the aperiodic nature of the heart beat signal tends to flatten the autocorrelation. If 25 viding a first reference signal and a second reference signal, said reference signals having a relative phase the number of samples is too small, the record might not include an entire heart beat cycle. Each successive autodifference of 90°• 4. The apparatus of claim 3 wherein said modulator correlation calculation starts at 100 points (correspondmeans further comprises an analog ramp generator ciring to one second) further in the data record. The contrast between 7C and 7D clearly demonstrates the abil- 30 cuit means for forming the time-varying ramp waveity of the autocorrelation to enhance the presence of the form used to frequency modulate said radio frequency signal output of said voltage controllable oscillator heart beat signal. While the invention has been described with respect means. 5. The apparatus of claim 3 wherein the receiving to a preferred physical embodiment constructed in accordance therewith, the same is by way of example only 35 means comprises: and is not to be taken by way of limitation. It will be antenna means for receiving said phase-modulated apparent to those skilled in the art that various modifireflected signal from said subject; mixing means for combining said received phasecations and improvements may be made without departmodulated reflected signal with said second signal ing from the scope and spirit of the invention. For example, while specific frequencies or frequency ranges 40 output from said voltage controllable oscillator have been given for various components within the means to produce a different signal output containVSM 10 it will be evident to those skilled in the art that ing harmonic frequency components of the timevarying ramp weigh form; other embodiments can be developed employing different frequencies or frequency ranges. Thus, specific oppreamplifier means for amplifying and filtering noise erating frequencies of the voltage controlled micro- 45 from the different signal output from said mixing wave oscillator 12, crystal oscillator 40 filters 56, 60, 72 means to produce a detected signal; and are not limitations of the VSM 10. Similarly, the VSM demodulator means for synchronously detecting inis not limited by the. type of ramp circuitry 36, phase and quadrature components from said deweighting circuitry 52, liquid low pass filter 62, or syntected signal. chronous detection circuitry 58 disclosed herein. Other 50 6. The apparatus of claim 5 wherein the demodulator digital sampling rates may also be used with the VSM means comprises: 10. Furthermore, while the present system has been a first bandpass filter means to remove said harmonic described as applicable for the non-contact measurefrequency components from said detected signal ment of heart rate and respiratory rate, it will be evident output from said preamplifier means and said first to those skilled in the art that the present system may be 55 bandpass filter means providing a first detected used to measure other types of potentially useful diagoutput signal and a second detected output signal; nostic information concerning cardiac and respiratory a first synchronous detector having inputs comprismechanical functions which can be derived from the ing said first reference signal from said digital timVSM waveform by appropriate signal processing. Acing circuit means and said first detected output cordingly, it is to be understood that the invention is not 60 signal, and the output of said first synchronous limited by the specific illustrative embodiment, but only detector comprising the in-phase component of by the scope of the appended claims. said detected signal; What is claimed is: a second synchronous detector having inputs com1. An apparatus for measuring simultaneously the prising said second reference signal from said digiheart and respiration rates of a subject comprising: 65 tal timing circuit means and said second detected transmitting means for directing a beam of frequency output signal, and the output of said second synmodulated, continuous wave radio frequency enchronous detector comprising a quadrature comergy towards a body portion of said subject; ponent of said detected signal; and
4,958,638·
duce a frequency modulated, continuous wave a plurality of second bandpass filters to filter the radio frequency signal; output of each of said synchronous detectors. 7. The apparatus of claim 6 wherein the demodulator splitting said frequency modulated, continuous wave means further comprises a weighting circuit to'suppress radio frequency signal into a first signal for transrange sidelobes in said detected signal by multiplying 5 mission and a second signal for mixing with said said detected signal by a weighting function to form a phase modulated reflected signal; weighted detected signal. controlling the radiated power of said first signal; and 8. The apparatus of claim 7 wherein the weighting transmitting said first signal towards the body portion function consists of a sinusoidal term and a DC offset of said subject. term which describes the Hanning window. 10 15. The method of claim 14 wherein the step of mod9. The apparatus of claim 7 wherein the weighting ulating further comprises generating a pair of reference function consists of a sinusoidal term and a DC offset signals having, a relative phase difference of 90·. term which describes the Hamming window. 16. The method of claim 15 wherein the step of re10. The apparatus of claim 9 wherein the weighting ceiving a motion-related, phase modulated reflected circuit comprises an operational amplifier-based bi-quad 15 signal comprises: lowpass filter having a first potentiometer connected to receiving said frequency modulated beam as phasethe input of said bi-quad low pass filter to vary the modulated reflected signal from said subject; amplitude of the sinusoidal term in the weighting funcmixing said phase-modulated reflected signal with tion, a second potentiometer connected to the input of said second signal to produce a difference signal said bi-quad low pass filter to generate a DC offset term, 20 output containing harmonics of said time-varying a summing junction connected to the input of said biramp waveform; quad low pass filter to add the sinusoidal term and the amplifying and filtering said difference signal output DC offset term, and a third potentiometer connected to produce a detected signal; between the input and the output of said bi-quad low demodulating said detected signal to produce inpass filter to set precisely the cutoff frequency of said 25 phase and quadrature components of said phase bi-quad lowpass filter. modulated reflected signal. 11. The apparatus of claim 6 wherein the signal pro17. The method of claim 16 wherein the step of de" cessing means comprises: modulating said detected signal comprises:, a plurality of sampling means for measuring the filfiltering said detected signal to remove from said tered outputs of each of said synchronous detectors 30 detected signal the harmonic frequency compoat a specified rate and producing a plurality of nents; digital sampled outputs; inverting said filtered detected signal to produce a a plurality of digital ftltering means for passing each first filtered signal and a second ftltered signal, said of said plurality of digital sampled outputs above a first filtered signal and said second filtered signal cutoff frequency to produce a plurality of filtered 35 being equal in amplitude but opposite in phase; digital sampled outputs; and mixing said first filtered signal with the first of said algorithm means for computing the autocorrelation pair of said reference signals to form an in-phase function of the plurality of said filtered digital samoutput signal; pled outputs. • mixing said second filtered signal with the second of 12. The apparatus of claim 5 further comprising: 40 said pair of reference signals to form a quadrature a single antenna means for transmitting a beam of output signal, frequency modulated, continuous wave radio freftltering the- in-phase output signal to remove DC quency energy and for receiving a phase moduand high frequency mixing components to form a lated reflected signal; filtered in-phase signal; arid a circulator means for recovering said received phase 45 filtering the quadrature output signal to remove DC modulated reflected signal from 'said antenna and high frequency mixing components to form a means; and filtered quadrature signal. isolator means to prevent leakage of said second sig18. The method of claim 16 wherein the step of denal output from said voltage controlled oscillator means from reaching said antenna means. 50 modulating said detected signal includes multiplying said detected signal output by a weighting function to 13. A method for measuring simultaneously the heart suppress range sidelobes contained in said detected and respiration rates of a subject comprising the steps signal. of: 19. The method of claim 18 wherein the step of protransmitting a beam of frequency modulated, continuous wave radio frequency energy towards a body 55 cessing the phase-modulated reflected signal to extract the heart rate and respiration rates comprises: portion of said subject; sampling the ftltered in-phase signal at a specified rate receiving said frequency modulated beam as a moto produce a digital sampled in-phase output; tion-related, phase modulated reflected signal from sampling the ftltered quadrature signal at a specified said body portion of said subject; and rate to produce a digital sampled quadrature outprocessing the phase modulated reflected signal to 60 put; extract the heart and respiration rates from said high-pass filtering the digital sampled in-phase output phase modulated reflected signal. to remove low frequency components; 14. The method claim 13 wherein the step of transmitting a beam of frequency modulated, continuous wave high-pass filtering the digital sampled quadrature radio frequency energy comprises: 65 . output to remove low frequency components; and producing a continuous wave radio frequency signal; performing an autocorrelation on the high-pass filmodulating said continuous wave radio frequency tered digital sampled in-phase output and the highsignal with a time-varying ramp waveform to propass filtered digital sampled quadrature output.
signal processing means for extracting the diagnostic information of interest from said motion-related tion regarding the mechanical function of the heart, phase modulated reflected signal. 21. A method for measuring diagnostic information lungs, chest wall, or any anatomical part of the body of 5 regarding the mechanical function of the heart, lungs, a subject undergoing motion comprising: chest wall, or any anatomical part of the body of a subject undergoing motion comprising: transmitting means for directing a beam of frequency transmitting a beam of frequency modulated, continumodulated, continuous wave radio frequency enous wave radio frequency energy towards a seergy towards a selected body portion of said sub- 10 lected body portion of said subject; receiving said frequency modulated beam as a moject; tion-related, phase modulated reflected signal from receiving said frequency modulating beam as means said selected body portion of said subject; and processing the phase modulated reflected signal to for receiving a motion-related, phase modulated 15 extract the diagnostic information of interest from reflected signal from said selected body portion; said phase modulated reflected signal. and ***** 20. An apparatus for measuring diagnostic informa20
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