Source: https://patents.google.com/patent/US8233858B2/en
Timestamp: 2019-08-22 00:59:49
Document Index: 10074330

Matched Legal Cases: ['Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60']

US8233858B2 - RF power transmission, modulation, and amplification embodiments, including control circuitry for controlling power amplifier output stages - Google Patents
RF power transmission, modulation, and amplification embodiments, including control circuitry for controlling power amplifier output stages Download PDF
US8233858B2
US8233858B2 US11/636,970 US63697006A US8233858B2 US 8233858 B2 US8233858 B2 US 8233858B2 US 63697006 A US63697006 A US 63697006A US 8233858 B2 US8233858 B2 US 8233858B2
US11/636,970
US20070096806A1 (en
2007-05-03 Publication of US20070096806A1 publication Critical patent/US20070096806A1/en
2012-07-31 Publication of US8233858B2 publication Critical patent/US8233858B2/en
The present application is a continuation of pending U.S. patent application Ser. No. 11/508,989 filed Aug. 24, 2006, which claims the benefit of U.S. Provisional Patent Application No. 60/794,121 filed on Apr. 24, 2006, U.S. Provisional Patent Application No. 60/797,653 filed on May 5, 2006, and U.S. Provisional Patent Application No. 60/798,705 filed on May 9, 2006, and is also a continuation-in-part of U.S. patent application Ser. No. 11/256,172, filed Oct. 24, 2005, which claims the benefit of U.S. Provisional Patent Application No. 60/620,972 filed on Oct. 22, 2004, U.S. Provisional Patent Application No. 60/671,542 filed on Apr. 15, 2005, U.S. Provisional Patent Application No. 60/671,536 filed on Apr. 15, 2005, U.S. Provisional Patent Application No. 60/673,397 filed on Apr. 21, 2005, U.S. Provisional Patent Application No. 60/706,003 filed on Aug. 8, 2005, U.S. Provisional Patent Application No. 60/709,092 filed on Aug. 18, 2005, U.S. Provisional Patent Application No. 60/717,244 filed on Sept. 16, 2005, and U.S. Provisional Patent Application No. 60/721,114 filed on Sept. 28, 2005, all of which are incorporated herein by reference in their entireties.
A high-level overview of vector power amplification is now provided. FIG. 1D illustrates the power amplification of an exemplary time-varying complex input signal 172. Signals 114 and 126 as illustrated in FIGS. 1A and 1B may be examples of signal 172. Further, signal 172 may be generated by or comprised of two or more constituent signals such as 104 and 106 (FIG. 1A), 108 and 112 (FIG. 11A), 116 and 118 (FIG. 1B), and 122 and 124 (FIG. 1B).
b 2.1) Phasor Signal Representation
FIG. 1 illustrates a phasor representation {right arrow over (R)} 102 of a signal r(t). A phasor representation of a signal is explicitly representative of the magnitude of the signal's envelope and of the signal's phase shift relative to a reference signal. In this document, for purposes of convenience, and not limitation, the reference signal is defined as being aligned with the real (Re) axis of the orthogonal space of the phasor representation. The invention is not, however, limited to this embodiment. The frequency information of the signal is implicit in the representation, and is given by the frequency of the reference signal. For example, referring to FIG. 1, and assuming that the real axis corresponds to a cos(ωt) reference signal, phasor {right arrow over (R)} would translate to the function r(t)=R(t) cos(ωt+φ(t)), where R is the magnitude of {right arrow over (R)} .
Still referring to FIG. 1, it is noted that phasor {right arrow over (R)} can be decomposed into a real part phasor {right arrow over (I)} and an imaginary part phasor {right arrow over (Q)} . {right arrow over (I)} and {right arrow over (Q)} are said to be the in-phase and quadrature phasor components of {right arrow over (R)} with respect to the reference signal. It is further noted that the signals that correspond to {right arrow over (I)} and {right arrow over (Q)} are related to r(t) as I(t)=R(t)·cos(φ(t)) and Q(t)=R(t)·sin(φ(t)), respectively. In the time domain, signal r(t) can also be written in terms of its in-phase and quadrature components as follows:
FIG. 3C, it can be noted that the resulting signal r(t) has a time-varying envelope. Further, it is noted, from FIG. 3C, that r(t) undergoes a reversal in phase at the moment when the modulating signal m(t) crosses zero. Having both non-constant envelope and phase, r(t) is said to be a time-varying complex envelope signal.
Still referring to FIG. 4, at instant t1, phasor {right arrow over (I1)} can be obtained by the sum of upper and lower phasors {right arrow over (IU 1 )} and {right arrow over (IL 1 )}. Similarly, at instant t2, phasor {right arrow over (I2)} can be obtained by the sum of upper and lower phasors {right arrow over (U 2 )} and {right arrow over (IL 2 )}. Note that phasors {right arrow over (IU 1 )} and {right arrow over (IU 2 )} have equal or substantially equal magnitude. Similarly, phasors {right arrow over (IL 1 )} and {right arrow over (IL 2 )} have substantially equal magnitude. Accordingly, the real part phasor of the time-varying envelope signal can be obtained at any time instant by the sum of at least two substantially constant envelope components.
ϕ 1 2 = cot - 1 ( I 1 2 ⁢ 1 - I 1 2 4 ) ; ⁢ and ( 2 ) ϕ 2 2 = cot - 1 ( I 2 2 ⁢ 1 - I 2 2 4 ) , ( 3 )
{right arrow over (R)}={right arrow over (IU)}+{right arrow over (I L)}+{right arrow over (Q U)}+{right arrow over (Q L)};
r ⁡ ( t ) = I U ⁡ ( t ) + I L ⁡ ( t ) + Q U ⁡ ( t ) + Q L ⁡ ( t ) ; I U ⁡ ( t ) = sgn ⁡ ( I → ) × I U × cos ⁡ ( ϕ I 2 ) × cos ⁡ ( ω ⁢ ⁢ t ) + I U × sin ⁡ ( ϕ I 2 ) × sin ⁡ ( ω ⁢ ⁢ t ) ; I L ⁡ ( t ) = sgn ⁡ ( I → ) × I L × cos ⁡ ( ϕ I 2 ) × cos ⁡ ( ω ⁢ ⁢ t ) - I L × sin ⁡ ( ϕ I 2 ) × sin ⁡ ( ω ⁢ ⁢ t ) ; Q U ⁡ ( t ) = - sgn ⁡ ( Q → ) × Q U × cos ⁡ ( ϕ Q 2 ) × sin ⁡ ( ω ⁢ ⁢ t ) + Q U × sin ⁡ ( ϕ Q 2 ) × cos ⁡ ( ω ⁢ ⁢ t ) ; Q L ⁡ ( t ) = - sgn ⁡ ( Q → ) × Q L × cos ⁡ ( ϕ Q 2 ) × sin ⁡ ( ω ⁢ ⁢ t ) - Q L × sin ⁡ ( ϕ Q 2 ) × cos ⁡ ( ω ⁢ ⁢ t ) . ( 5 )
where sgn({right arrow over (I)})=±1 depending on whether {right arrow over (I)} is in-phase or 180° degrees out-of-phase with the positive real axis. Similarly, sgn({right arrow over (Q)})=±1 depending on whether {right arrow over (Q)} is in-phase or 180° degrees out-of-phase with the imaginary axis
corresponds to the phase shift of {right arrow over (QU)} and {right arrow over (QL)} relative to the imaginary axis. φ1/2 and
r ⁡ ( t ) = ⁢ I U ⁡ ( t ) + I L ⁡ ( t ) + Q U ⁡ ( t ) + Q L ⁡ ( t ) ; I U ⁡ ( t ) = ⁢ sgn ⁡ ( I → ) × I UX × cos ⁡ ( ω ⁢ ⁢ t ) + I UY × sin ⁡ ( ω ⁢ ⁢ t ) ; I L ⁡ ( t ) = ⁢ sgn ⁡ ( I → ) × I UX × cos ⁡ ( ω ⁢ ⁢ t ) - I UY × sin ⁡ ( ω ⁢ ⁢ t ) ; Q U ⁡ ( t ) = ⁢ - Q UX × cos ⁡ ( ω ⁢ ⁢ t ) + sgn ⁡ ( Q → ) × Q UY × sin ⁡ ( ω ⁢ ⁢ t ) ; Q L ⁡ ( t ) = ⁢ Q UY × cos ⁡ ( ω ⁢ ⁢ t ) - sgn ⁡ ( Q → ) × Q UY × sin ⁡ ( ω ⁢ ⁢ t ) . ⁢ where ⁢ I UX = ⁢ I U × cos ( ϕ I 2 ) = I L × cos ( ϕ I 2 ) , I UY = ⁢ I U × sin ( ϕ I 2 ) = I L × sin ( ϕ I 2 ) , Q UX = ⁢ Q U × sin ( ϕ Q 2 ) = Q L × sin ( ϕ Q 2 ) , ⁢ and ⁢ Q UY = ⁢ Q U × cos ( ϕ Q 2 ) = Q L × cos ( ϕ Q 2 ) . ( 6 )
In the example of FIG. 5, a frequency generator such as a synthesizer 510 generates a reference signal A*cos(ω)t) 511 having the same frequency as that of output signal r(t) 578. It can be understood by a person skilled in the art that the choice of the reference signal is made according to the desired output signal. For example, if the desired frequency of the desired output signal is 2.4 GHz, then the frequency of the reference signal is set to be 2.4 GHz. In this manner, embodiments of the invention achieve frequency up-conversion.
In the example embodiment of FIG. 5, each of vector modulators 520, 530, 540, 550 includes an input phase splitter 522, 532, 542, 552 for phasing the signals 522, 531, 541,551. Accordingly, input phase splitters 522, 532, 542, 552 are used to generate an in-phase and a quadrature components or their respective input signals.
Step 640 includes processing the Q component to generate third and fourth signals having the output signal frequency. The third and fourth signals have substantially constant and equal magnitude envelopes and a sum equal to the Q component. The third and fourth signals correspond to the QU(t) and L(t) constant envelope constituents described above. In the example of FIG. 5, step 630 is achieved by vector modulators 540 and 550, in conjunction with their appropriate input signals.
FIGS 9A and 9B conceptually illustrate the CPCP 2-Branch VPA embodiment using a phasor signal representation. In FIG. 9A, phasor {right arrow over (Rin)} represents a time-varying complex envelope input signal r(t). At any instant of time, {right arrow over (Rin)} reflects a magnitude and a phase shift angle of signal r(t). In the example shown in FIG. 9A, {right arrow over (Rin)} is characterized by a magnitude R and a phase shift angle θ. As described above, the phase shift angle is measured relative to a reference signal.
Still referring to FIG. 9A, it is noted that, at any time instant, {right arrow over (R′)} can be obtained by the sum of an upper phasor {right arrow over (U′)} and a lower phasor {right arrow over (L′)}. Further, {right arrow over (U′)} and {right arrow over (L′)} can be maintained to have substantially constant magnitude. The phasors, {right arrow over (U′)} and {right arrow over (L′)}, accordingly, represent two substantially constant envelope signals r′(t) can thus be obtained, at any time instant, by the sum of two substantially constant envelope signals that correspond to phasors {right arrow over (U′)} and {right arrow over (L′)}.
U′(t)=A×cos(ωt+φ/2);
L′(t)=A×cos(ωt−φ/2); (9)
where C denotes the real part component of phasors {right arrow over (U′)} and {right arrow over (L′)} and is equal to A×cos
ϕ ( 2 )
Note that C is a common component of {right arrow over (U′)} and {right arrow over (L′)}. α and β denote the imaginary part components of phasors {right arrow over (U′)} and {right arrow over (L′)}, respectively. α=β=A×sin(φ/2).
r ′ ⁡ ( t ) = 2 ⁢ C × cos ⁡ ( ω ⁢ ⁢ t ) = 2 ⁢ A × cos ⁢ ϕ ( 2 ) × cos ⁡ ( ω ⁢ ⁢ t ) .
Still referring to FIG. 10, Rclk signal 1016 is input, in parallel, into two vector modulators 1060 and 1062. Vector modulators 1060 and 1062 generate the U(t) and L(t) substantially constant envelope constituents, respectively, of the desired output signal r(t) as described in (12). In vector modulator 1060, an in-phase Rclk signal 1020, multiplied with Common signal 1028, is combined with a 90° degree shifted version 1018 of Rclk signal, multiplied with first signal 1026. In parallel, in vector modulator 1062, an in-phase Rclk signal 1022, multiplied with Common signal 1028, is combined with a 90° degrees shifted version 1024 of Rclk signal, multiplied with second signal 1030. Common signal 1028, first signal 1026, and second signal 1030 correspond, respectively, to the real part C and the imaginary parts a and β described in equation (12).
As described above, signals 1040 and 1042 are characterized by having substantially equal and constant magnitude envelopes.
Accordingly, when signals 1040 and 1042 are input into corresponding power amplifiers (PA) 1044 and 1046, corresponding amplified signals 1048 and 1050 are substantially constant envelope signals.
Step 1130 includes processing the clock signal to generate a normalized clock signal having a phase shift angle according to the received I and Q components. In an embodiment, the normalized clock signal is a constant envelope signal having a phase shift angle according to a ratio of the I and Q components. The phase shift angle of the normalized clock is relative to the original clock signal. In the example of FIG. 10, step 1130 is achieved by multiplying clock signal 1010 's in-phase and quadrature components with Iclk_phase 1012 and Qclk_phase 1014 signals, and then summing the multiplied signal to generate Rclk signal 1016.
Still referring to FIG. 12, information signals 1222 and 1224 include normalized in-phase Iclk_phase and quadrature Qclk_phase signals, respectively. Iclk_phase and Qclk_phase are normalized versions of the I and Q information signals included in signal 1210. In an embodiment, Iclk_phase and Qclk_phase are normalized such that that (I2clk_phase+Q2 clk_phase=constant). It is noted that the phase of signal 1250 corresponds to the phase of the desired output signal and is created from Iclk_phase and Qclk_phase. Referring to FIG. 9B, Iclk_phase and Qclk_phase are related to I and Q as follows:
θ = tan - 1 ( Q I ) = tan - 1 ( Q clk_phase I clk_phase ) ( 12.1 )
As described and verified above with respect to FIG. 9A