Source: https://patents.google.com/patent/US7481111?oq=6233389
Timestamp: 2018-03-17 21:10:17
Document Index: 362846343

Matched Legal Cases: ['art 2', 'arts 2', 'art 108', 'art 108', 'arts 107', 'art 108', 'art 108']

US7481111B2 - Micro-electro-mechanical sensor with force feedback loop - Google Patents
US7481111B2
US7481111B2 US11843579 US84357907A US7481111B2 US 7481111 B2 US7481111 B2 US 7481111B2 US 11843579 US11843579 US 11843579 US 84357907 A US84357907 A US 84357907A US 7481111 B2 US7481111 B2 US 7481111B2
US11843579
US20070289382A1 (en )
Each part 2 a, 2 b comprises a respective actuation system 5, a respective inertial sensor 6, and a mechanical connection 7, which connects the actuation system 5 to the inertial sensor 6. In FIG. 3, the microstructure 102 has been sketched in a simplified way with reference to just one of the two parts 2 a, 2 b.
In detail, the actuation system 5 comprises the driving mass 107 having an open concave shape (C shape), movable actuation electrodes 11 connected to the driving mass 107, and first and second fixed actuation electrodes 13 a, 13 b, comb-fingered to the movable actuation electrodes 11. The driving mass 107 is supported by first and second anchorages 15 a, 15 b via two first and two second anchoring springs 16 a, 16 b connected to the driving mass 107 in the proximity of the outer edges of the driving mass 107 itself.
In detail, the first insulating region 23 extends approximately parallel to the central axis of symmetry A so that the first part 108 a of the sensing mass 108 is supported by and connected to the driving mass 107 only via the first coupling springs 25 a, whilst the second part 108 b of the sensing mass 108 is supported by and connected to the driving mass 107 only via the second coupling springs 25 b.
Furthermore, the second insulating regions 24 extend transversely to the respective C-shaped arms so that the main portion 107 a of the driving mass 107 is connected only to the first coupling springs 25 a and to the first anchoring springs 16 a, whereas the end parts 107 b of the driving mass 107 is connected only to the second coupling springs 25 b and to the second anchoring springs 16 b. The position of the second insulating regions 24 is moreover such that the movable actuation electrodes 11 extend from the main portion 107 a of the driving mass 107 and are electrically connected thereto.
Actuation biasing regions 27, of a buried type, are connected to the first anchoring regions 15 a; first detection biasing regions 28, which are also of a buried type, are connected to the second anchoring regions 15 b; second detection biasing regions 29 are connected to the first fixed sensing electrodes 22 a; and third detection biasing regions 30 are connected to the second fixed sensing electrodes 22 b.
In this way, the first part 108 a of the sensing mass 108, the first coupling springs 25 a, the main portion 107 a of the driving mass 107, the movable actuation electrodes 11, the first anchoring springs 16 a, and the first anchoring regions 15 a are all set at one and the same potential, applied via the actuation biasing regions 27, and are electrically insulated, by the insulating regions 23, 24, from the rest of the suspended structures, which include the second part 108 b of the sensing mass 108, the second coupling springs 25 b, the end portions 107 b of the driving mass 107, the second anchoring springs 16 b and the second anchoring regions 15 b, biased via the first detection biasing regions 28.
The transimpedence amplifier 109 is of a fully differential type and has a pair of inputs connected to reading outputs 107 c, 107 d of the actuation system 5 for receiving first reading currents IRD1, IRD2, which are correlated to the linear velocity of oscillation of the driving mass 107 along the first axis X. On the outputs of the transimpedence amplifier 109 there are hence present first reading voltages VRD1, VRD2, which also indicate the linear velocity of oscillation of the driving mass 107 along the first axis X. Also the first reading voltages VRD1, VRD2 have equal amplitude and frequency and are 180° out of phase.
T ⁡ ( s ) = K ⁢ ⁢ s 1 + sT P
where s is a complex variable, K is a constant coefficient and Tp=1/ωP is the time constant associated to the pole of the differentiator stage 110 (see also FIG. 6). Preferably, the coefficient K is such that, for frequencies higher than the frequency ωP of the pole, the gain K/Tp of the differentiator stage 110 is greater than unity. In practice, then, the differentiator stage 110 amplifies the harmonic components of the first reading voltages VRD1, VRD2 close to the resonance frequency ωR, whereas possible constant components are eliminated (for example offset voltages). Furthermore, in the bandpass B of the differentiator 110, i.e., for frequencies greater than the frequency ωP of the pole, the offset introduced by the differentiator stage 110 is substantially zero, since the contributions of the pole and of the zero compensate one another.
The PLL circuit 113 has inputs connected to the outputs of the differentiator stage 110 through a comparator 116, of an analog type with hysteresis, and an output 113 a, connected to a clock input 112 c of the controller 112. The comparator 116 supplies at output to the PLL circuit 113 the first clock signal CK, which is a square-wave voltage having a first value in a first half-period, in which the voltages on the outputs of the differentiator stage 110 have a respective sign, and a second value in a second half-period, in which the voltages on the outputs of the differentiator stage 110 have a sign opposite to the one corresponding to the first half-period. In practice, the first clock signal switches at each change of sign of the first reading voltages VRD1 VRD2, which are in phase with the voltages on the outputs of the differentiator stage 110. Hysteresis prevents multiple switching due to noise in the proximity of the changes of sign of the voltages on the outputs of the differentiator stage 110. On the output 113 a, moreover, the PLL circuit 113 supplies the second clock signal CK90. In particular, (see FIGS. 7 and 8) the first clock signal CK has edges synchronized with zero-crossing instants of the first reading voltages VRD1, VRD2 (one of which is represented with a dashed line). The second clock signal CK90 is 90° out of phase with respect to the first clock signal CK and is in phase with the peak values of the first reading voltages VRD1, VRD2. In other words, the first clock signal CK and the second clock signal CK90 are in phase, respectively, with the linear velocity and with the displacement of the driving mass 107 along the first axis X.
The differentiator stage 110 amplifies the first reading voltages VRD1, VRD2 and eliminates any possible intrinsic offset of the microstructure 102 or any offset introduced by the transimpedence amplifier 109. The elimination of the offset is particularly important for correct operation of the PLL circuit 113 and, consequently, of the controller 112. As illustrated in FIG. 9, when an offset OS is present, the first reading voltages VRD1, VRD2 do not change sign at each half-period, but at different instants. Consequently, the comparator 116 switches at instants in which the phase of the first reading voltages VRD1, VRD2 is not known and phase-locking fails. Thus, the first and second clock signals CK, CK90 do not contain useful information because their edges do not correspond to the changes in sign or to the peak values of the first reading voltages VRD1, VRD2. Instead, the differentiator stage 110 suppresses the offset, and hence the comparator 116 switches at instants significant for phase-locking. For this reason, the first and second clock signals CK, CK90 are synchronized to the first reading voltages VRD1, VRD2 with a zero and 90° phase lag, respectively. Also the controller 112 is hence correctly clocked. The use of the differentiator stage 110 is additionally advantageous because it enables amplification of the first reading voltages VRD1, VRD2, without introducing any phase lag around the resonance frequency ωR of the microstructure 102.
The charge amplifier 120 has inputs connected to the terminals 108 c, 108 d of the sensing mass 108 for receiving second reading currents IRS1, IRS2, which are correlated to the linear velocity of oscillation of the sensing mass 108 along the second axis Y. On account of the charge amplification, on the outputs of the charge amplifier 120 second reading voltages VRS1, VRS2 are present, which indicate the displacement of the sensing mass 108 along the second axis Y; also the second reading voltages VRS1, VRS2 have equal magnitude and frequency and are 180° out of phase with respect to one another.
The raw signal SRAW is generated within the sensing feedback loop 106 and is correlated to the displacements of the sensing mass 108 along the second axis Y. Furthermore, the raw signal SRAW is amplitude-modulated in the DSB-SC (Double Side Band-Suppressed Carrier) mode and is the sum of two components. A first component, useful for measurement of the instantaneous angular velocity, is in phase with the displacement of the sensing mass 108 and has an amplitude correlated to the Coriolis acceleration (along the second axis Y), to which the sensing mass 108 itself is subjected on account of the oscillation along the first axis X and of the rotation of the microstructure 102. A second component, 90° out of phase, is correlated to the spurious drawing motions. For example, if the driving mass 107 oscillates in a direction which is not perfectly aligned to the first axis X, the sensing mass 108 can be driven in oscillation along the second axis Y even in the absence of rotation of the microstructure 102.
second filtering means for filtering the raw signal to generate the feedback voltages and impose stability conditions on a phase of the gain of the analog force feedback loop.
US11843579 2004-08-03 2007-08-22 Micro-electro-mechanical sensor with force feedback loop Active US7481111B2 (en)
EP20040425600 EP1624286B1 (en) 2004-08-03 2004-08-03 Micro-electro-mechanical sensor with force feedback loop
EP04425600.6 2004-08-03
US11195363 US7275433B2 (en) 2004-08-03 2005-08-02 Micro-electro-mechanical sensor with force feedback loop
US11843579 US7481111B2 (en) 2004-08-03 2007-08-22 Micro-electro-mechanical sensor with force feedback loop
US20070289382A1 true US20070289382A1 (en) 2007-12-20
US7481111B2 true US7481111B2 (en) 2009-01-27
US11195363 Active US7275433B2 (en) 2004-08-03 2005-08-02 Micro-electro-mechanical sensor with force feedback loop
US11843579 Active US7481111B2 (en) 2004-08-03 2007-08-22 Micro-electro-mechanical sensor with force feedback loop
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