Source: http://www.google.com/patents/US5568088?dq=6233389
Timestamp: 2017-11-20 15:51:44
Document Index: 471671680

Matched Legal Cases: ['application No. 08', 'application No. 08', 'application No. 08', 'application No. 08', 'application No. 08', 'application No. 08', 'application No. 08']

Patent US5568088 - Waste energy control and management in power amplifier - Google Patents
An amplifying apparatus for linearly amplifying a desired signal using a pair of coupled non-linear amplifiers is disclosed. The amplifying apparatus comprises a limiter for separating amplitude variations from the desired signal and producing a constant amplitude signal bearing the phase of the desired...http://www.google.com/patents/US5568088?utm_source=gb-gplus-sharePatent US5568088 - Waste energy control and management in power amplifier
Publication number US5568088 A
Application number US 08/472,247
Also published as CA2157183A1, DE19580085T0, DE19580085T1, US5574967, US5631604, US5638024, US5732325, US5771444, US5818298, US5842140, WO1995019066A1
Publication number 08472247, 472247, US 5568088 A, US 5568088A, US-A-5568088, US5568088 A, US5568088A
Inventors Paul W. Dent, Ross W. Lampe
Patent Citations (19), Referenced by (103), Classifications (33), Legal Events (4)
US 5568088 A
1. A device for increasing the energy efficiency of an amplifier using feedforward linearization, comprising:
a direct current power source to power said amplifier;
a first non-linear amplifier for producing a main output signal;
a second amplifier for amplifying an error signal;
combining means for combining said error signal with said main signal to produce a corrected sum signal and a waste energy signal; and
rectifier means for converting said waste energy signal to a direct current that is supplied back to said direct current power source to reduce net power consumption.
A transmitter power amplifier can also be constructed by combining two similar, smaller sized amplifiers. If the amplifier devices are driven in antiphase and their outputs are combined with a 180 degree relative phase so that their outputs add constructively, the amplifier is known as a push-pull amplifier. Sometimes, two similar amplifiers 20 and 21 can be driven 90 degrees out of phase and their outputs combined using a 90 degree or quadrature coupler as illustrated in FIG. 2. The quadrature coupler 23 can be formed by running two strip transmission lines in parallel proximity to each other. The energy is transferred between the lines in such a manner that a signal flowing from left to right on one line induces a signal flowing from right to left on the other line but with a 90 degree phase shift. Thus, two amplifiers connected respectively to the left hand end of a first line and the right hand end of a second line win produce signals travelling from left to right on the first line and from right to left on the second line.
FIGS. 1(a) and 1(b) illustrate a conventional power amplifier coupling to an antenna;
An overdimensioned matrix power amplifier according to one embodiment of the present invention is illustrated in FIG. 4. A set of N input signals is connected to the input ports of an M+M port butler matrix 40 chosen according to Babcock or other optimum spacing. To apply Babcock spacing, the butler matrix ports are numbered with increasing integers according to the phase increments with which successive input signals appear combined at the outputs. For example, port 0 refers to the butler matrix output corresponding to the sum of the input signals with no phase shift. Port 1 refers to the output corresponding to the sum of input signals with an increasing phase shift in the series 0, d Φ, 2d Φ, 3d Φ etc. Port 2 corresponds to phase shifts 0, 2d Φ, 4d Φ, 6d Φ. . . etc. The M outputs of the butler matrix 40 are amplified in non-linear power amplifiers 42. Amplifier outputs are combined using a similar M+M port butler matrix 44 and the N Babcock-spaced output ports yield desired amplified signals, while the remaining M-N signals are terminated to dissipate unwanted intermodulation as heat or energy in dummy loads 46.
The amplifier 21 is of a known type and produces a hard-limited output signal which preserves phase information but has amplitude variations removed. The input signal's amplitude variations are encoded by the logarithmic amplifier's progressive detection process into a signal proportional to the logarithm of the amplitude which, after optional low-pass filtering in low pass filter 60, is applied to a function generator 62. The function generator 62 converts the time-varying logamplitude signal into two time varying signals labelled COSΘ and SINΘ to indicate that the sum of their squares is always unity. COSΘ is equal to the ratio of instantaneous amplitude to peak amplitude, or, expressed in terms of logamplitudes, is the antilog of the difference between instantaneous logamplitude and peak logamplitude. The peak logamplitude can either be determined by the function generator 62 over a sufficiently long period, or set from outside by means of a "SCALE" input. The SINΘ function is merely 1-COS2 Θ. One means of implementing such a function generator is by digitizing LOG(A) and then using digital signal processing circuitry comprising function look-up tables. Analog means can also be used however, employing diode or transistor networks that synthesize a piece-wise linear approximation to the desired functions.
In the digital implementation, the LOGAMPLITUDE signal is sampled and digitized at a rate at least equal to the total signal bandwidth. The SCALE signal is set to the peak of the LOGAMPLITUDE signal and subtracted from it to produce a value equal to the LOG of the ratio of instantaneous to peak amplitude. This value is always negative but may of course be complemented to produce a digital value that is always positive. This binary value between 000 . . . 000 and 111 . . . 111 is then used as the address to a pre-computed table of corresponding cos(Θ) and sin(Θ) values. The values are precomputed using the formula Θ=2ARCOSINE[EXP(-λA)] where A is the address and λ is a suitable scaring value depending on the number of bits of the address and their significance is to make the resultant Θ value equal to twice the ARCOSINE of the instantaneous to peak amplitude ratio. The digital cos(Θ) and sin(Θ) values are then converted to analog voltage waveforms using D to A convertors and low-pass filters.
The COS(Θ and SINΘ functions are used to multiply the constant amplitude signal supplied by the logarithmic amplifier 50, the low-pass filter 51 and the upconvertor 52 if used. The multiplier 58 effectively re-applies the amplitude modulation removed by the hard limiting amplifier 50 on the upper channel while applying a complementary amplitude modulation to the lower channel such that the sum of the squares of the new amplitudes A1 and A2 is unity. This arises through choice of the functions COSΘ and SINΘ whose sum square is automatically unity. The new signals from the multipliers 58 are A1·EXP(jΦ) and A2·EXP(jΦ) where Φ is the phase of the original input signal. These two signals are combined using a quadrature coupler 56 at the inputs of the class-C power amplifier to produce (A1+jA2)EXP(jΦ) and (A2+jA1)EXP(jΦ). Since the sum of the squares of the real and imaginary parts of these are by design unity, the class-C power amplifiers 57 receive constant amplitude drive signals so as to operate at maximum efficiency. The entire chain of components described up to the quadrature coupler 56 is suitable for integration on a small, low-cost silicon chip with a size of only 3 mm×3 mm.
N input signals are combined in an N+N port input Butler matrix 70 to produce N output signals. These are split into DRIVE and COMPLEMENTARY DRIVE signals in drive signal splitters 71 according to the above described procedure so that each signal is a constant amplitude signal. These constant amplitude signals may be amplified efficiently by the N pairs of class-C power amplifiers 72, whose outputs are then pair-wise combined in combiners 73 to produce N wanted signals and N waste energy signals. The waste energy signals are dissipated in dummy loads 74 as heat, while the N wanted signals are alecombined in output Butler matrix 75 to produce the original N input signals at an amplified power level. As a result, unwanted intermodulation products are dissipated as heat, light, or other electromagnetic radiation in dummy loads 74.
This embodiment of the present invention comprises a multi-beam satellite antenna for dividing the area illuminated on the earth into cells. A matrix power amplifier is provided with an output corresponding to each antenna beam and comprised of a plurality of power amplifiers connected by means of Butler matrices at their outputs to provide each beam signal. Means are also provided to TDMA-modulate to vary the power level of the drive signals on a timeslot by timeslot basis. Means are also provided to inhibit transmission on groups of unused timeslots on any TDMA signal such that available power amplifier power can during that period be taken up by increasing the drive signals in another beam. The TDMA modulation and power varying means are preferably located on the ground. A ground station comprises the necessary signal generation for each beam signal and transmits these to the satellite. Moreover, the matrix-PA input Butler matrix combiners can also preferably be located on the ground whereby the ground station produces already combined drive signals for directly driving each of the power amplifiers and these signals are transmitted to the satellite by means of coherent feeder links as disclosed in U.S. patent application No. 08/179,953, entitled "A Cellular/Satellite Communications System With Improved Frequency Re-use", filed Jan. 11, 1994, which is expressly incorporated by reference.
A completely genera/passive combining structure 80 combines the outputs of 2N power amplifier stages 82 to produce N wanted signals and N unwanted signals that are dissipated in dummy loads 81 as heat or light. In a satellite application, the latter can be achieved by use of incandescent filament lamps as the dummy loads, and the resultant light focused back on the solar cell array. The passive combining network can be the known prior an Butler matrix, or a reduced Butler matrix formed by omitting combining waste energy signals, or a simplified Butler matrix corresponding to a Fast Walsh Transform structure as opposed to the usual FFT structure.
S(k)--S(k-1)=jGAIN·C·T·dT
where C is shorthand notation for the coefficient matrix, and T is the diagonal matrix ##EQU2## and dT is a vector of the 2N phase changes to be found. By denoting the product jGAIN·C·T by the N by 2N complex matfix U, and the difference between successive signal samples S(k)--S(k-1) as dS, the following equation is obtained:
The N by 2N complex matrix U may be regarded as a 2N by 2N real matrix if the N×2N real coefficients URij are row-interleaved with the N×2N imaginary parts Ulij in the following pattern: ##EQU3## Likewise, the N complex dS values comprise 2N real values and can be regarded as a 2N point real vector by interlacing real and imaginary parts vertically. Thus, the N complex equations U·dT=dS are actually 2N real equations that are simply solved by a real equation solving process to yield
The dT values so found are added to corresponding Θ values to obtain new Θ values. These can be used to calculate new values of EXP(jΘ) which are the required drive signals for the power amplifiers. This conversion of Θ values to drive signals can take place by simply applying digitized Θ values to a COS/SIN look-up table or ROM to obtain values for COS(Θ) and SIN(Θ) which are the real and imaginary parts of EXP(jΘ). The numerical values of COS(Θ) and SIN(Θ) are then DtoA converted and used to drive a quadrature modulator to produce the radio-frequency drive signals as shown in FIG. 8, which will be explained in detail below. Since modem technology is capable of realizing DtoA convertors of adequate precision (e.g. 8 bits) which can run at 1000 Megasamples per second, such a digital implementation is practically useable for all practical bandwidths.
The new Θ values and the C matrix can be used to calculate the S-values actually achieved by the linearizing approximation of differentiation. Alternatively, the S-values achieved may be measured at the wanted signal outputs. Either way, it is the actually achieved values of S that are used as S(k-1) to form the differences dS with the next desired set of samples S(k). In this way, errors do not propagate and are limited. In the case where measured S values are subtracted from the next set of desired values S(k), the system can be described as N-channel Cartesian feedback. Cartesian feedback is a known technique for reducing distortion in a quasi linear power amplifier through using a signal assessment demodulator to measure the achieved complex values of a signal at the output of a power amplifier and comparing them with desired values to produce error values. The error values are integrated and fed to a quadrature modulator to produce new values of the power amplifier drive signals that will cause the power amplifier more nearly to deliver the desired complex signal output. An advantageous method for the above is described in U.S. patent application No. 08/068,087, entitled "Selfadjusting Modulator", which is incorporated herein by reference. In the case of the above inventive matrix power amplifier, feeding back measured complex values of the N output signals so as better to achieve the next set of complex values is a form of Cartesian feedback for correcting signal matrices instead of single complex signal values.
By determining the new dS values in the above-described manner, the new Θ values are used with the C matrix to determine the new U matrix. A simplified way to perform this function is to note that the effect of adding DΘ1 to DΘ2 will be to rotate the previous values of U1j through an angle DΘ1, which causes a transfer of a fraction of their imaginary parts to their real parts and vice versa. For small values of DΘ, this allows the U matrix to be updated with no multiplies. The DΘ values can always be kept small by choosing successive sample values S(k-1), S(k), S(k+1) . . . to be sufficiently closely spaced in time. A high speed digital logic machine or computer may be envisaged for performing these calculations in real time to give continuous drive waveforms to the power amplifier stages of several MHz bandwidth. For a satellite application, such a machine is preferably located on the ground and the resulting drive waveforms only communicated to the satellite via 2N mutually coherent feeder links as disclosed in U.S. patent application No. 08/119,953, entitled "A Cellular/Satellite Communications System With Improved Frequency Re-use", filed Jan. 11, 1994, which is expressly incorporated herein by reference.
A simplified alternative exists when the signals to be communicated Si are radio signals modulated with digital information streams. If the information streams on each of the signal paths are symbol or bit synchronous, then the waveforms Si at any instant depend on a limited number of past and future bits according to the smearing produced by the impulse response of the premodulation filtering. In the limit, an S-value in the center of a symbol at least may depend only on that symbol. With binary signalling, the symbol can only take on one of two values, 0 or 1, corresponding to an Si of +1 or -1 with suitable scaling to the desired output power and frequency. For a matrix power amplifier of limited size, for example 16 channels, this means that there can only be 216 =65536 different vectors S when sampled mid-bit. The 2N values of Θ corresponding to each of these S vectors can be precomputed and stored in a reasonably sized ROM, and may be retrieveel when needed by applying the current 16-bits from the 16 channels as an address. For practical purposes, adequate shaping of the data transitions from one bit period to the next in order to control the spectrum may be obtainable by smoothly transitioning between these sets of Θ values by means of interpolation. Thus, if a particular Θ value for a current set of 16 bits being transmitted was 130 degrees and a next value was -170 degrees, the Θ value would be moved from the old value to the new by means of the sequence 135, 140, 145, 150, 155, 160, 165, 170, 175, 180, -175, -170 noting that the shortest path is taken. A Θ value that had less far to move would take the same number of smaller steps. At each step, the values of THETA would be applied via COS/SIN ROMs and DtoA convertors to a quadrature modulator in order to convert them to the desired radio-frequency drive signals for the power amplifiers.
In another embodiment of the present invention, the amplifier outputs are not combined before being fed to a multi-beam antenna but rather are fed directly to the elements of an antenna array. FIG. 9 illustrates the use of a cylindrical array of slot antennas as might be used for a phased array base station as disclosed in U.S. patent application No. 08/179,053, entitled "A Cellular/Satellite Communications System With Improved Frequency Reuse", which is incorporated herein by reference above. In such a phased array, a number of antenna elements, such as 8 horizontal columns around the cylinder and 20 vertical columns, are used for transmission or reception of signals from mobile radio telephones. For transmission, the elements in each of the vertical columns may be driven in phase from the output of a single power amplifier by use of a 20-way, passive power-splitter. Alternatively, each element may be equipped with its own associated smaller power amplifier (of 1/20th the output power) and the 20 amplifiers in each column are driven in phase using the same drive signal.
When either of the above arrangements is used for transmitting multiple signals each in a different desired direction, the signal for each column comprises the sum of the different signals with an appropriate set of amplitude and phase (complex) weightings for that column. For example, the signals S1, S2, S3 . . . for the eight columns may be formed from the signals T1, T2, T3 to be transmitted as follows: ##EQU4## etc., where j=.check mark.(-1) signifying a 90-degree phase-changed component.
It can be seen that the signal Si to be amplified for application to a column of elements comprises the complex-weighted sum of the independent signals T1, T2 etc. Thus, the amplifier or amplifiers for that column have to faithfully reproduce not just a single signal (that could have been a constant amplitude signal as with analog FM) but the sum of independent signals that is not a constant amplitude signal. Thus, in prior art phased array base stations, the use of linear amplifiers that reproduced both the amplitude and phase variations of the composite drive signals is required. In U.S. Pat. No. 3,917,998 to Welti, the use of a coupled matrix of power amplifiers is disclosed with the property that no single amplifier necessarily has to generate the peak power that any one antenna element or feed may require; however the possibility of using constant-amplitude power amplifiers was not disclosed. In a previous embodiment of the present invention, the possibility of using class-C or constant amplitude power amplifiers in a coupled matrix is disclosed, through overdimensioning the matrix by a factor of two at least relative to the number of independently-specifiable amplified output signals desired. The previous embodiment consisted of a number at least 2N of class-C power amplifiers coupled at their outputs by an at least 2N+2N port Butler matrix or lossless coupling network, wherein the 2N input ports being connected to said amplifier outputs and N of said coupling network output ports being said desired amplified signal outputs, the rest being terminated in dummy loads. As a result, the unwanted intermodulation products are dissipated in the dummy loads as heat.
In the present embodiment of the present invention, the 2N+2N Butler matrix is not needed. Instead, each column of elements in the array is split into even and odd elements numbered vertically from 1 at the top to at least 2N at the bottom. In one implementation of this embodiment, elements 1, 3, 5 etc. of a column are connected by a passive N-way power splitter driven by a first power amplifier, while elements 2, 4, 6, 8 . . . are connected by a second N-way power splitter driven by a second power amplifier. In a second implementation of the present embodiment, each element is equipped with a smaller power amplifier and the amplifiers attached to even elements are all driven with a ftrst drive signal while amplifiers attached to odd elements are all driven with a second drive signal. If the drive signals are the same, radiation from the even and odd elements will reinforce in the horizontal plane, while if the drive signals are in antiphase, there will be no radiation in the horizontal plane. Thus it is possible by varying the relative phasing between in phase and antiphase to produce varying amplitudes of signal radiation in the horizontal plane even though the individual elements of the column receive constant amplitude signals from their respective power amplifiers.
D1=Ao·EXP[j(Φ+Θ)]
D2=Ao·EXP[j(Φ-Θ)]
COS(Θ)=A(t)/2Ao
It may be verified that, when D1 and D2 are added to calculate the radiation in the horizontal plane, we obtain ##EQU5##
The principle of the present embodiment of the present invention may be extended by dividing a column of elements into four equal groups, i.e., nos. 1, 5, 9 . . . 2, 6, 10 . . . 3, 7, 11 . . . 4, 8, 12 and so on. Each group is connected to a single power amplifier for that group by means of a power splitter, or each element is equipped with its own power amplifier and the amplifiers of a group are driven in phase with a drive signal adapted for each group. Then, a set of four drive signals is found such that desired signal radiation occurs at up to two desired elevation angles, which may if desired both be in the horizontal plane. In general, the problem of how to produce radiation of N desired signals at N desired angles of elevation from a column of at least 2N constant-amplitude radiating elements may be solved by a similar mathematical process to that described above for generating N desired signals of varying amplitude and phase using at least 2N signals of constant amplitude and varying phase.
The amplitude and phase relationship of the array elements for reception can be predicted from the theoretical or measured polar patterns of the constituent elements and their physical disposition in the army. Thus, the relative phases and amplitudes for transmission in any direction can be predetermined by changing the signs of the phases. It is often sufficient to quanfize the number of possible directions the array can be called upon to transmit in to a limited number of beams spaced, for example, at 5 degree intervals around 360 degrees of azimuth. The phase relationships and amplitude ratios can thus be precomputed for each of these 72 directions. If the array is formed by 8 columns of elements, there exists furthermore an 8-fold symmetry such that said 72 possible phase and amplitude relationships reduce to only 9 distinct patterns that are repeated by shifting the whole pattern in steps of one column around the array. It is thus a relatively straightforward process to store these 9 patterns in a signal processor and to select the pattern and the shift needed to radiate a given signal in a given direction to the nearest 5 degrees. When this has been done for all signals desired to be radiated, and the results summed to determine the composite radiation to be produced by each column of elements, the drive signals for the constant-amplitude amplifiers associated with each column are generated using the method described above.
FIG. 10 illustrates a block diagram of a transmit signal processor designed to accomplish the above-described process. A set of signals to be transmitted T1, T2 . . . Tn1 along with associated direction information Θ1, Θ2 . . . is applied to transmit matrix processor 110. Signals T1, T2 etc. are preferably in the form of a digitized sample stream produced by other signal processor(s) (not shown) that can include speech coding, error correction coding and digital conversion to modulated radio signal form. The latter represents each signal as a stream of complex numerical samples having a real and an imaginary part or alternatively in polar form using a phase and amplitude. The matrix processor 110 uses the direction information Θ1, Θ2 etc. to select or compute a set of complex weighting coefficients using stored data in a memory 112 which is adapted to the array configuration. In its simplest form, direction information Θ1, Θ2 etc. can consist of a beam number, and the beam number is used to select a set of precomputed stored coefficients from the memory 112. The coefficients are used in complex multiplication with signals T1, T2 to produce weighted sums S1, S2 . . . etc.
The complex weighting coefficients are calculated for each signal according to its desired direction of radiation and using the principle of phase-conjugating the signals that would be received from the desired direction, as disclosed above. The weighting coefficients may also be calculated such as to minimize the transmit power consumed to produce a given signal strength at the receivers, as disclosed in the incorporated U.S. patent application No. 08/179,953, entified "A Cellular/Satellite Communications System With Improved Frequency Re-use".
The composite output signals now represent the sum of signals at different frequencies and thus are wide band signals with a commensurately increased sample rate. Each signal Si is then convened in drive signal splitters 114 to a pair of constant amplitude signals whose sum has the desired instantaneous phase and amplitude of Si. This process of drive splitting is preferably still performed in the digital signal domain, but shortly thereafter it is appropriate to convert said drive signals to analog form with the aid of DtoA convertors. Since the numerical form consists of complex numbers, one convertor may be used for the real part and another for the imaginary part. The two signals produced are termed in the known art as I,Q signals and may be applied to a known quadrature modulator device to translate them to a desired radio frequency band. Translated signals are then amplified to a transmit power level using efficient, constant-amplitude power amplifiers 116 and 118 for even and odd column elements respectively. These amplifiers may be distributed among the array elements themselves.
It will be appreciated that the present invention may be applied similarly to arrays of elements disposed on a cylindrical surface, a planar surface, or any other surface. The general principle is to provide a superfluity of elements of preferably at least a factor of two over and above the number of distinctly different signal directions the array is desired to resolve. In this way, the array when transming can comprise efficient constant-amplitude power amplifiers and unwanted signal intermodulation products that arise can be arranged to be radiated in directions other than those in which the array radiates wanted signal energy. For example, an orbiting satellite carrying such an array can be arranged to irradiate the earth with wanted signals at different locations while intermodulation products (unwanted signals) are radiated harmlessly into space. In this application, the invention disclosed in the incorporated U.S. patent application No. 08/179,953, entitled "A Cellular/Satellite Communications System With Improved Frequency Re-use", may be employed to place the generation of the drive signals for said array elements and associated amplifiers on the ground, the resulting signals being conveyed to the satellite using coherent feeder links from a ground-based hub-station. An advantage of the present invention is a reduction of the heat that would normally have to be dissipated in a less efficient, linear power amplifier designed not to produce such unwanted intermodulation signals.
The ability to radiate heat from an orbiting satellite can often be the dominant factor in limiting the capacity of a satellite communications system. Satellite communications systems designed to provide communication with a large number of mobile stations are of so-called multiple access type and may employ Frequency Division Multiple Access, Time-Division Multiple Access, Code Division Multiple Access, or any combination of these techniques. In FDMA or CDMA systems, a large number of signals must be radiated simultaneously leading to the problem of intermodulation in transmitters. In TDMA signals, a frame period is divided into timeslots, and each signal occupies a timeslot. Thus, in a pure TDMA system, it is not necessary to radiated many signals simultaneously, and efficient constant amplitude transmitters can be used. In practice, however, lack of available frequency spectrum requires that a fourth multiple access method also be employed, called Space Division Multiple Access (SDMA) or "Frequency Re-use". Frequency Re-use is the well-known cellular radio-telephone technique of dividing the earth into cells and permitting cells that are sufficiently separated to employ the same frequencies. Thus, even when a satellite system employs pure TDMA within each cell, it may have to employ SDMA to permit other cells to use the same frequency; thus the satellite ends up having to radiate several signals at the same time in different directions. The invention described above may be employed to allow a phased array of antenna elements with associated efficient, class-C power amplifiers to generate a number of TDMA signals for radiation instantaneously in different directions. The set of directions may change from one timeslot to the next as disclosed in U.S. patent application No. 08/179,953, entitled "A Cellular/Satellite Communications System With Improved Frequency Re-use". The overall efficiency of such an inventive arrangement expressed in terms of conversion of DC power from a solar array or battery into usefully radiated signal energy may be no greater than if a prior art arrangement using linear amplifiers had been employed. However, the inefficiency shows up less in terms of heat dissipation, and instead as the radiation of radio energy in the form of intermodulation products harmlessly into space. Alternatively, the intermodulation may be subject to the waste energy recovery procedure herein described.
Another embodiment of the present invention relates to recycling waste energy generated in a power amplifier. In FIG. 12, a power amplifier 140 is coupled through an isolator/circulator 141 and an optional transmit filter 142 to an antenna 143. In duplex systems requiring simultaneous transmission and reception on different frequencies, the transmit filter may form part of the antenna duplex filter, the other part of the filter 144 being associated with the receiver. In non-duplexing systems such as Time Division Duplex/Time Division Multiple Access systems or simply press-to-talk systems, there may be no duplex filter but rather a transmit/receive antenna switch. The present invention is applicable to both of these configurations. In the present invention, the dummy load normally connected to the reflected power port of a circulator is replaced by a rectifier 145. The rectifier 145 converts AC radio frequency energy into a direct current which is then fed back to the battery or power source that powers the transmitter. If the efficiency of the transmitter is E1 and the efficiency of the rectifier is E2, then a fraction E1·E2 of the total energy will be recovered in the case where the antenna is a complete mismatch, reflecting all the transmitted power fed to it. The fraction E1·E2 can of course never be greater than unity. In the case where the antenna reflects a fraction R of the power fed to it, the net consumption of energy will be reduced by the factor 1-R·ER1·E2. For example, if R=10%, E1=55%, and E2=70%, the battery consumption is reduced by 3.85%, which is a significant amount of savings. When the effective energy is calculated as the ratio of the power actually radiated from the antenna to the net power consumption, the efficiency curves with and without the use of the present invention are plotted in FIG. 13. FIG. 13 shows that the effective efficiency is less sensitive to antenna rnismatch when the present invention is used. FIG. 14 illustrates the percentage of increased talk-time by using the present invention, as a function of the percentage of energy reflected by the antenna.
Another embodiment of the present invention is illustrated in FIG. 15. Two similar power amplifier stages 150 and 152 are combined using a 3 db directional coupler 154 in order to obtain the sum of their output powers at the antenna 156. An unwanted difference signal is produced which, instead of being dissipated in a dummy load as in FIG. 2, is convened to a DC current in rectifier 158 and the current is fed back to the battery or power source feeding the amplifiers. The same curves illustrated in FIGS. 13 and 14 apply to this embodiment as well.
Without the present invention, the efficiency is given by: ##EQU6## where k is the voltage coupling factor of the directional coupler 164, Ec is the efficiency of the amplifier 160,
When using the present invention, the effective efficiency is given by: ##EQU7## where R is the rectifier efficiency and D is the difference signal given by
If the single carrier amplifier approach is selected, means must be provided to connect the plurality of amplifiers to the antenna. Multiple antennas are not favorable because they tend to be large, expensive and need more rear estate. One way to couple multiple amplifiers to the antenna is to use a multicoupling filter, which uses frequency selectivity to isolate the different amplifiers from one another. Multicoupling filters are large, and expensive and are only feasible when the frequency difference between the multiple carriers is not too small. The alternative is to use dissipative combining, in which amplifiers may be combined in pairs using hybrid couplers or directional couplers, which are essentially sum and difference networks. Using dissipative combining for two signals, for example, a power amplifier of power 2P is used for a first signal and a second power amplifier of output power 2P for a second signal. The outputs of the two power amplifiers are then combined using a 3 db coupler. The coupler allows half of the power of each power amplifier, i.e., P from each power amplifier, to reach the antenna while the other half is normally dissipated in a dummy load. In general, dissipative combining of N signals requires that each signal be amplified to a power N times the desired power P, so that the total power of all N amplifiers is N×NP, while only NP (P from each amplifier) reaches the antenna. Where N is a power of 2, the dissipative combiner is a binary tree which combines pairs of signals and then pairs of pairs and so on to the final output. Each pair-combining network may be a 3 dB coupler.
At each coupler, half of the power is extracted in a sum signal and passed to the next coupler or to the final output, while the other half is normally wasted in a dummy load. Thus, half the total power is wasted in combining two carriers, 3/4 of the total power is wasted in combining four carriers, and in genera/a fraction (N-1)/N of the total power is wasted when N carriers are combined. By employing the current invention, this normally wasted power is recovered at least to the extent of the rectifier efficiency R. Thus, instead of the net efficiency being only E/N where E is the efficiency of one single carrier power amplifier, the effective efficiency becomes E/(N-(N-1)E·R). For example, when E=60% and R=70%, the effective efficiency is plotted versus N in FIG. 17 with and without the present invention. The figure illustrates that efficiency can be improved by a factor of more than 1.5 by using the present invention.
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U.S. Classification 330/151, 330/124.00R, 330/295, 330/202, 455/127.5, 455/13.4, 330/297
International Classification H04B1/16, H04B7/185, H03F1/32, H03F3/60, H03F3/68, H03F1/02, H04B1/04, H04B1/52
Cooperative Classification H03F1/3252, H04B1/0483, H04B1/52, H04B1/04, H03F1/0294, H04B2001/0433, H03F1/3229, H04B1/1607, H03F3/602, H03F3/24, H03F1/34
European Classification H04B1/16A, H04B1/04P, H03F3/60C, H04B1/52, H03F1/32P4, H03F1/32F2, H03F1/02T6