Source: http://www.google.com/patents/US8130853?dq=5,371,548
Timestamp: 2017-03-29 17:58:00
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Matched Legal Cases: ['§119', 'Application No. 60', 'art 16', 'art 11', 'art 11', 'art 16', 'art 16', 'art 11', 'art 11', 'art 11', 'art 11']

Patent US8130853 - Method and apparatus for performing baseband equalization in symbol ... - Google PatentsSearch Images Maps Play YouTube News Gmail Drive More »Sign inPatentsA method and apparatus for processing a frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol. The method includes generating an estimate of a channel for each sub-carrier of the frequency domain digital OFDM symbol; generating a plurality of demodulated symbols based, at...http://www.google.com/patents/US8130853?utm_source=gb-gplus-sharePatent US8130853 - Method and apparatus for performing baseband equalization in symbol modulated communicationsAdvanced Patent SearchTry the new Google Patents, with machine-classified Google Scholar results, and Japanese and South Korean patents.Publication numberUS8130853 B1Publication typeGrantApplication numberUS 12/645,644Publication dateMar 6, 2012Filing dateDec 23, 2009Priority dateFeb 15, 2002Fee statusPaidAlso published asUS7133473, US7564913, US7643586, US8437417, US8675786, US9036743Publication number12645644, 645644, US 8130853 B1, US 8130853B1, US-B1-8130853, US8130853 B1, US8130853B1InventorsHui-Ling Lou, Kok-Wui CheongOriginal AssigneeMarvell International Ltd.Export CitationBiBTeX, EndNote, RefManPatent Citations (19), Non-Patent Citations (16), Referenced by (1), Classifications (10), Legal Events (1) External Links: USPTO, USPTO Assignment, EspacenetMethod and apparatus for performing baseband equalization in symbol modulated communications
US 8130853 B1Abstract
A method and apparatus for processing a frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol. The method includes generating an estimate of a channel for each sub-carrier of the frequency domain digital OFDM symbol; generating a plurality of demodulated symbols based, at least in part on, the estimate of the channel for each sub-carrier, in which each demodulated symbol corresponding to a given sub-carrier of the frequency domain digital OFDM symbol; calculating a decision metric for each sub-carrier based on (i) the channel estimate corresponding to the sub-carrier and (ii) the demodulated symbol corresponding to the sub-carrier; and estimating a most likely transmitted symbol for each sub-carrier of the frequency domain digital OFDM symbol based on the decision metric corresponding to the sub-carrier.
1. A receiver baseband processor comprising:
a channel estimator configured to generate an estimate of a channel for each sub-carrier of a frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol;
a demodulator configured to generate a plurality of demodulated symbols based, at least in part on, the estimate of the channel for each sub-carrier of the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol, each demodulated symbol corresponding to a given sub-carrier of the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol; and
a decoder including
a metric calculator configured to calculate a decision metric for each sub-carrier of the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol based, at least in part, on (i) the channel estimate corresponding to the sub-carrier and (ii) the demodulated symbol corresponding to the sub-carrier, and
a Maximum Likelihood Sequence Estimation unit configured to estimate a most likely transmitted symbol for each sub-carrier of the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol based on the decision metric corresponding to the sub-carrier.
2. The receiver baseband processor of claim 1, wherein the metric calculator calculates a branch metric (BM) for each sub-carrier of the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol in accordance with the following equation:
BM k,n,i=−2Z k,n,i ·|H k,n|2 ·X+(|H k,n|2 ·X)2
where i∈{0,1}, k corresponds to each sub-carrier index of an nth OFDM symbol, and X∈{±1} for QPSK, X∈{±1, ±3} for a 16-QAM constellation, and X∈{±1, ±3, ±5, ±7} for a 64-QAM constellation.
3. The receiver baseband processor of claim 1, wherein the metric calculator calculates a branch metric (BM) for each sub-carrier of the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol in accordance with the following equation:
BM k,n,i=−2Z k,n,i ·X+(|H k,n |·X)2
4. The receiver baseband processor of claim 1, wherein the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol is formatted in compliance with at least one of the following standards: IEEE 802.11a, IEEE 802.11g, or IEEE 802.16a.
5. The receiver baseband processor of claim 4, wherein the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol is mapped to one of: a DBPSK modulation constellation, a QPSK modulation constellation, a 16-QAM modulation constellation, a 64-QAM modulation constellation, a 256-QAM modulation constellation, or an N-QAM modulation constellation where N exceeds 256.
6. The receiver baseband processor of claim 1, further comprising a slicer configured to perform a hard-decision decoding on each demodulated symbol and generate a corresponding hard-decision decoded symbol.
7. A transceiver comprising the receiver baseband processor of claim 1.
8. A network apparatus comprising the transceiver of claim 7.
9. The network apparatus of claim 8, wherein the network apparatus comprises one or more of a PC card, a network interface card, or a wireless communications access point.
generating an estimate of a channel for each sub-carrier of a frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol;
generating a plurality of demodulated symbols based, at least in part on, the estimate of the channel for each sub-carrier of the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol, each demodulated symbol corresponding to a given sub-carrier of the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol;
calculating a decision metric for each sub-carrier of the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol based, at least in part, on (i) the channel estimate corresponding to the sub-carrier and (ii) the demodulated symbol corresponding to the sub-carrier; and
estimating a most likely transmitted symbol for each sub-carrier of the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol based on the decision metric corresponding to the sub-carrier.
11. The method of claim 10, wherein calculating a decision metric for each sub-carrier of the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol comprises:
calculating a branch metric (BM) for each sub-carrier of the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol in accordance with the following equation:
12. The method of claim 10, wherein calculating a decision metric for each sub-carrier of the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol comprises:
13. The method of claim 10, wherein the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol is formatted in compliance with at least one of the following standards: IEEE 802.11a, IEEE 802.11g, or IEEE 802.16a.
14. The method of claim 13, wherein the frequency domain digital Orthogonal Frequency Division Multiplexing (OFDM) symbol is mapped to one of: a DBPSK modulation constellation, a QPSK modulation constellation, a 16-QAM modulation constellation, a 64-QAM modulation constellation, a 256-QAM modulation constellation, or an N-QAM modulation constellation where N exceeds 256.
15. The method of claim 10, further comprising performing a hard-decision decoding on each demodulated symbol and generating a corresponding hard-decision decoded symbol.
This application is a continuation of U.S. patent application Ser. No. 11/645,881, filed Dec. 27, 2006, which is a continuation of U.S. patent application Ser. No. 11/590,708 (now U.S. Pat. No. 7,564,913), filed Oct. 31, 2006, which is a continuation of U.S. patent application Ser. No. 10/259,142 (now U.S. Pat. No. 7,133,473), filed Sep. 26, 2002, which claims priority benefit under 35 U.S.C. §119(e)(1) to U.S. Provisional Application No. 60/357,317, filed Feb. 15, 2002. The disclosures of the above applications are incorporated herein by reference.
The receiver portion of a wireless communications system compliant with the aforementioned 802.11a/802.11g and 802.16a standards generally includes an RF receiving unit to handle RF downconversion and filtering of received signals in one or more stages and a baseband processing unit to demodulate OFDM encoded symbols bearing the data of interest. FIG. 2 illustrates a known receiver baseband processing unit 200, a channel model 201 and a corresponding encoding transmitter baseband processor 260 to conceptually illustrate the relationship between a received OFDM symbol Yn in relation to its originally transmitted counterpart OFDM symbol Xn and the pre FEQ version {tilde over (X)}n, taking into account the channel impulse response h(t) and intervening noise v(t) (Additive White Gaussian Noise or AWGN in particular), assuming that the channel impulse response h(t) is shorter than the guard interval (using e.g. cyclic prefix). To ease understanding, conventional RF upconversion transmission, and downconversion interposing the baseband processing units 200 and 260, are omitted from FIG. 2. More specifically, the channel model 201 illustrates the following relationship:
Y k,n =g n H k,n ·X k,n +g n v k,n (0)
X _ k , n = Y k , n H _ k , n = H _ k , n * · Y k , n  H _ k , n  2 ( 1 ) where: 1) X k,n is the demodulated symbol of the kth sub-carrier during the nth OFDM symbol; 2) Yk,n is the received noisy symbol of the kth subcarrier during the nth OFDM symbol; and 3) H k,n is the channel estimate corresponding to the kth subcarrier during the nth OFDM symbol.
Z k,n = H* k,n ·Y k,n, (2)
P k,n =e −jΦ k,n ·Y k,n, (3)
H k,n =| H k,n |e jΦ k,n . (4)
The in-phase and quadrature demodulated symbols are the real and imaginary parts of
X _ k , n ( i . e . Re { X _ k , n } and Im { X _ k , n } ) respectively. The following discusses the changes required in the conventional soft decision Viterbi decoder (such as decoder 230 shown in FIG. 2) as well as the hard-decision symbol-by-symbol (such as the slicer 225 shown in FIG. 2).
Y k,n =H k,n ·X k,n +v k,n, (5)
H k,n=Hk,n (6)
{H k,n }≡H k,n,0 , {H k,n }≡H k,n,1. (7)
Z k,n =H* k,n ·Y k,n =|H k,n|2 X k,n +H* k,n v k,n (8)
{Z k,n }≡Z k,n,0 =|H k,n|2 X k,n,0 +H k,n,0 ·v k,n,0 +H k,n,1 ·v k,n,1 (9)
{Z k,n }≡Z k,n,1 =|H k,n|2 X k,n,1 +H k,n,0 ·v k,n,1 −H k,n,1 ·v k,n,0. (10)
BM k,n,i=(Z k,n,i −|H k,n|2 ·X)2 =Z k,n,i 2−2Z k,n,i ·|H k,n|2 ·X+(|H k,n|2 ·X)2, (11)
where i∈{0,1}, and X∈{±1} for QPSK, X∈{±1, ±3} for 16-QAM, and X∈{±1, ±3, ±5, ±7} for 64-QAM constallations. Since the term (Zk,n,i 2) is common to all the branch metrics at each stage of the Viterbi decoding process, this term can be subtracted from all the path metrics without changing the MLSE (described below) comparison results. Thus,
BM k,n,i=−2Z k,n,i ·|H k,n|2 ·X+(|H k,n|2 ·X)2 (12)
For QPSK, X∈{±1}, and the term (|Hk,n|2·X)2=|Hk,n|4 is common to all the branch metrics at each decoding stage. Thus, the term can be subtracted from all the path metrics of all the states without changing the comparison results. The scaling factor (2) can also be removed. Thus,
BM k,n,i =−Z k,n,i ·|H k,n|2 ·X for QPSK. (13)
{ W k , n } ≡ W k , n , 0 =  H k , n  · X k , n , 0 + ( H k , n , 0 · v k , n , 0 + H k , n , 1 · v k , n , 1 )  H k , n  and ( 14 ) { W k , n } ≡ W k , n , 1 =  H k , n  · X k , n , 1 + ( H k , n , 0 · v k , n , 1 - H k , n , 1 · v k , n , 0 )  H k , n  ( 15 ) From which, the Viterbi decoder branch metrics, BMk,n,i, can be derived as follows:
BM k,n,i=(W k,n,i −|H k,n |·X)2 =W k,n,i 2−2W k,n,i ·|H k,n |·X+(|H k,n |·X)2, (16)
where i∈{0,1}, and X∈{±1} for QPSK, X∈{±1, ±3} for 16-QAM, and X∈{±1, ±3, ±5, ±7} for 64-QAM constellations. As before, the branch metric can be simplified to
BM k,n,i=−2W k,n,i ·|H k,n |·X+(|H k,n |·X)2 (with CSI), (17)
BM k,n,i =−W k,n,i ·|H k,n |·X for QPSK (with CSI). (18)
BM k,n,i=−2Z k,n,i ·X+(|H k,n |·X)2 (with CSI). (19)
Again, for QPSK, X∈{±1}, and
BM k,n,i =−Z k,n,i ·X for QPSK (with CSI). (20)
P k,n =e −jΦk,n ·Y k,n =|H k,n |·X k,n +e −jΦk,n ·v k,n (21)
BM k,n,i=−2P k,n,i ·|H k,n |·X+(|H k,n |·X)2 (with CSI), (22)
where i∈{0,1}, and X∈{±1} for QPSK, X∈{±1, ±3} for 16-QAM, and X∈{±1, ±3, ±5, ±7} for 64-QAM constellations, and thus
BM k,n,i =−P k,n,i |H k,n |·X for QPSK (with CSI). (23)
Once each BMi is calculated by the branch metric computation unit 332 for the current symbol Yn, they are transferred to the Maximum Likelihood Sequence Estimation unit 334 (or “MLSE”), which performs conventional survivor path and path metric updating, as well as most likely path trace back to recover {circumflex over (X)}n. For more information regarding MLSE 334 operation and functions, see e.g. Lou, Hui-Ling “Implementing the Viterbi Algorithm”, IEEE Signal Processing Magazine, September 1995, pages 42-52 which is incorporated herein fully by reference.
The MLSE unit 334 estimates {circumflex over (X)}n representing the most likely symbol transmitted ({tilde over (X)}n). In view of known limitations of the Viterbi algorithm {circumflex over (X)}n may or may not be the same as {tilde over (X)}n. As shown in FIG. 3, Zn or Pn may also be fed to a hard-decision slicer 350 that, based on slicing rules detailed below with reference to FIGS. 6 and 7, will generate dec{ X n} which is turn sent to the channel estimator 310 to improve the channel estimates H n+1. More detail on the functions and interactions of the dynamic slicer 350 will be discussed below with reference to FIGS. 6 and 7.
Hk,0≈Hk,1≈Hk,2 . . . , (24)
where bk,0,0 is the Most Significant Bit (MSB). We further assume that the resultant |Hk,0|2 are to be quantized to 6 bits. Denoting ‘Λ’ a logical OR operation and ‘&’ a logical AND operation, we propose the following pseudocode to arrive at |Hk, 0|2:
if (b0,0,0Λb1,0,0Λb2,0,0Λ . . . Λb51,0,0=1)
|Hk,0|2=|Hk,0|2 & 1111110000 exit else
if (b0,0,1Λb1,0,1Λb2,0,1Λ . . . Λb51,0,1=1)
|Hk,0|2=|Hk,0|2 & 0111111000 exit else
if (b0,0,2Λb1,0,2Λb2,0,2Λ . . . Λb51,0,2=1)
|Hk,0|2=|Hk,0|2 & 0011111100 exit else
if (b0,0,3Λb1,0,3Λb2,0,3Λ . . . Λb51,0,3=1)
|Hk,0|2=|Hk,0|2 & 0001111110 exit else
|Hk,0|2=|Hk,0|2 & 0000111111 endif
As mentioned previously, in many practical receiver designs, hard-decision decoding is performed on the output symbols of the FEQ to find their closest constellation points in order to perform decision-directed equalization or channel estimation. To determine the hard decision, we denote Dec{ X k,n} as the closest QAM constellation point to X k,n, using hard decision decoding, and X k,n is computed using Equation 1,
X _ k , n = Y k , n H _ k , n = X k , n + v k , n H _ k , n , where Xk,n,i∈{±1} for QPSK, Xk,n,i∈{±1, ±3} for 16-QAM, and Xk,n,i∈{±1, ±3, ±5, ±7} for 64-QAM constellations, and i∈{0,1} represents the in-phase (I) and quadrature (Q) components of Xk,n. In this case, the hard-decision decoded symbol, Dec{ X k,n}, can be computed by rounding X k,n,i to its corresponding nearest constellation point, Xk,n,i, in each dimension.
Z k,n =| H k,n|2 · X k,n =|H k,n|2 ·X k,n +H* k,n ·v k,n, (28)
P k,n =| H k,n |· X k,n =|H k,n |·X k,n +e −jΦ k,n ·v k,n, (29)
If ((|Zk,n,i|−2| H k,n|2)≦0), Dec{ X k,n,i}=±1,
otherwise Dec{ X k,n,i}=±3.
if ((|Zk,n,i|−2| H k,n|2)≦0), Dec{ X k,n,i}=±1, exit;
if ((|Zk,n,i|−3| H k,n|2)≦0), Dec{ X k,n,i}=±3, exit;
if ((|Zk,n,i|−6| H k,n|2)≦0), Dec{ X k,n,i}=±5, exit;
Dec{ X k,n,i}=±7.
if ((|Pk,n,i|−2| H k,n|)≦0), Dec{ X k,n,i}=±1,
if ((|Pk,n,i|−2| H k,n|)≦0), Dec{ X k,n,i}=±1, exit;
if ((|Pk,n,i|−4| H k,n|)≦0), Dec{ X k,n,i}=±3, exit;
if ((|Pk,n,i|−6| H k,n|)≦0), Dec{ X k,n,i}=±5, exit;
BM k,n,i=−( X k,n,i −m)·sign(b i), (31)
where m=0 for QPSK, m∈{0,2} for 16-QAM and m∈{0,2,4} for 64-QAM constellations, and bi∈{±1}.
To improve the Viterbi decoder performance, CSI may be incorporated to normalize the expected noise power appropriately as described above in Section 1. If Wk,n is demodulated using From Equation 18, the piecewise branch metric should be (±Wk,n,i·|Hk,n|). That is, the slope of the branch metrics will be ±|Hk,n|, depending on whether the sign of bi is −1 or 1. Furthermore, the piecewise decision regions are separated by m·|Hk,n| (see Equations 14 and 15), where m=0 for QPSK, m∈{0,2} for 16-QAM and m∈{0,2,4} for 64-QAM constellations. In short, the branch metrics will be of the form,
BM k,n,i =−|H k,n|(|W k,n |−|H k,n |·m)·sign(b i) (with CSI), (32)
BM k,n,i =−|H k,n|2(| X k,n,i |−m)·sign(b i) (with CSI) (34)
BM k,n,i=−(|Z k,n,i |−|H k,n|2 ·m)·sign(b i), (35)
BM k,n,i =−|H k,n|(|P k,n,i |−|H k,n |·m)·sign(b i), (36)
The previous two sections assume a single receive antenna is used. This section addresses the MRC multiple receive antennae case. A simplified conceptual block diagram of a D-path receive diversity system is shown in FIG. 8 for a single subcarrier k (channel mode 801). In this example, Xk,n was received through D independent pathways and the D received symbols Yk,n (0), Yk,n (1), . . . Yk,n (D−1) are to be demodulated and decoded. In Maximum Ratio Combining (MRC), the received symbols are co-phased to provide coherent voltage addition and are individually weighted to provide optimal SNR. Assuming that perfect channel knowledge is available at the receiver, each receiver chain has the same average noise power σ2, and the channel is flat-fading, the optimal MRC output generated by the MRC combine FEQ and demodulation unit 810, is given by
X _ k , n = ∑ d = 0 D - 1 H k , n ( d ) * Y k , n ( d ) ∑ d = 0 D - 1  H k , n ( d )  2 . ( 37 ) Substituting Yk,n (d)=Hk,n (d)·Xk,n+vk,n (d), Equation (37) may be rewritten as (38)
This signal is demodulated, and the I and Q components,
Re { X _ k , n } and
Im { X _ k , n } , are sent to the Viterbi decoder 830.
X _ k , n CSI = ∑ d = 0 D - 1 H k , n ( d ) * Y k , n ( d ) ∑ d = 0 D - 1  H k , n ( d )  2 . ( 40 ) H ^ k , n = ∑ d = 0 D - 1  H k , n ( d )  2 , ( 41 ) and similar to the one receive antenna case, the branch metric computation handled by the branch metric computation unit 1032 of the Viterbi decoder 1030 can be expressed as:
BM k,n,i=−2{tilde over (X)} k,n,i CSI ·Ĥ k,n ·X+(Ĥ k,n ·X)2 (MRC with CSI), (42)
where i∈{0,1}, and X∈{±1} for QPSK, X∈{±1, ±3} for 16-QAM, and X∈{±1, ±3, ±5, ±7} for 64-QAM constellations. For QPSK, the relationship simplifies to
BM k,n,i =− X k,n,i CSI ·Ĥ k,n ·X (MRC with CSI). (43)
Z k,n=Σd=0 D−1 H k,n (d) *·Y k,n (d) (44)
BM k,n,i=−2Z k,n,i ·X+Ĥ k,n 4 ·X 2 (MRC with CSI), (45)
where i∈{0,1}, and X∈{±1} for QPSK, X∈{±1, ±3} for 16-QAM, and X∈{±1, ±3, ±5, ±7} for 64-QAM constellations. For QPSK, the branch metrics simplify to
BM k,n,i =−Z k,n,i ·X for QPSK (MRC with CSI). (46)
BM k,n,i=−(|Z k,n,i |−Ĥ k,n 2 ·m)·sign(b i) (47)
In a practical MRC receiver design, the received signal in each receive pathway will pass through an Automatic Gain Control (AGC) (not shown) prior to frequency domain conversion through an individual pathway FFT (not shown) so that the dynamic range of the Analog-to-Digital Convertor (ADC) can be fully utilized. A simplified conceptual block diagram illustrating this feature is shown in FIG. 9. A gain, denoted g(d) for branch or pathway d is applied to the received symbol Yk,n (d) as noted in the channel model 901. Thus, the available received symbol at the receiver is,
{tilde over (Y)} k,n (d) =g (d) ·Y k,n (d), (48)
{tilde over (H)} k,n (d) =g (d) ·H k,n (d), (49)
Z k,n=Σd=0 D−1(g (d))−2 {tilde over (H)} k,n (d) *·{tilde over (Y)} k,n (d) (51)
BM k,n,i=(−|Z k,n,i |+ k,n 2 ·m)sign(b i) (MRC with CSI) (52)
H ⋓ k , n = ∑ m = 0 D - 1 ( g ( m ) ) - 2  H ~ k , n ( m )  2 ( 53 ) and where m=0 for QPSK, m∈{0,2} for 16-QAM and m∈{0,2,4} for 64-QAM constellations, and bi∈{±1}.
Z k,n ={tilde over (H)} k,n (o) *·{tilde over (Y)} k,n (0) +β·{tilde over (H)} k,n (1) *·{tilde over (Y)} k,n (1) (54)
BM k,n,i=(−|Z k,n,i|+({tilde over (H)} k,n (0)2 +β·{tilde over (H)} k,n (1)2)·m)·sign(b i) (MRC with CSI) (55)
These practical receiver branch metric calculations may be conveniently undertaken in the branch metric computation unit 932 of the Viterbi decoder 930 as part of the MRC capable receiver baseband processing unit 900 shown in FIG. 9. If utilized, simulation results show at least a I dB SNR improvement.
Turning briefly to FIG. 1, FIG. 1 illustrates a wireless communications transceiver 100 capable of implementing the receiver baseband processing units shown in FIG. 3 or 4. In this embodiment, inbound RF signals conveying a 802.11a/g or 802.16a compliant frame of OFDM encoded symbols are picked up by the duplex antenna 110 and routed to the RF receiver unit 115 of a receiver 150 arranged in a manner consistent with the present invention. The RF receiver unit 115 performs routine downconversion and automatic gain control of the inbound RF signals, and presents an analog baseband signal containing the aforementioned 802.11a/g or 802.16a frame of OFDM symbols to the receive baseband processor 120, which can conveniently comprise the receiver baseband processing unit 300, 400 described above. Generally speaking, the receive baseband processor 120 performs symbol demodulation of each inbound 802.11a/g or 802.16a compliant frame to recover bitstream data for receiver synchronization (preamble), frame or packet definition (header), or the actual inbound data of interest (payload). Consistent with the present invention, this processor 120 may include either units 300 or 400 to carry out joint decoding and equalization or reduced complexity decoding as described above.
Though only a single duplex antenna arrangement is shown in FIG. 1, the transceiver 100 can be easily adapted to incorporate multiple receive pathways or chains to take advantage of selection diversity or MRC diversity techniques as discussed above. To illustrate, consider the transceiver 101 in FIG. 11 which includes plural receive antenna (e.g. antennae 114, 111, and 112) respectively coupled to individual receive pathway RF receivers (e.g 115, 116 and 117) as part of the receiver 151. In turn, the individual baseband signals recovered by these distinct receive pathway RF receivers are fed to a common receive baseband processor 121, which can conveniently implement an MRC aware receive baseband processing unit 1000 or 900 shown in FIGS. 10 and 9 respectively. A separate antenna 113 is shown in FIG. 11 to radiate the RF signal generated by the transmitter 160, though other types of antennae configuration may be utilized, as is known in the art.
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