Source: https://www.google.co.uk/patents/US9131569
Timestamp: 2017-12-16 09:26:20
Document Index: 160429134

Matched Legal Cases: ['art=0', 'Application No. 201180022813', 'Application No. 2010', 'Application No. 201280044038', 'Application No. 201280054168', 'Application No. 201180004266', 'Application No. 201080053889', 'Application No. 11777867', 'Application No. 10819249', 'Application No. 2012', 'Application No. 2013', 'Application No. 099131743']

Patent US9131569 - AC driven solid state lighting apparatus with LED string including switched ... - Google Patents
A diode selection circuit for a light emitting apparatus according to some embodiments includes a plurality of light emitting devices coupled in series. The diode selection circuit includes a comparator configured to receive a rectified AC input signal and a reference voltage and to generate a control...https://www.google.co.uk/patents/US9131569?utm_source=gb-gplus-sharePatent US9131569 - AC driven solid state lighting apparatus with LED string including switched segments
Publication number US9131569 B2
Application number US 13/919,528
Also published as CN102870501A, CN102870501B, EP2567597A1, EP2567597A4, EP2567597B1, US8476836, US20110273102, US20130285564, WO2011139624A1
Publication number 13919528, 919528, US 9131569 B2, US 9131569B2, US-B2-9131569, US9131569 B2, US9131569B2
Inventors Antony P. Van de Ven, Terry GIVEN
Patent Citations (402), Non-Patent Citations (52), Referenced by (3), Classifications (9), Legal Events (1)
US 9131569 B2
1. A circuit for a light emitting apparatus including a plurality of light emitting devices coupled in a series, the diode selection circuit comprising:
a comparator configured to receive a rectified AC input signal and a reference voltage and to generate a control signal in response to comparison of the rectified AC input signal with the reference voltage;
a voltage controlled current source configured to supply a current that is proportional to a voltage of the rectified AC input signal to the plurality of light emitting diodes in response to the rectified AC input signal; and
a transistor configured to receive the control signal and to shunt current away from more than one of the plurality of light emitting devices in response to the control signal.
2. The circuit of claim 1, wherein the voltage controlled current source comprises a first PNP bipolar transistor, an emitter resistor coupled to an emitter of the first PNP bipolar transistor, a diode coupled to a base of the first transistor, a first resistor coupled to a first terminal of the diode and a second resistor coupled to a second terminal of the diode, and a terminal configured to receive the rectified AC input signal coupled to the emitter resistor and to the first resistor.
3. The circuit of claim 2, wherein a collector of the first PNP bipolar transistor is coupled to the series of light emitting devices.
4. The circuit of claim 2, further comprising a second transistor including a base coupled to a collector of the first PNP bipolar transistor and a collector coupled to the emitter of the first PNP bipolar transistor.
an adjustable current sink coupled to the voltage controlled current source.
6. The circuit of claim 5, wherein the adjustable current sink is coupled to the base of the first PNP bipolar transistor.
7. The circuit of claim 6, wherein the adjustable current sink comprises a third transistor having a collector coupled to the base of the first PNP bipolar transistor and an emitter coupled to ground, and a diode coupled to a base of the third transistor.
8. The circuit of claim 7, wherein the base of the third transistor is configured to receive a pulse width modulation (PWM) control signal configured to control a conductivity of the third transistor.
9. The circuit of claim 1, wherein the diode selection circuit is configured to shunt current away from the at least one light emitting device in response to a voltage level of the rectified AC input signal being below a threshold level.
10. The circuit of claim 1, wherein the switch comprises a field effect transistor, and wherein the control signal is applied to a gate of the field effect transistor.
11. The circuit of claim 10, wherein the switch further comprises a second transistor coupled in a cascode configuration with the field effect transistor so that conductivity of the second transistor is controlled by the field effect transistor.
12. The circuit of claim 11, wherein the second transistor comprises a bipolar transistor including a base, a collector and an emitter, and wherein a drain of the field effect transistor is coupled to the emitter of the bipolar transistor, and the collector of the bipolar transistor is coupled to an anode of the at least one light emitting device.
13. The circuit of claim 12, further comprising a resistor coupled between the drain of the field effect transistor and the emitter of the bipolar transistor.
14. A circuit for a light emitting apparatus including a plurality of light emitting devices coupled in a series, the diode selection circuit comprising:
a voltage controlled current source configured to supply a current to the plurality of light emitting diodes that is proportional to the rectified AC input signal;
a switch configured to receive the control signal and to shunt current away from at least one of the plurality of light emitting devices in response to the control signal; and
an adjustable current sink coupled to the voltage controlled current source, wherein the adjustable current sink is configured to be controlled in response to a pulse width modulation (PWM) signal.
15. A circuit for a light emitting apparatus including a plurality of light emitting devices coupled in a series, the diode selection circuit comprising:
a voltage controlled current source configured to supply a current that is proportional to a voltage level of the rectified AC input signal to the plurality of light emitting diodes in response to the rectified AC input signal; and
a switch configured to receive the control signal and to shunt current away from at least one of the plurality of light emitting devices in response to the control signal;
wherein the voltage controlled current source comprises a first PNP bipolar transistor, an emitter resistor coupled to an emitter of the first PNP bipolar transistor, a diode coupled to a base of the first transistor, a first resistor coupled to a first terminal of the diode and a second resistor coupled to a second terminal of the diode, and a terminal configured to receive the rectified AC input signal coupled to the emitter resistor and to the first resistor.
This application is a continuation of U.S. patent application Ser. No. 12/775,842 filed May 7, 2010, the disclosure of which is incorporated herein by reference in its entirety.
The present invention relates to solid state lighting, and more particularly to lighting fixtures including solid state lighting components:
Although solid state light sources having high CRI and/or high efficiency have been demonstrated, one problem with the large-scale adoption of such light sources in architectural applications is that solid state lighting devices are typically designed to be driven using direct current (DC) power, while electric power is distributed using alternating current (AC), which is more efficient for long distance power distribution. Typically, a solid state lighting source is provided or coupled with a power converter that converts AC power into DC power, and the DC power is used to energize the light source. However, the use of such power converters increases the cost of the lighting source and/or the overall installation, and introduces additional efficiency losses.
Some attempts at providing AC-driven solid state lighting sources have involved driving an LED or string or group of LEDs using a rectified AC waveform. However, because the LEDs require a minimum forward voltage to turn on, the LEDs may turn on for only a part of the rectified AC waveform, which may result in visible flickering, may undesirably lower the power factor of the system, and/or may increase resistive loss in the system.
A circuit for a light emitting apparatus according to some embodiments includes a plurality of light emitting devices coupled in series. The circuit includes a comparator configured to receive a rectified AC input signal and a reference voltage and to generate a control signal in response to comparison of the rectified AC input signal with the reference voltage, a voltage controlled current source configured to supply a current to the plurality of light emitting diodes that is proportional to the rectified AC input signal, and a switch configured to receive the control signal and to shunt current away from at least one of the plurality of light emitting devices in response to the control signal.
The voltage controlled current source may include a first transistor, an emitter resistor coupled to an emitter of the first transistor, a diode coupled to a base of the first transistor, a first resistor coupled to a first terminal of the diode and a second resistor coupled to a second terminal of the diode, and a terminal configured to receive the rectified AC input signal coupled to the emitter resistor and to the first resistor. The diode may include a Zener diode.
A collector of the first transistor may be coupled to the series of light emitting devices.
In some embodiments, the circuit may further include a second transistor including a base coupled to a collector of the first transistor and a collector coupled to the emitter of the first transistor.
The circuit may further include an adjustable current sink coupled to the base of the first transistor. The adjustable current sink may include a third transistor having a collector coupled to the base of the first transistor and an emitter coupled to ground, and a diode coupled to a base of the third transistor. The base of the third transistor is configured to receive a pulse width modulation (PWM) control signal configured to control a conductivity of the third transistor.
The circuit may be configured to shunt current away from the at least one light emitting device in response to a level of the rectified AC input signal being below a threshold level.
The switch may include a field effect transistor, and the control signal may be applied to a gate of the field effect transistor.
The switch may further include a second transistor coupled in a cascode configuration with the field effect transistor so that conductivity of the second transistor may be controlled by the field effect transistor.
The second transistor may include a bipolar transistor including a base, a collector and an emitter, and a drain of the field effect transistor may be coupled to the emitter of the bipolar transistor, and the collector of the bipolar transistor may be coupled to an anode of the at least one light emitting device.
The circuit may further include a resistor coupled between the drain of the field effect transistor and the emitter of the bipolar transistor.
The diode selection circuit may further include a voltage divider ladder configured to generate a plurality of reference voltages, a plurality of comparators configured to receive the rectified AC input signal and a respective one of the plurality of reference voltages and to generate respective control signals in response to comparison of the rectified AC input signal with the respective reference voltages, and a plurality of switches configured to receive respective ones of the control signals and to shunt current away from respective ones of the plurality of light emitting devices in response to the control signals.
The switch may include a field effect transistor including a gate terminal and source/drain terminals, one of the source/drain terminals may be coupled to an anode of the at least one light emitting device and another of the source/drain terminals may be coupled to an cathode of the at least one light emitting device.
The circuit may further include a slew rate control capacitor coupled between the gate terminal of the field effect transistor and the anode of the at least one light emitting device.
The circuit may further include a voltage divider ladder configured to generate a plurality of reference voltages, a plurality of comparators configured to receive the rectified AC input signal and a respective one of the plurality of reference voltages and to generate respective control signals in response to comparison of the rectified AC input signal with the respective reference voltages, and a plurality of switches configured to receive respective ones of the control signals and to shunt current away from respective ones of the plurality of light emitting devices in response to the control signals.
The switch may include a bipolar transistor including a base coupled to an output of the comparator and having an emitter coupled to ground and a collector coupled to an anode of the at least one light emitting device.
A lighting apparatus according to some embodiments includes a terminal configured to receive an AC power signal, a full wave rectifier configured to generate a rectified AC input signal in response to the AC power signal, a reference voltage generator configured to generate a DC reference voltage in response to the rectified AC input signal, a plurality of light emitting devices coupled in series, a voltage controlled current source configured to supply a current to the plurality of light emitting diodes that is proportional to the rectified AC input signal, and a diode selection circuit including a comparator configured to receive the rectified AC input signal and the reference voltage and to generate a control signal in response to comparison of the rectified AC input signal with the reference voltage. The diode selection circuit includes a switch configured to receive the control signal and to shunt current away from at least one of the plurality of light emitting devices in response to the control signal.
The diode selection circuit may be configured to shunt current away from the at least one light emitting device in response to a level of the rectified AC input signal being below a threshold level.
The lighting apparatus may further include a current supply circuit coupled to an anode of a first one of the plurality of light emitting devices, the current supply circuit including a bipolar transistor including a collector coupled to the anode of the first light emitting device, an emitter resistor coupled to an emitter of the bipolar transistor, and a bias circuit coupled to a base of the bipolar transistor.
FIG. 1C is a schematic circuit diagram illustrating series interconnection of light emitting devices (LEDs) in a solid state lighting apparatus.
FIG. 2 is a schematic circuit diagram illustrating a drive circuit for a solid state lighting apparatus in accordance with various embodiments of the invention.
FIGS. 3A, 3B and 3C are schematic circuit diagrams illustrating voltage controlled current sources for a solid state lighting apparatus in accordance with various embodiments of the invention.
FIG. 3D is a graph of simulated output current of voltage controlled current sources for a solid state lighting apparatus in accordance with various embodiments of the invention.
FIG. 4A is a graph illustrating output current, and FIG. 4B is a graph illustrating input and output voltage waveforms, for solid state lighting systems in accordance with various embodiments of the invention.
FIG. 5 is a schematic circuit diagram illustrating a diode selection circuit according to some embodiments of the invention.
FIG. 6 is a graph illustrating a rectified voltage signal according to some embodiments.
FIGS. 7, 8 and 9 are schematic circuit diagrams illustrating diode selection circuits according to further embodiments of the invention.
FIG. 10 is a schematic circuit diagram illustrating a voltage controlled current sources with an adjustable current sink for a solid state lighting apparatus in accordance with some embodiments of the invention.
“Warm white” generally refers to white light that has a CCT between about 3000K and 3500K. In particular, warm white light may have wavelength components in the red region of the spectrum, and may appear yellowish to an observer. Warm white light typically provides a relatively high CRI, and accordingly can cause illuminated objects to have a more natural color. For illumination applications, it is therefore desirable to provide a warm white light.
The LEDs 22 in the lighting apparatus 10 may include white/BSY emitting LEDs, while the LEDs 24 in the lighting apparatus may emit red light. The LEDs 22, 24 in the lighting apparatus 10 may be electrically interconnected in respective strings, as illustrated in the schematic circuit diagram in FIG. 1G. As shown therein, the LEDs 22, 24 may be interconnected such that the white/BSY LEDs 22 are connected in series to form a first string 34A. Likewise, the red LEDs 24 may be arranged in series to form a second string 34B. Each string 32, 34 may be connected to a respective anode terminal 23A, 25A and a cathode terminal 23B, 25B.
Although two strings 34A, 34B are illustrated in FIG. 1C, it will be appreciated that the lighting apparatus 10 may include more or fewer strings. Furthermore, there may be multiple strings of white/BSY LEDs 22, and multiple strings of red or other colored LEDs 24.
Referring now to FIG. 2, a drive circuit 100 for driving a plurality of solid state light emitting devices in a solid state lighting apparatus is illustrated. In particular, the drive circuit 100 is configured to drive a plurality of LEDs, or sets of LEDs, that are connected in series. As illustrated in FIG. 2, the sets of LEDs may include and sets S1, S2, . . . SN. Each set of LEDs S1 to SN may include one or more LEDs connected in series and/or parallel. For example, as illustrated in set S1, the LEDs in a set may be connected in respective series strings that are themselves connected in parallel. The total number of LEDs in series may be selected so that the circuit has a suitably high efficiency when the input voltage to the string is at the maximum line voltage. Efficiency may also be increased by keeping the voltage across the selected LEDs close to an applied AC voltage, as discussed below.
An alternating current signal VAC is applied to a full wave rectifier bridge B1. The output of the full wave rectifier bridge B1 is a rectified AC signal Vrect. The rectified AC signal Vrect may be applied to a current source, such as a series resistor or a voltage controlled current source, which supplies current to the first set S1 of LEDs. Subsequent sets of solid state lighting devices S2 to SN are connected in series with the first set S1. Control lines L2, L3, . . . LN are coupled to anodes of respective sets of LEDs S2 to SN.
The drive circuit 100 includes a diode selection circuit 10, a power supply 12, and a voltage divider 14. The rectified signal Vrect is divided by the voltage divider 14 and supplied as a control signal to the diode selection circuit 10. The rectified signal Vrect is also supplied to a power supply 12, which responsively generates a substantially steady reference voltage Vref, which is also supplied to the diode selection circuit 10. In some embodiments, the power supply 12 may generate more than one reference voltage, e.g., a family of reference voltages of various levels.
The diode selection circuit 10 is configured to dynamically adjust the number of LEDs that are being driven by the rectified AC voltage Vrect in response to a level of the rectified AC signal Vrect, so that a voltage drop of diodes that are being driven at any given point in time is less than the level of the rectified AC signal Vrect. In this manner, the AC waveform may be more fully utilized, potentially increasing light output from the circuit, improving the power factor of the circuit, and or reducing visible flicker of light generated by the circuit. In general, the diode selection circuit is designed to keep the instantaneous LED string voltage close to but always slightly less than Vrect, which may reduce or minimize the voltage dropped across the current source 20, thereby increasing or maximizing efficiency, and more fully utilizing the AC waveform to generate more light. In some embodiments, compared to apparatus including non-switched LED strings, the THD may improve from 85% to 25%.
In operation, when the level of the rectified AC voltage Vrect is low, only the first group LEDs S1 may be driven by the rectified AC signal Vrect, while the other sets of LEDs S2 to SN may be switched out of the circuit and/or bypassed, so that they are not being driven by the rectified signal Vrect. As the rectified AC signal Vrect increases, successive sets of LEDs S2 to SN may be switched into the circuit so that they are being driven by the rectified AC signal Vrect until all of the sets of LEDs S1 to SN have been switched into the circuit and are being driven by the rectified AC signal Vrect.
In a switching arrangement that utilizes a resistor (Rseries) as the current source, the voltage across Rseries starts off small, and increases until the next switching threshold is reached, at which point it returns to zero and the process begins again. This is during the first half-cycle of Vrect(t), where Vrect(t) is rising. During the second half-cycle of Vrect(t), Vrect(t) is falling and the LED current is the mirror image of the first half-cycle. It is therefore sufficient to consider only the first half-cycle. Assuming a Binary Weighted Switching approach in which LEDs are controllably switched in groups of 1, 2, 4, 8, etc., . . . , the LED current is given by:
Vrect(t)=Vpeak*|sin(2*□*Fac*t)|−2*Vf
Vr_series=Vrect(t)−floor([Vrect(t)−Vdrop]/Vled)*Vled
ILED(t)=[Vrect(t)−floor([Vrect(t)−Vdrop]/Vled)*Vled]/Rseries
Rather than using a series resistor, some embodiments drive the bank of dynamically switched LEDs with a Voltage Controlled Current Source (or sink)—a VCCS, as illustrated in FIG. 3A.
According to some embodiments, current to the LED string is provided by a voltage controlled current source (VCCS) 20A. The current source 20A includes a PNP transistor Q1 having an emitter resistor RE coupled to the emitter thereof and a bias circuit including resistors RH and RL and a zener diode VZ coupled to the base thereof. Output current IOUT is sourced by the current source 20 according to the following equations:
I SOURCE = V Z R E + Vrect ( R H R H + R L ) ( 1 R E ) I SOURCE = I 0 + kVrect
Accordingly, the output current is proportional to the rectified input voltage Vrect. This circuit may therefore further regulate the output current to help reduce spikes as sets of LEDs are switched into/out of the circuit. It will be appreciated that the zener diode could be replaced with another type of diode that has a voltage drop that is substantially matched to the base-emitter junction of the transistor Q1.
A more general illustration of a VCCS 20B is shown in FIG. 3B, and will be discussed below.
The VCCS 20B illustrated in FIG. 3B includes resistors R1, R2 and R3, diode D1 and PNP transistor Q1. Assuming D1 and Q1 are matched and thermally coupled, Vf(D1)=Vbe(Q1). Assuming transistor Q1 has sufficient gain, then the voltage across resistor R3 is the same as the voltage across resistor R1.
Referring to FIG. 3C, a Sziklai pair 122 may be used in a VCCS current source 120 according to some embodiments. The Sziklai pair 122 includes a PNP transistor Q1 and an NPN transistor Q2. The collector of the PNP transistor Q1 is coupled to the base of the NPN transistor Q2, and the emitter of the PNP transistor Q1 is coupled to the collector of the NPN transistor Q2, as shown in FIG. 3C. The emitter of the NPN transistor Q2 is coupled to the switched LED string 124, which is controlled by the diode selection circuit 10. A Sziklai pair may have a low saturation voltage and temperature coefficient, but very high gain (and the high power transistor Q2 is NPN). Darlington transistors may be used, but the saturation voltage is larger and there are two base-emitter junctions requiring thermal compensation. MOSFET current sources may also be used in some embodiments.
If we then assume that Vrect(t)>>Vf(D1), the voltage across R1 is given by:
Vr1=Vrect(t)−Vrect(t)*R2/(R1+R2)
which re-arranges to give:
Vr1=Vrect(t)*R1/(R1+R2)
Vr3=Isource*R3=Vr1
Isource*R3=Vrect(t)*R1/(R1+R2)
Isource=[Vrect(t)/R3]*[R1/(R1+R2)]
Isource=[Vrect(t)/R3]*[1/(R1+R2)/R1]
Isource=[Vrect(t)/R3]*[1/(1+R2/R1)]
Isource=Vrect(t)/[R3*(1+R2/R1)]
Isource(t)=Vrect(t)/Req
Req=R3*(1+R2/R1)
Although it is called a constant current source, it is clear that the current Isource is directly proportional to Vrect(t). Hence the term Voltage Controlled Current Source (VCCS) is more properly used to describe the current source 20B.
Typical circuit designs attempt to remove the dependence on Vrect(t). However, in some embodiments, it may be more appropriate not to have a constant current. If D1 is used to compensate for the static and dynamic base-emitter voltage Vbe of Q1 (−2.2 mV/K), a Voltage Controlled Current Source (VCCS) is provided. The VCCS has a minimum dropout voltage of:
Vdropout=Vcesat(Q1)+Isource*R3
Where Vcesat(Q1) increases with increasing Isource. It is sufficient to use Isource_max when calculating Vcesat(Q1) and the voltage drop across R3. If a string of dynamically switched LEDs is driven with a VCCS, and the circuit is designed so that Vdrop>Vdropout (in other words the dynamic switching controller ensures the VCCS always has enough headroom to regulate Isource), then the LED current is always proportional to Vrect(t).
Vrect(t)=Vpeak*|sin(2*□*Fac*t)|=2*Vf
Assuming Vpeak*|sin(2*□*Fac*t)|>>2*Vf, then:
Vrect(t)=Vpeak*|sin(2*□*Fac*t)|
Isource(t)=Vpeak*|sin(2*□*Fac*t)|/Req
Isource(t)=[Vpeak/Req]*|sin(2*□*Fac*t)|
Isource(t)=Ipeak*|sin(2*□*Fac*t)|
where Ipeak=Vpeak/Req and Req=R3*(1+R2/R1)
The input current is therefore substantially sinusoidal, although the foregoing analysis has ignored the bridge rectifier, D1 and Vdrop voltages, which cause a small amount of dead time and prevent Isource(t) from being perfectly sinusoidal.
A simulation comparing a VCCS with a resistor current source is shown in FIG. 3D. Curve 301 is a graph of Isource(t) for a dynamically switched LED driver circuit including a VCCS current source with an equivalent resistance Req=2.22 k□, while curve 302 is a graph of Isource(t) for a dynamically switched LED driver circuit including a series resistor current source having a resistance R series=270□□.
Both circuits use the same Dynamically Switched LED string which is non-linear (but not binary), and designed for minimum loss with 7 switches. Two LEDs are always in series, to provide for a free low voltage supply (and to allow for flicker-free dimming).
For a resistor current source using Rseries=270□, the total harmonic distortion is 77.4%. For the VCCS with Req=2.22 k□□, the □input current is very close to a pure sinusoid, and has 2.4% THD. The power factor is greater than 0.99.
In embodiments in which the LED current is set using a Voltage Controlled Current Source, such as the VCCS circuits 20A, 20B illustrated in FIG. 3A and/or FIG. 3B, to set the LED current proportional to Vrect, the output current waveform may exhibit some spikes at the switching instant, as illustrated in FIG. 4A, which is a graph 403 of simulated output current for a circuit according to some embodiments. Current spikes are visible in the output current signal 403 at the switching instants.
When a set of k LEDs is switched in (or out) of the string, the LED string voltage rises (or falls) by k*Vled. This occurs fairly quickly—shorting a set is as fast as the shorting switch itself. The time taken removing the short circuit from a set is governed by the current source charging the stray capacitance of the set, which is fairly small.
Once a shorting switch is turned OFF (and assuming there are not minority carrier storage-time issues), the current source sets the slew rate as it charges up the stray capacitance. However, the current source is haversinusoidal, so over one quarter AC line period the current ranges from near zero to 1.4*Irms.
The uppermost LED anode connects to the collector of the PNP current source transistor Q4, so changes in the LED string voltage change the PNP collector voltage an equal amount.
This feeds current I=Cm*dV/dt from the collector to the base through the BJT Miller capacitance Cm, and causes the current source to misbehave, which is what creates the spikes shown in FIG. 4B.
It may be possible to reduce or avoid these spikes by designing the VCCS so they are reduced, or by controlling the slew rate of the switches.
Simulated example voltage and current waveforms for a drive circuit 100 according to some embodiments are illustrated in FIG. 4B. Referring to FIG. 4B, curve 401 is a rectified AC sine wave Vrect, while curve 402 is a graph of the voltage across the LEDs that are switched into the circuit at any given instant, that is, the sum of the voltage drops of all of the LEDs that are energized at a particular instant. As illustrated in FIG. 4A, as the rectified AC signal Vrect increases, successive groups of LEDs are switched into the circuit, and the output voltage increases in steps as successive sets of LEDs are energized.
According to some embodiments, the numbers of LEDs that are switched on at any given step and the threshold levels of Vrect that result in such sets being switched in our chosen so that the level of Vrect remains greater than the sum of the forward voltages of the energized LEDs at each successive step. Accordingly, as the level of Vrect rises, all LEDs that are switched into the circuit may remain energized when a successive set of LEDs is switched into the circuit.
As the level of Vrect falls, successive sets of LEDs may be switched out of the circuit to ensure that the level of the voltage across the energized diodes remains below Vrect at substantially each time instant.
Referring again to FIG. 2, once Vrect exceeds the voltage required to turn on the LEDs in set S1, the LEDs are forward biased and current begins to flow through the LEDs. The voltage dropped across these resistors is equal to Vrect-Vied (where Vled is the voltage across the LEDs in S1), so the current ramps up with the same slope as Vrect. When Vrect is sufficiently above Vled (as a function of losses in the current source), the diode selection circuit 100 switches the next set S2 of LEDs in series, so the total LED string voltage increases by Vnew_led_string, which may be chosen such that:
Vrect(t=Tswitch)>Vled+Vnew_led_string.
This ensures that when the new LED string is switched in series, the current through the current source is still slightly positive.
We can define dV=Vrect(Tswitch)−(Vled+Vnew_led_string). If dV>0, current through the LEDs drops down to a positive, non-zero value when the new set is switched into the series. If dV=0, the current drops down to zero when the new set is switched into the series.
If dV<0, the current tries to drop below zero, but the LEDs are unipolar so current cannot flow, and current continues to not flow until Vrect rises enough to make Vled>0.
Thus, when dV<=0, current in the LEDs is equal to zero, so dV should be made greater than zero keep current flowing through the diodes, regardless of current control methodology. If not, the current drops to zero at the switching edge, and stays there until Vrect is greater than the voltage drop of all LEDs switched into the series.
Furthermore, as dV becomes more negative the zero current period gets longer. So, the switching edge may give rise to notches (down/up to 0 A) in the input current waveform, if dV is not chosen well.
With a resistor as the input current source, all that is needed is dV>=0. With a current source, however, there is a finite dropout voltage Vcsdo, that may, for example, be about 1V, below which the current source is OFF. The constraint on dV when using a Voltage Controlled Current Source (VCCS) is therefore dV>=Vcsdo.
A drive circuit 100A including a diode selection circuit 10A is illustrated in more detail in FIG. 5. As illustrated therein, the diode selection circuit 10A includes a voltage divider ladder 18 including resistors R51 to R54 that generate a plurality of reference voltages Vref-1, Vref-2 and Vref-3 from the reference voltage Vref.
The diode selection circuit 10A further includes a plurality of comparators C51 to C53 and switching transistors Q51 to Q53 having gates coupled to respective outputs of the comparators C51 to C53. In the embodiments illustrated in FIG. 5, sources of the switching transistors Q51 to Q53 are coupled to ground, and drains of the switching transistors Q51 to Q53 are coupled to emitters of respective shunting bipolar transistors Q54 to Q56 through respective resistors R56, R57 and R58. Collectors of the bipolar shunting transistors Q54 to Q56 are coupled to respective control lines L1, L2, and L3, which are in turn coupled to anodes of respective ones of the sets S2, S3 and S4 of LEDs.
Reference voltages Vrect-1 to Vrect-3 generated by the voltage divider ladder 18 are applied to the non-inverting inputs of the comparators C51 to C53. In particular, the lowest reference voltage Vref-1 is applied to the non-inverting input of the comparator C51, the reference voltage Vref-2 is applied to non-inverting input of the comparator C52, and the highest reference voltage Vref-3 is applied to the non-inverting input of the comparator C53.
An optional voltage divider 16 supplies a bias voltage to the bases of the respective bipolar shunting transistors Q54 to Q56.
One or more of the sets S2 to S4 of series connected LEDs may be switched out of the drive circuit based on the level of the rectified AC input signal Vrect. In particular, based on a comparison of the level of Vrect′ with one of the reference voltages Vref-2 to Vref-3, the drive current of the LEDs may be shunted through one of the bipolar shunting transistors Q54 to Q56 through a respective emitter resistor R56 to R58 to ground, bypassing one or more sets of LEDs S2 to S4. Current through the remaining LEDs may be controlled by the bias level of the bipolar shunting transistor Q54 to Q56 through which the LED drive current is shunted.
Operation of the drive circuit 100A and the diode selection circuit 10A illustrated in FIG. 5 will now be described further reference to FIG. 6, which is a graph of one cycle of the scaled AC rectified signal Vrect′. Referring to FIGS. 5 and 6, when the value of Vrect′ is lower than the lowest reference voltage Vref-1, the non-inverting inputs to the comparators C51 to C53 are all less than the corresponding reference voltages Vref-2 to Vref-3 that are input into the non-inverting inputs to the comparators C51 to C53. Accordingly, each of the comparators C51 to C53 generates a high output voltage, which switches each of the respective switching transistors Q51 to Q53 to an ON state. In particular, the transistor Q51 is switched on, which also is emitter switches the bipolar shunting transistor Q54 to a conductive state. When the bipolar shunting transistor Q54 is conductive, the anode of the second set S2 of LEDs is coupled to ground though the switching transistor Q51, thereby bypassing the subsequent sets S2 to S4 of LEDs.
When the level of Vrect exceeds the forward voltage drop of the LEDs in the first set S1, the LEDs in the first set S1 will turn on and begin to generate light.
As the level of Vrect′ continues to increase, it reaches the lowest reference voltage Vrect-1. At that point, the output of the comparator C51 switches to low, thereby turning off the switching transistor Q51 and the bipolar shunting transistor Q54. However, because the voltage Vrect′ is still less than the second reference voltage Vref-2, comparators C52 and C53 continue to output a high level, so that the third and fourth sets S3 and S4 of LEDs continue to be switched out of the circuit and not energized. Thus, when the level of Vrect′ is between Vref-1 and Vref-2, only the first and second sets S1 and S2 of LEDs may be energized.
As Vrect′ continues to increase, it reaches the value of Vref-2, at which point the second comparator C52 also switches to a low output, turning off switching transistor Q52 and bipolar shunting transistor Q55. This switches the third set S3 of LEDs into the drive circuit, while continuing to bypass the fourth set S4 of LEDs.
Finally, as Vrect′ reaches the value of the third reference signal Vref-3, the third comparator C53 also switches to low output, turning off transistors Q53 and Q56 and switching the fourth set S4 of LEDs into the drive circuit.
The bipolar transistors Q54 to Q56 provide a means both for controlling current flow through the LEDs that are currently switched into the drive circuit, as well as providing an inexpensive and reliable transistor for sustaining the voltage of LEDs that are in the ON state. For example, if the transistor Q54 to Q56 were not included and the MOS transistors Q51 to Q53 were directly coupled to the output lines L1 to L3, when the Vrect′ greater than Vrect-3 and all three of the switching transistors Q51 to Q53 are switched off, the transistor Q51 would have to be capable of sustaining entire voltage drop over sets S2, S3 and S4 of LEDs. This may require a larger and more expensive MOS transistor. Furthermore, it might be difficult to regulate current at the output of the energized LEDs.
The bipolar shunting transistors Q54 to Q56 may be biased in a linear mode and emitter-switched by the switching transistors Q51 to Q53.
A circuit according to some embodiments can drive LEDs with a relatively high power factor. In general, “power factor” refers to how closely aligned the output current and voltage waveforms are over each cycle. Instead of switching current on and off only near peaks of the rectified waveform when an entire string is energized, current is drawn in steps as the rectified waveform changes. Thus, the output current may follow the input voltage waveform more closely in circuits according to some embodiments.
A drive circuit 100B including a diode selection circuit 10B according to further embodiments is illustrated in FIG. 7. The drive circuit 100B includes similar elements as the drive circuit 100A illustrated in FIG. 5, including a bridge rectifier B1, a power supply 12, and a voltage divider 14. Operation of these elements is similar to the operation of like elements in drive circuit 100A, and need not be described again in detail.
The diode selection circuit 10B includes a voltage divider ladder 18 that is configured to generate a range of reference voltages Vref-1 to Vref-3, which are supplied to the non-inverting inputs of respective comparators C71 to C73, as illustrated in FIG. 7. A scaled version Vrect′ of the rectified input voltage Vrect is supplied to the inverting inputs of the comparators C71 to C73. The outputs of the comparators C71 to C73 are coupled to respective gates of shunting MOS transistors Q71 to Q73 through bias resistors R75, R76 and R77. Drain and source terminals of the shunting transistors Q71 to Q73 are coupled to anode and cathode contacts, respectively, of respecting sets of LEDs S1, S2 and S3. Optional slew-rate control capacitors Cslew may be coupled between the gates and drains of the transistors Q71 to Q73.
Referring to FIGS. 6 and 7, when Vrect′ is less than Vref-1, all three comparators C71 to C73 output a high voltage level, which turns on the respective transistors Q71 to Q73, thereby bypassing sets S1, S2 and S3 of LEDs. When Vrect′ reaches Vref-1, comparator C73 outputs a low level which turns transistor Q73 off, thereby energizing set S3, while sets S1 and S2 remain bypassed.
Similarly, when Vrect′ reaches Vref-2, comparator C72 outputs a low level which turns transistor Q72 off, thereby energizing set S2, while set S1 remains bypassed.
Finally, when Vrect′ reaches Vref-3, comparator C71 outputs a low level, which turns transistor Q71 off, thereby energizing set S1, at which point all three sets S1 to S3 are energized and emit light.
The number of LEDs in a given set S1 to S3 may be varied to produce a desired turn-on characteristic. Furthermore, the values of resistors in the voltage divider ladder 18 may be selected to provide an appropriate voltage threshold for each set S1 to S3 depending on the number of LEDs in such sets.
Although circuits including three or four sets S1 to S4 are illustrated in FIGS. 3 through 7, a circuit according to some embodiments may have more or fewer sets of LEDs. Furthermore, each set may include one or more LEDs connected in series, parallel, or series/parallel as noted above.
In some embodiments, the first set S1 may include only a single LED in each branch to reduce the turn on voltage for the first string to a minimum level.
In some embodiments, the number of series LEDs in sets that are switched on at low voltages may be higher than the number of LEDs in sets that are switched on at higher voltages to match the sinusoidal shape of the rectified AC waveform, which may further reduce resistive losses in the apparatus. For example, referring to FIG. 5, set S2, which is switched into the circuit when Vrect′ reaches Vref-1, may include more LEDs in series than set S3, which is not switched into the circuit until Vrect′ reaches Vref-2.
Referring again to FIG. 7, it will be appreciated that the peak voltage across each switch is always limited to the total forward voltage of the LED segment it shorts out. All switches can thus be low voltage, and it is quite possible to use the same FET for all switches. The low-voltage FETs make this suitable for integration.
All N switches are ON and in series when Vrect(t) and Isource(t) are both low, so although the total on resistance is the sum of the N individual switch RDS_ON, conduction losses are low.
Furthermore because the switch voltage is relatively low, fairly low-voltage MOSFETs with much lower RDS_ON can be used. The switches that spend the most amount of time ON, must be low RDS_ON.
One potential disadvantage of the circuit of FIG. 7 is that, once the switches are all OFF, each gate drive sits at a higher voltage than the one below. However when all switches are ON, all of the FET Sources are at roughly the same potential. If the uppermost FET Q1 is turned off first, all of the lower FETs remain at the same potential. The switching sequence is thus:
uppermost FET OFF
second highest FET OFF
second-to-lowest FET OFF
• lowest FET OFF (all FETs now OFF)
• lowest FET ON
• second-to-lowest FET ON
second highest FET ON
• uppermost FET ON (all FETs now ON)
This can be exploited to implement a fully integrable design.
Referring to FIG. 8, a drive circuit 100C according to further embodiments is illustrated. The drive circuit 100C includes similar elements as the drive circuit 100A illustrated in FIG. 5, including a bridge rectifier B1 and a power supply 12. Operation of these elements is similar to the operation of like elements in drive circuit 100A, and need not be described again in detail.
The drive circuit 100C further includes a diode selection circuit 100 and a voltage controlled current source 20. The current source 20 is configured to supply current to the sets S0 to S3 of light emitting devices at a current level that is proportional to the rectified input voltage Vrect, which may improve the power factor of the circuit.
In particular, the diode selection circuit 100 includes a voltage divider 18 that includes a plurality of resistors R81 to R84 in series. The voltage divider 18 divides the rectified input voltage Vrect into a plurality of voltages VR1, VR2, VR3, which are applied to the inverting inputs of respective comparators C81 to C83. Outputs of the comparators C81 to C83 are applied to bases of respective bipolar transistors Q81 to Q83, which act as switches that controllably switch anodes of respective sets S1 to S3 of light emitting devices to ground. A reference voltage Vref is applied to the noninverting inputs of the comparators C81 to C83.
In operation, when the rectified input voltage Vrect is low, all three voltages VR1, VR2, VR3, which are applied to the inverting inputs of respective comparators C81 to C83, are less than the reference voltage Vref, and consequently all three comparators C81 to C83 output a high voltage, causing the transistors Q81 to Q83 to be in an ON state, bypassing sets S1 to S3 of light emitting devices.
As the rectified input voltage Vrect increases, the highest comparator input voltage VR1 reaches the level of the reference voltage Vref, and the output of the first comparator C81 switches to a low voltage, placing the transistor Q81 in the OFF state, and switching set S1 into the drive circuit to be energized by the rectified input voltage Vrect.
As the rectified input voltage Vrect increases further, the next highest comparator input voltage VR2 reaches the level of the reference voltage Vref, and the output of the second comparator C82 switches to a low voltage, placing the transistor Q82 in the OFF state, and switching set S2 into the drive circuit to be energized by the rectified input voltage Vrect.
These operations continue until all of the comparators C81 to C83 are switched low and all sets S1 to S3 of light emitting devices are energized.
Further embodiments of the invention are illustrated in FIG. 9, which illustrates an LED drive circuit 100D including a diode selection logic circuit 200 that is coupled to a plurality of comparators C1 to C6 through control lines L1 to L6. The outputs of the comparators control the conductivity of respective bypass switching transistors Q91 to Q96, which bypass respective sets of LEDs S1 to S6. The operation of comparators C91 to C96, transistors Q91 to Q96 and sets of LEDs S1 to S6 is similar to the corresponding elements described above with respect to FIG. 7. However, the circuit illustrated in FIG. 9 uses a series resistor Rseries to regulate current through the LED string.
The diode selection logic may be implemented in some embodiments as a programmed microcontroller that includes an integrated analog to digital converter, such as the PIC 16F88 microcontroller manufactured by Microchip Technology Inc. However, it will be understood that the diode selection logic circuit 200 can be implemented as an application specific integrated circuit (ASIC) or with discrete circuitry.
The diode selection logic circuit 200 may be configured to selectively bypass one or more sets of LEDs in response to a voltage level of the reference voltage Vref sensed at an ADC input to the diode selection logic. In particular embodiments the diode selection logic circuit 200 may be configured to switch the sets S1 to S6 using Binary Weighted Switching as described below.
Using Binary Weighted Switching, the LED current is given by:
I LED(t)=[Vrect(t)−floor([Vrect(t)−Vdrop]/Vled)*Vled]/Rseries
Assuming a peak voltage of 163V and 3V LEDs, a binary switched approach would use 64 LEDs, allowing the lamp to operate up to 137 Vrms, and 6 switches. In a binary switched approach, Set 1 includes 1 LED, Set 2 includes 2 LEDs, Set 3 includes 4 LEDs, Set 4 includes 8 LEDs, Set 5 includes 16 LEDs and Set 6 includes 32 LEDs. Accordingly, transistor switch Q1 (corresponding to the least significant bit of the control word) would switch one LED, and while transistor switch Q6 (corresponding to the most significant bit of the control word) would short 16 LEDs.
Switches Q1-Q6 are operated in a binary manner, choosing n(t) such that n(t)*Vled=Vac(t)−Vdrop. A simple way to do this would be to use a 6-bit A/D converter, driven by a voltage divider from the DC bus. A small zener diode or LED in series with the top of the voltage divider may be used to set Vdrop, and each of the ADC outputs directly drive the switches through inverters—when the ADC bit is 1 the relevant switch is off, when the ADC bit is zero the switch is ON. Accordingly, the diode selection logic circuit can, in some embodiments, simply include and appropriately configured analog to digital conversion circuit.
Returning to the previous example, assuming Vdrop=3V and t=2 ms, then:
Vac(t)=163*sin(2*□*60*2 ms)=122.9V
Vrect(t)=|Vac(t)|−2*Vf=120.9V is the DC bus
[Vrect(t)−Vdrop]/VLED=39.3
The voltage divider and ADC reference voltage will be selected such that the peak DC bus voltage gives an output of 111111 (all switches off) so the ADC output will be:
Nadc=floor([Vrect(t)−Vdrop]/VLED)=39=32+4+2+1=100111
MSB Switch Q6=OFF (LEDs 32-64 operating)
Switch Q5=ON (LEDs 16-31 short-circuited)
Switch Q4=ON (LEDs 8-15 short-circuited)
Switch Q3=OFF (LEDs 4-7 operating)
Switch Q2=OFF (LEDs 2, 3 operating)
LSB Switch Q1=OFF (LED 1 operating)
The total LED forward voltage is (16+4+2+1)*3V=117V and 3.9V is dropped across the series resistance Rseries.
The binary approach is not the only method of switch ordering, but it may result in high overall efficiency, although the gate drivers may be floating in such a circuit.
By its very design, the switched-LED approach has good efficiency. As mentioned above, a Binary Weighted Switching approach may theoretically gives high efficiency, as the LED string voltage can track the rectified AC line voltage with low error. However it has a two major disadvantages compared to Linear-Sequential Switching, namely, a much higher switching frequency, and fully floating switches requiring floating gate drivers.
The efficiency of a Linear-Sequential Switching circuit can be improved by driving the dynamically switched LED string with a Voltage Controlled Current Source, so that the LED current tracks the rectified AC line voltage. Assuming a reasonably sinusoidal AC supply, the VCCS drives the LEDs with:
Because the LED current is sinusoidal, it starts out at zero then ramps up sinusoidally, reaching a peak after 90 electrical degrees. The losses in the current source at any point are given by:
Psource(t)=[Vrect(t)−VLED_string(t)]*Isource(t)
Vdrop(t)=[Vrect(t)−VLED_string(t)]
So the VCCS instantaneous power dissipation is:
Psource(t)=Vdrop(t)*Ipeak*|sin(2*□*Fac*t)|
As the LED Segments are switched at discrete intervals, it is reasonable to consider the VCCS losses during each interval. The AC line period is long compared to the thermal time constant of the VCCS transistor(s), but short compared to typical heatsink time constants, so to reduce/minimize peak junction temperature, the average VCCS losses during each interval Psource_average[N] should be kept roughly constant.
When Isource(t) is small, keeping Psource_average[N] constant requires a large Vdrop(t). Conversely when Isource(t) is large, keeping Psource_average[N] constant requires a small Vdrop(t). Therefore in order to reduce/minimize the total number of switches and increase/maximize the efficiency, LED Segment voltages that are different should be chosen—large for the first few segments that are turned on and small for the last few. The following process can be used to choose LED groupings:
choose Tstart=0
choose Vstart=Vrect(Tstart)
choose Nstart=Ntotal=total number of LEDs in the string
choose some arbitrary number of LEDs Nmax for the uppermost segment
for each N=1, 2 . . . Nmax calculate:
The number of LEDs remaining Nleft=(Nstart−N) when N LEDs are switched OFF
The voltage at which this occurs, Vstop=(Nstart−N)*VLED
the time Tstop(N) at which Vrect(t)=Vstop
the instantaneous VCCS losses during the interval Tstart−Tstop
the average VCCS losses Psource_average[N] during this interval
the percentage VCCS losses Psource_average[N]/Pin during this interval
then display a column vector of each of the Nmax percentage losses
Choose the value of N=Nopt which gives the desired % Loss
This then is the number of LEDs in that segment, Nsegment=Nopt
Choose Tstart=Tstop(Nsegment)
Calculate Nstart=Nstart−Nsegment
As the algorithm starts when Isource(t) is maximum, the number of LEDs in the uppermost segment will be quite small, e.g. Nmax=3. However by the time the lower segments are reached, the number of LEDs may be quite high.
Binary-Weighted Switching places a number of constraints on both the FETs and gate drivers: FET voltage is also Binary Weighted—the MSB FET will switch half the LEDs, so must be rated at least half of the peak DC bus voltage; the LSB FET will switch only one or two LEDs so can be very low voltage.
Binary-Weighted FETs switch on and off constantly—the LSB FET has the highest switching frequency, halving for the next bit and so on down to the MSB FET, which has the lowest switching frequency.
Because of this switching behavior, the peak current rating seen by each FET during a quarter-linecycle can be calculated as follows:
The MSB FET turns OFF and stays OFF once Vrect(t) and hence Isource(t) are above h. The peak current is therefore 0.5*Isource_peak.
The next most significant bit FET turns OFF and stays OFF once Vrect(t) and hence Isource(t) are above 75%. The peak current is therefore 0.75*Isource_peak and so on, down to the LSB, which is switching furiously right up until the peak of Vrect(t) and Isource(t), so its peak current is 1.0*Isource_peak.
Assuming a 6-bit ADC, the percentage currents are:
Bit 5=50.0%
Bit 4=75.0%
Bit 3=87.5%
Bit 2=93.8%
Bit 1=96.9%
Bit 0=98.4%
As the ADC resolution increases the MSB rating stays the same and the extra LSBs get closer and closer to 100%. Therefore almost all of the FETs in a Binary-Weighted Switching scheme need to be rated for the peak current. This increases/maximizes both Qg and CDS which is unfortunate, as the switches with the highest current also switch at the highest frequency.
Finally, all but one of the FET gate drives must be completely floating, although one can sit at the low side of the DC bus.
By its very design, the switched-LED approach generates very low electromagnetic interference (EMI)—there are no switching power supplies, very little switching of LEDs occurs, and the LEDs are switched smoothly. In this respect Linear-Sequential Switching may be best—with a total of N segments, there are no more than N switching instants per quarter-cycle, as a segment is not switched ON until the voltage dropped across the current source exceeds the segment voltage, then stays ON for the rest of the quarter-cycle.
Fswitch_linear=4*Fac*Nsegments
For a circuit with a total of 8 segments, there are 8 switching events per quarter cycle, giving:
Fswitch_linear=4*50 Hz*8 segments
Fswitch_linear=1.6 kHz
Note that EMI scans typically start at 150 kHz. Even if the switching edges are fast enough to fall within the conducted EMI limits, the duty cycle is so low that the total amount of energy involved is miniscule. The amount of EMI filtering required for Linear-Sequential Switching will be negligible. However with Binary-Weighted Switching within one quarter line cycle there will be a switching instant every time the voltage dropped across the current source exceeds one LSB, which is the smallest segment voltage:
Fswitch_linear=4*Fac*Vrect_peak/Vsegment_min
Assuming a peak DC bus voltage of 165V and an LSB of 1 LED=3V this gives:
Fswitch_linear=4*50*165V/3V
Fswitch_linear=11 kHz
This is more than 10× higher the switching frequency of a similar Linear-Sequential Switching approach, but still more than an order of magnitude below the start of a conducted EMI scan. The amount of EMI filtering required for Binary Weighted Switching will be much greater than that required for Linear-Sequential Switching, but should still be negligible.
The slope of the switching edges can also be controlled by connecting a capacitor Cslew between the MOSFET Drain and Gate, as illustrated in FIG. 7. Current flowing through Cslew is proportional to the Drain voltage slew-rate, and opposes the current flowing in from the Gate resistor Rgate. Rgate and Cslew can be designed to produce a wide range of desired slew rates.
In designing an LED apparatus, it may be important to consider color control, which can be affected by temperature variations. By its very design, the switched-LED approach means different LEDs are ON for different amounts of time. Provided the LEDs are all tightly thermally coupled (e.g. using an MCPCB) they will be at the same temperature.
The VCCS current source illustrated in FIG. 3B above can be designed to have essentially any arbitrary temperature coefficient (linear or otherwise), by controlling the diode D1 voltage drop and/or using combinations of linear resistors, negative temperature coefficient (NTC) and/or positive temperature coefficient (PTC) thermistors for the current source resistors R1 & R2. Assuming the sensing elements are also tightly coupled to the LEDs, the LED Current vs Temperature curve can be controlled. Of course the LED current varies widely over a half-cycle, so the actual LED color may be somewhat harder to control. If however a segment comprises a series-parallel array of, for example, BSY and red LEDs with a shunt current-flow controller to control the ratio of BSY to red current, full color control can be attained (and the current source temperature coefficient may simplify the design).
A VCCS current source according to some embodiments may inherently provide excellent power factor and total harmonic distortion (THD). However as the input line voltage varies, so too does the LED current. An LED lamp will be designed for some specific optical output, efficiency and input power at some nominal voltage. As the RMS input voltage varies, so too does the current and hence power. The current source looks like an equivalent resistor of:
The input power and current is thus:
Pin=Vrms2/Req
Irms=Vrms/Req
A 10% drop in input voltage therefore gives a 10% drop in input current and a 19% drop in input power. Conversely a 10% rise in AC line voltage gives a 10% rise in input current and a 21% increase in input power. This will affect the color balance of the apparatus. In an apparatus including both BSY and red strings, shunt current flow controllers can correct the color balance by adjusting the ratio of BSY to Red LED current.
However, if the current source is made to be adjustable, it is possible to adjust Isource(t). Using thermistors as part of the R1-R2 network is one way of achieving this by adjusting Isource(t) as a function of temperature. If it is desired to keep the input power constant over a given voltage range, the current source must be adjusted such that Pin=Pconstant. This leads to:
Req=R3*(1+R2/R1)=Vrms2 /Pconstant
There are a number of ways that this can be done, either analog or digital. One way of doing this would be to choose:
R3=Vrms2/k
Pconstant=Vrms2/[(Vrms2/k)*(1+R2/R1)]
Pconstant=k/(1+R2/R1)
An analog approach involves measuring Vrms2 indirectly, e.g. using a fast-attack-slow-decay envelope detector, to measure the long-term average peak DC bus voltage Vpeak_avg then squaring it and using (Vpeak_avg)2 as the setpoint for a voltage controlled resistor in parallel with a much larger R3′, which now sets the minimum operating current. Making R3 adjustable is more difficult than adjusting either (or both) R1 & R2, as R3 carries the full load current. However the adjustment is linear with respect to (Vpeak_avg)2. Alternatively, it is possible to adjust either (or both) of the divider resistors R1 & R2 as follows:
Pin=(1+R2/R1)*Vrms2 /R3=Pconstant, which requires:
(1+R2/R1)*Vrms2 =Pconstant*R3,
which re-arranges to:
(R1+R2)/R1=Pconstant*R3/Vrms2
The voltage divider itself adds further nonlinearities. It may be difficult and/or unnecessary to adjust both R2 and R1. If the control circuitry is referenced to 0V, R2 can be made adjustable with little extra circuitry, again by placing a VCCS in parallel with a much larger R2′, which once again sets the minimum operating current. Re-arranging further still yields:
R2=[Pconstant*R3*R1/Vrms2 ]−R1
The required adjustment is non-linear, and may require more analog circuitry, the need for which is at least partially offset by the much lower currents involved. If a microcontroller with ADC is used to control the switch signals, this adjustment is fairly straightforward.
Typically, a microcontroller already has an ADC and a reference, with which it can measure an attenuated version of the DC bus:
VADC(t)=Vrect(t)*□1
where □1 is the attenuation factor of the DC bus sensing network. The absolute minimum sample rate required to generate Linear-Sequential Switching signals is Nsegment samples per quarter-cycle, giving:
Fsample_linear≧□4*Fac*Nsegment
where Nsegment is the total number of LED segments. So 10 segments at 60 Hz requires:
Fsample_linear≧□2400 Samples/s
The minimum ADC resolution required is given by the ratio of peak DC bus voltage to minimum segment voltage. The simulations above use 165V and 5.6V respectively, giving:
ADCresolution_linear>log2(Vrect_peak/Vsegment_min)
ADCresolution_linear>log2(165V/5.6V)>4.88 bits
This minimum value sets one LSB=minimum segment voltage. If we choose, say, to measure the minimum segment voltage to an accuracy of not less than □□(0≦□□<1), we need to add more bits:
ADCresolution_linear≧log2(Vrect_peak/(Vsegment_min)−log2(□)
If Vsegment_min is measured to within 10% (□□=0.1), then ADCresolution_linear≧log2(165V/5.6V)−log2(0.1)≧4.88+3.32=8.2 bits.
Binary-Weighted Switching requires far higher sample rates, governed by the ratio of peak DC bus voltage to LSB segment voltage:
Fsample_binary≧□4*Fac*(Vrect_peak/VLSB)
Fsample_binary≧□013200 Samples/s
The ADC resolution, as with linear switching, is given by the ratio of peak DC bus voltage to the LSB voltage multiplied by the required tolerance □. If we set 1 LSB=1 LED and □□□=0.1 (10%) this gives:
ADCresolution_binary≧log2(Vrect_peak/VLSB)−log2(□)
ADCresolution_binary≧log2(165V/3V)−log 2(0.1)≧5.78+3.32=9.1 bits
In practice an 8-bit ADC would suffice for either linear-sequential or binary-weighted switching. Modern microcontrollers make it trivial to achieve sample rates in excess of 10 kS/s, with 8- or 10-bit ADCs.
Assuming the VCCS is implemented using a microcontroller with ADC generating the switching signals, it is fairly simple (and inexpensive) to get the microcontroller to control the LED string current. First the current source is changed to a current mirror. Using a high-gain Sziklai pair as illustrated in FIG. 3D and D1 for thermal compensation ensures that the voltage across R3 is equal to the voltage across R1. If R2 is replaced with an adjustable current sink, the voltage across R1 is:
VR1=Isink*R1
But this is equal to the voltage across R3, so:
Isource=Isink*R1/R3
So the current source has been converted into a Current Mirror with gain
□1=R1/R3>>1
An adjustable current sink according to some embodiments can therefore operate at a much lower current than the current mirror, and can be both small and cheap. In practice we would increase R2 to R2′ rather than removing it, and connect the adjustable current sink in parallel with R2. This ensures the current source always operates, regardless of the current sink setpoint, which is mandatory if a non-switched pair of LEDs is used at the bottom of the string as our controller power supply.
An adjustable current sink circuit 220 according to some embodiments is illustrated in FIG. 10.
Referring to FIG. 10, R6-C1 and R7//R5-C2 form a second-order low pass filter (LPF), converting the high-frequency pulse width modulation (PWM) signal into a scaled, smoothed analog level Vbase. D5 thermally compensates the Q3 base-emitter junction, and R4 converts Vbase into Isink. If the microcontroller has a DAC, the PWM filter may be omitted.
Assuming Isource_peak=74 mA, R3=22R, and R1=2.4 k, □1=2.4 k/22=109 and Isink_peak=74 mA/109=0.678 mA
The transfer function of the adjustable current sink 220 shown above is roughly:
Isink=1.49 mA*D−0.424 mA for 0.285<=D<=1, and
Isink_max=1.07 mA
Linearity is not as important as repeatability, as look-up tables can be used for linearization. If temperature is measured (many microcontrollers have this built in) it is possible to correct for thermal drift.
Allowing some headroom for Isink allows for calibration in the factory. The duty cycle required for 0.678 mA is thus:
Dpeak=(Isink_peak+0.424 mA)/1.49 mA=0.739
The duty-cycle to Source Current transfer function is:
Isource(D)=162.7 mA*D−46.3 mA for 0.285<=D<=1
The DC Bus Voltage measured by the microcontroller is given by:
V ADC(t)=Vrect(t)*□1
N ADC=2m *V ADC(t)Nref=2m*□1*Vdc(t)/Vef
Where □1 is the attenuation factor of the DC bus sensing network and Vref is the ADC reference voltage. As with the adjustable current sink, some headroom is required for calibration. The circuit is trying to emulate a constant resistance, given by:
Req=Vrms2 /Pin
which forces the input current to be: Irms=Pin/Vrms.
Vrms is moderately difficult to calculate, but peak quantities are trivial:
Ipeak/√2=Pin/(Vpeak/∞2)
Ipeak=2*Pin/Vpeak
All the necessary data and transfer functions to calculate the duty cycle (D) are thus available. First, the peak current is calculated:
Isource_peak=2*Pin/Vrect_peak
The instantaneous current is therefore:
Isource(t)=Isource_peak*(Vrect(t)/Vrect_peak)
Isource(t)=(2*Pin/Vrect_peak)*(Vrect(t)/Vrect_peak)
Isource(t)=(2*Pin/Vrect_peak2)*Vrect(t)
Measuring Vrect_peak can be performed in software, for example using a fast-attack-slow-decay numerical envelope detector. At this point constant power operation may be implemented as follows:
Isource(t)=(2*Pconstant/Vrect_peak2)*Vrect(t)
It is then straightforward to calculate the requisite duty cycle from the current sink transfer function:
The duty cycle can be directly calculated as:
D(t)=(Isource(t)+46.3 mA)/162.7 mA
D(t)=0.285+Isource(t)/162.7 mA
The instantaneous duty cycle is thus:
D(t)=0.285+Vrect(t)*(2*Pconstant/Vrect_peak2)/162.7 mA
Note that the term (2*Pconstant/Vrect_peak2)/162.7 mA need not be calculated every sample time, reducing processor overhead if required.
If constant input power is not maintained as the line voltage varies, the calculations are much easier:
where Isource_peak is constant, and is given by
Because it does not matter if the variation in AC line voltage causes changes in string current, both power and peak voltage can be assumed to be nominal values, giving:
(2*Pin/Vrect_peak2)=(Pin_rated/Vrms_rated2)
D(t)=0.285+Vrect(t)*(Pin_rated/Vrms_rated2)/162.7 mA
D(t)=0.285+Z*Vdc(t),
where Z=(Pin_rated/Vrms_rated2)/162.7 mA is a pre-defined constant. Assuming an 8-bit ADC is used, the entire calculation can be dropped into a single 256-byte Look-Up Table:
D(t)=LOOK_UP_TABLE[NADC]
Some possible choices of switches used to switch the LEDs in and out of the string are NPN Bipolar Junction Transistor (BJT), PNP BJT, N-Channel MOSFET and/or P-Channel MOSFETs.
BJTs require a continuous supply of base current to remain conducting once switched ON, whereas MOSFETs do not—once the gate-source capacitance (and the gate-drain capacitance) has been charged, a MOSFET will stay ON without drawing any gate current (assuming of course there is no external leakage path). MOSFETs are therefore a good choice for switching LED segments. However there are a few constraints. For example, the maximum Gate-source voltage, which is device dependant, is typically +20V. The minimum Gate-source voltage, which is also device dependant, is typically −20V.
High-voltage FETs have much higher RDS_ON than low-voltage FETs, and for a given voltage rating small FETs have much higher RDS_ON than large FETs.
Small FETs have much lower gate charge (Qg) than large FETs. Also, small FETs have much lower drain-Source capacitance (CDS) than large FETs. Qg and CDS respectively are responsible for gate drive and switching losses.
For Linear-Sequential Switching, the switching frequency is very low, so Qg and CDS are not particularly important.
US446142 28 Feb 1890 10 Feb 1891 F One Half to josiaii knight
US3560728 23 Mar 1967 2 Feb 1971 Stonco Electric Products Co Floodlight and heat dissipating device
US3638042 31 Jul 1969 25 Jan 1972 Borg Warner Thyristor with added gate and fast turn-off circuit
US3655988 10 Dec 1969 11 Apr 1972 Sharp Kk Negative resistance light emitting switching devices
US3913098 23 Mar 1973 14 Oct 1975 Hayakawa Denki Kogyo Kabushiki Light emitting four layer device and improved circuitry thereof
US4504776 12 Nov 1980 12 Mar 1985 Bei Electronics, Inc. Power saving regulated light emitting diode circuit
US4798983 * 21 May 1987 17 Jan 1989 Mitsubishi Denki Kabushiki Kaisha Driving circuit for cascode BiMOS switch
US4839535 * 22 Feb 1988 13 Jun 1989 Motorola, Inc. MOS bandgap voltage reference circuit
US5059788 7 Mar 1990 22 Oct 1991 Nec Corporation Optical logic device with PNPN detection and laser diode output
US5059890 6 Dec 1989 22 Oct 1991 Fujitsu Limited Constant current source circuit
US5125675 30 Jul 1990 30 Jun 1992 Engelbrecht Jan C Trolley
US5138541 6 Jan 1992 11 Aug 1992 Nafa-Light Kurt Maurer Lamp with ventilated housing
US5334916 27 May 1992 2 Aug 1994 Mitsubishi Kasei Corporation Apparatus and method for LED emission spectrum control
US5357120 14 Jul 1992 18 Oct 1994 Hitachi Ltd. Compound semiconductor device and electric power converting apparatus using such device
US5397938 * 28 Oct 1993 14 Mar 1995 Siemens Aktiengesellschaft Current mode logic switching stage
US5467049 15 Jul 1993 14 Nov 1995 Hitachi, Ltd. Solid-state switch
US5504448 1 Aug 1994 2 Apr 1996 Motorola, Inc. Current limit sense circuit and method for controlling a transistor
US5528467 25 Sep 1995 18 Jun 1996 Wang Chi Industrial Co., Ltd. Head light structure of a car
US5646760 12 Apr 1995 8 Jul 1997 Interuniversitair Micro-Elektronica Centrum Vzw Differential pair of optical thyristors used as an optoelectronic transceiver
US6079852 5 Jun 1997 27 Jun 2000 Piaa Corporation Auxiliary light
US6137235 29 Jul 1999 24 Oct 2000 Patent-Treuhand-Gesellschaft Fur Elektrische Gluehlampen Mbh Low-resistance bipolar bridge circuit
US6153980 4 Nov 1999 28 Nov 2000 Philips Electronics North America Corporation LED array having an active shunt arrangement
US6264354 21 Jul 2000 24 Jul 2001 Kamal Motilal Supplemental automotive lighting
US6323597 10 Jul 2000 27 Nov 2001 Jlj, Inc. Thermistor shunt for series wired light string
US6411155 8 Jan 2001 25 Jun 2002 Sgs-Thomson Microelectronics S.A. Power integrated circuit
US6501630 17 Dec 1999 31 Dec 2002 Koninklijke Philips Electronics N.V. Bi-directional ESD diode structure
US6556067 12 Jun 2001 29 Apr 2003 Linfinity Microelectronics Charge pump regulator with load current control
US6697130 28 Dec 2001 24 Feb 2004 Visteon Global Technologies, Inc. Flexible led backlighting circuit
US6755550 6 Feb 2003 29 Jun 2004 Amy Lackey Recessed illuminated tile light
US6784622 5 Dec 2001 31 Aug 2004 Lutron Electronics Company, Inc. Single switch electronic dimming ballast
US7108238 24 Jul 2001 19 Sep 2006 Regent Lighting Corporation Outdoor light mounting bracket
US7109664 16 Dec 2003 19 Sep 2006 Tsu-Yeh Wu LED light with blaze-like radiance effect
US7271545 7 Oct 2005 18 Sep 2007 Delta Electronics, Inc. Ballast and igniter for a lamp having larger storage capacitor than charge pump capacitor
US7291983 5 Mar 2007 6 Nov 2007 Delta Electronics, Inc. Ballast and igniter for a lamp having larger storage capacitor than charge pump capacitor
US7307391 9 Feb 2006 11 Dec 2007 Led Smart Inc. LED lighting system
US7408308 27 Apr 2006 5 Aug 2008 Sharp Kabushiki Kaisha LED drive circuit, LED lighting device, and backlight
US7427838 22 Apr 2005 23 Sep 2008 Nec Corporation Light source controlling circuit and portable electronic apparatus
US7513639 29 Sep 2006 7 Apr 2009 Pyroswift Holding Co., Limited LED illumination apparatus
US7535180 4 Apr 2005 19 May 2009 Cree, Inc. Semiconductor light emitting circuits including light emitting diodes and four layer semiconductor shunt devices
US7614767 9 Jun 2006 10 Nov 2009 Abl Ip Holding Llc Networked architectural lighting with customizable color accents
US7628513 18 Jul 2007 8 Dec 2009 Primo Lite Co., Ltd. Led lamp structure
US7649326 27 Mar 2007 19 Jan 2010 Texas Instruments Incorporated Highly efficient series string LED driver with individual LED control
US7656371 26 Jul 2004 2 Feb 2010 Nichia Corporation Light emitting apparatus, LED lighting, LED light emitting apparatus, and control method of light emitting apparatus
US7677767 1 Apr 2008 16 Mar 2010 Wen-Long Chyn LED lamp having higher efficiency
US7679292 26 Oct 2006 16 Mar 2010 Fiber Optic Designs, Inc. LED lights with matched AC voltage using rectified circuitry
US7780318 1 Feb 2008 24 Aug 2010 Fu Zhun Precision Industry (Shen Zhen) Co., Ltd. Flood lamp assembly having a reinforced bracket for supporting a weight thereof
US7812553 28 Mar 2007 12 Oct 2010 Samsung Electronics Co., Ltd. LED lighting device and method for controlling the same based on temperature changes
US7862201 20 Jul 2006 4 Jan 2011 Tbt Asset Management International Limited Fluorescent lamp for lighting applications
US7862214 23 Oct 2007 4 Jan 2011 Cree, Inc. Lighting devices and methods of installing light engine housings and/or trim elements in lighting device housings
US7914902 6 Nov 2007 29 Mar 2011 Jiing Tung Tec. Metal Co., Ltd. Thermal module
US7994725 6 Nov 2008 9 Aug 2011 Osram Sylvania Inc. Floating switch controlling LED array segment
US8008845 30 Jul 2009 30 Aug 2011 Cree, Inc. Lighting device which includes one or more solid state light emitting device
US8157422 16 Mar 2011 17 Apr 2012 Lg Electronics Inc. Lighting apparatus
US8174201 10 Jul 2009 8 May 2012 Sheng-Hann Lee Self-oscillating transformerless electronic ballast
US8235555 18 May 2009 7 Aug 2012 Electraled, Inc. Multiple use LED light fixture
US8519630 28 Apr 2011 27 Aug 2013 Analog Integrations Corporation Driving circuit capable of enhancing energy conversion efficiency and driving method thereof
US8970131 15 Feb 2013 3 Mar 2015 Cree, Inc. Solid state lighting apparatuses and related methods
US20020027453 3 Apr 2001 7 Mar 2002 Kulhalli Suhas R. Amplifying signals in switched capacitor environments
US20020063534 14 May 2001 30 May 2002 Samsung Electro-Mechanics Co., Ltd Inverter for LCD backlight
US20020097095 15 May 2001 25 Jul 2002 Samsung Electronics Co., Ltd. Temperature compensation circuit for a power amplifier
US20040036418 21 Aug 2002 26 Feb 2004 Rooke Alan Michael Closed loop current control circuit and method thereof
US20040208009 12 Jun 2002 21 Oct 2004 Mardon Paul Francis Lighting unit with improved cooling
US20040208809 24 Nov 2003 21 Oct 2004 D Alesandro Raymond J Method of removing SO3 from flue gases
US20050007164 5 Aug 2004 13 Jan 2005 Callahan Michael J. Driver circuit having a slew rate control system with improved linear ramp generator including ground
US20050057179 26 Aug 2004 17 Mar 2005 Osram Sylvania Inc. Driver circuit for LED vehicle lamp
US20050111222 21 Nov 2003 26 May 2005 Olsson Mark S. Thru-hull light
US20050128752 20 Oct 2004 16 Jun 2005 Ewington Christopher D. Lighting module
US20050140282 29 Jun 2004 30 Jun 2005 Lg.Philips Lcd Co., Ltd. Organic electroluminescent display device and method of fabricating the same
US20050242742 30 Apr 2004 3 Nov 2005 Cheang Tak M Light emitting diode based light system with a redundant light source
US20050254234 17 May 2004 17 Nov 2005 Kuo-Tsai Wang LED flashlight
US20050280376 5 Apr 2005 22 Dec 2005 Stacoswitch, Inc. Transistorized, voltage-controlled dimming circuit
US20060018664 21 Sep 2005 26 Jan 2006 Levinson Frank H Optoelectronic device capable of participating in in-band traffic
US20060153511 18 Sep 2003 13 Jul 2006 Franklin James B Light emitting device
US20060176411 24 Mar 2005 10 Aug 2006 Norimasa Furukawa Constant current driver, back light source and color liquid crystal display
US20060221609 14 Jun 2006 5 Oct 2006 Ryan Patrick H Jr Lighting strip
US20060238465 25 Apr 2006 26 Oct 2006 Sanyo Epson Imaging Devices Corporation Led driving circuit, illuminating device, and electro-optical device
US20060244396 16 Nov 2005 2 Nov 2006 Constantin Bucur Serial powering of an LED string
US20070018594 28 Sep 2006 25 Jan 2007 Jlj. Inc. Holiday light string devices
US20070096661 28 Oct 2005 3 May 2007 David Allen Decorative lighting string with stacked rectification
US20070108843 17 Nov 2005 17 May 2007 Preston Nigel A Series connected power supply for semiconductor-based vehicle lighting systems
US20070115662 6 Mar 2006 24 May 2007 Cree, Inc. Adaptive adjustment of light output of solid state lighting panels
US20070182346 2 Feb 2007 9 Aug 2007 Exclara Inc. Current regulator for multimode operation of solid state lighting
US20070195023 10 Jan 2007 23 Aug 2007 Samsung Electronics Co., Ltd. Light emitting apparatus and control method thereof
US20070215027 16 Mar 2007 20 Sep 2007 Macdonald Ian Two piece view port and light housing with swivel light
US20070236920 31 Mar 2006 11 Oct 2007 Snyder Mark W Flashlight providing thermal protection for electronic elements thereof
US20070257623 27 Mar 2007 8 Nov 2007 Texas Instruments, Incorporated Highly efficient series string led driver with individual led control
US20070257999 5 Jul 2006 8 Nov 2007 Novatek Microelectronics Corp. Variable-gain amplifier circuit and method of changing gain amplifier path
US20070262724 14 May 2007 15 Nov 2007 Alexander Mednik Shunting type pwm dimming circuit for individually controlling brightness of series connected leds operated at constant current and method therefor
US20070267978 18 May 2007 22 Nov 2007 Exclara Inc. Digitally controlled current regulator for high power solid state lighting
US20070273299 25 Feb 2005 29 Nov 2007 Michael Miskin AC light emitting diode and AC LED drive methods and apparatus
US20080054281 20 Dec 2006 6 Mar 2008 Nadarajah Narendran High-efficient light engines using light emitting diodes
US20080062070 13 Sep 2006 13 Mar 2008 Honeywell International Inc. Led brightness compensation system and method
US20080089071 12 Oct 2006 17 Apr 2008 Chin-Wen Wang Lamp structure with adjustable projection angle
US20080094000 * 29 Aug 2007 24 Apr 2008 Kenji Yamamoto Device and method for driving led
US20080105887 20 Jun 2006 8 May 2008 Nadarajah Narendran Package Design for Producing White Light With Short-Wavelength Leds and Down-Conversion Materials
US20080122376 9 Nov 2007 29 May 2008 Philips Solid-State Lighting Solutions Methods and apparatus for controlling series-connected leds
US20080128718 28 Nov 2007 5 Jun 2008 Nichia Corporation Light emitting device
US20080129220 13 Aug 2007 5 Jun 2008 Exclara Inc. System and Method for Driving LED
US20080130283 1 Apr 2005 5 Jun 2008 Che-Min Chang Lamp and lamp string
US20080150440 20 Feb 2007 26 Jun 2008 Gemmy Industries Corporation LED light string with guaranteed conduction
US20080157688 28 Sep 2007 3 Jul 2008 Gibboney James W Light String of LEDS
US20080203946 15 Feb 2008 28 Aug 2008 Koito Manufacturing Co., Ltd. Light emitting apparatus
US20080211415 21 Dec 2007 4 Sep 2008 Altamura Steven J Resistive bypass for series lighting circuit
US20080215279 11 Dec 2007 4 Sep 2008 Tir Technology Lp Luminaire control system and method
US20080252197 13 Apr 2007 16 Oct 2008 Intematix Corporation Color temperature tunable white light source
US20080258628 17 Apr 2007 23 Oct 2008 Cree, Inc. Light Emitting Diode Emergency Lighting Methods and Apparatus
US20090015759 3 Jul 2008 15 Jan 2009 Nec Lcd Technologies, Ltd Light emission control circuit, light emission control method, flat illuminating device, and liquid crystal display device having the same device
US20090039791 1 Jul 2008 12 Feb 2009 Steve Jones Entryway lighting system
US20090059582 27 Aug 2008 5 Mar 2009 Texas Instruments Incorporated Heat Sinks for Cooling LEDS in Projectors
US20090079355 21 Sep 2007 26 Mar 2009 Exclara Inc. Digital Driver Apparatus, Method and System for Solid State Lighting
US20090079357 29 Oct 2007 26 Mar 2009 Exclara Inc. Regulation of Wavelength Shift and Perceived Color of Solid State Lighting with Intensity Variation
US20090079358 29 Oct 2007 26 Mar 2009 Exclara Inc. Regulation of Wavelength Shift and Perceived Color of Solid State Lighting with Temperature Variation
US20090079359 29 Oct 2007 26 Mar 2009 Exclara Inc. System and Method for Regulation of Solid State Lighting
US20090079360 29 Oct 2007 26 Mar 2009 Exclara Inc. System and Method for Regulation of Solid State Lighting
US20090079362 29 Oct 2007 26 Mar 2009 Exclara Inc. Regulation of Wavelength Shift and Perceived Color of Solid State Lighting with Intensity and Temperature Variation
US20090094000 8 Apr 2008 9 Apr 2009 Balachander Krishnamurthy System and method for profiling resource constraints of web servers
US20090140630 17 Mar 2006 4 Jun 2009 Mitsubishi Chemical Corporation Light-emitting device, white light-emitting device, illuminator, and image display
US20090147517 1 Feb 2008 11 Jun 2009 Fu Zhun Precision Industry (Shen Zhen) Co., Ltd. Led recessed lamp with screws fixing a recessed fixture thereof
US20090251934 6 Apr 2008 8 Oct 2009 Exclara Inc. Apparatus, System and Method for Cascaded Power Conversion
US20100026187 18 Oct 2007 4 Feb 2010 William Kelly Luminaire drive circuit
US20100026208 29 Jul 2008 4 Feb 2010 Exclara Inc. Apparatus, System and Method for Cascaded Power Conversion
US20100045187 5 Jul 2009 25 Feb 2010 Exclara Inc. System and Method for Driving LED
US20100045210 25 Aug 2009 25 Feb 2010 Suresh Hariharan Power Factor Correction in and Dimming of Solid State Lighting Devices
US20100060175 9 Sep 2008 11 Mar 2010 Exclara Inc. Apparatus, Method and System for Providing Power to Solid State Lighting
US20100060181 4 Sep 2009 11 Mar 2010 Seoul Semiconductor Co., Ltd. Ac led dimmer and dimming method thereby
US20100067227 16 Nov 2009 18 Mar 2010 Budike Lothar E S LED light pod with modular optics and heat dissipation structure
US20100079076 30 Sep 2008 1 Apr 2010 Chu-Cheng Chang Led light string without additional resistors
US20100079262 26 Sep 2008 1 Apr 2010 Albeo Technologies, Inc. Systems And Methods For Conveying Information Using A Control Signal Referenced To Alternating Current (AC) Power
US20100090604 11 Aug 2009 15 Apr 2010 Yasuhiro Maruyama Led drive circuit, led illumination component, led illumination device, and led illumination system
US20100123403 16 Nov 2009 20 May 2010 Reed William G Electronic control to regulate power for solid-state lighting and methods thereof
US20100134018 24 Nov 2009 3 Jun 2010 Microsemi Corp. - Analog Mixed Signal Group Ltd. Led string driver with light intensity responsive to input voltage
US20100135016 10 Apr 2007 3 Jun 2010 Miyoji Ishibashi Lamp unit
US20100141159 6 Mar 2009 10 Jun 2010 Green Solution Technology Inc. Led driving circuit and controller with temperature compensation thereof
US20100156763 17 Nov 2009 24 Jun 2010 Hyun Lee Organic electroluminescent display device including heat-radiating means
US20100177509 25 Aug 2009 15 Jul 2010 Cree Led Lighting Solutions, Inc. Lighting device
US20100194274 16 Jul 2008 5 Aug 2010 Nxp B.V. Light emitting diode (led) arrangement with bypass driving
US20100231135 17 Jul 2009 16 Sep 2010 Bridgelux,Inc. Reconfigurable LED Array and Use in Lighting System
US20100246177 26 Mar 2009 30 Sep 2010 Cree Led Lighting Solutions, Inc. Lighting device and method of cooling lighting device
US20100277084 16 Jul 2010 4 Nov 2010 Seoul Opto Device Co., Ltd. Light emitting device for ac power operation
US20100308738 22 Mar 2010 9 Dec 2010 Exclara Inc. Apparatus, Method and System for Providing AC Line Power to Lighting Devices
US20100308739 4 Jun 2009 9 Dec 2010 Exclara Inc. Apparatus, Method and System for Providing AC Line Power to Lighting Devices
US20110025217 24 Jun 2010 3 Feb 2011 Intersil Americas Inc. Inrush current limiter for an led driver
US20110031894 4 Aug 2009 10 Feb 2011 Cree Led Lighting Solutions, Inc. Lighting device having first, second and third groups of solid state light emitters, and lighting arrangement
US20110057571 4 May 2009 10 Mar 2011 Koninklijke Philips Electronics N.V. Device and method for controlling the color point of an led light source
US20110068701 12 Feb 2010 24 Mar 2011 Cree Led Lighting Solutions, Inc. Solid state lighting apparatus with compensation bypass circuits and methods of operation thereof
US20110074265 25 Sep 2009 31 Mar 2011 Cree Led Lighting Solutions, Inc. Lighting device with one or more removable heat sink elements
US20110074289 7 Jun 2010 31 Mar 2011 Van De Ven Antony Paul Lighting Devices Including Thermally Conductive Housings and Related Structures
US20110075411 25 Sep 2009 31 Mar 2011 Cree Led Lighting Solutions, Inc. Light engines for lighting devices
US20110075414 19 Nov 2009 31 Mar 2011 Cree Led Lighting Solutions, Inc. Light engines for lighting devices
US20110075422 25 Sep 2009 31 Mar 2011 Cree Led Lighting Solutions, Inc. Lighting devices comprising solid state light emitters
US20110075423 25 Sep 2009 31 Mar 2011 Cree Led Lighting Solutions, Inc. Lighting device with position-retaining element
US20110109228 6 Nov 2009 12 May 2011 Tsutomu Shimomura System and method for lighting power and control system
US20110115394 18 Aug 2010 19 May 2011 Exclara Inc. System and Method for Regulation of Solid State Lighting
US20110115411 5 Nov 2010 19 May 2011 Exclara Inc. Current Regulator for Multimode Operation of Solid State Lighting
US20110198984 12 Feb 2010 18 Aug 2011 Cree Led Lighting Solutions, Inc. Lighting devices that comprise one or more solid state light emitters
US20110211351 8 Feb 2011 1 Sep 2011 Cree, Inc. Lighting devices that comprise one or more solid state light emitters
US20110227484 21 Apr 2011 22 Sep 2011 Active-Semi, Inc AC LED lamp involving an LED string having separately shortable sections
US20110227485 19 Mar 2010 22 Sep 2011 Active-Semi, Inc. AC LED lamp involving an LED string having separately shortable sections
US20110227489 4 May 2010 22 Sep 2011 Active-Semi, Inc. Reduced flicker AC LED lamp with separately shortable sections of an LED string
US20110227490 31 Jul 2010 22 Sep 2011 Active-Semi, Inc. AC LED lamp involving an LED string having separately shortable sections
US20110273102 7 May 2010 10 Nov 2011 Van De Ven Antony P Ac driven solid state lighting apparatus with led string including switched segments
US20120091920 11 Apr 2011 19 Apr 2012 Long Yang LED Light Source with Direct AC Drive
US20120099321 10 Nov 2011 26 Apr 2012 Bridgelux, Inc. Ac led array module for street light applications
US20120176826 8 Jan 2012 12 Jul 2012 Braxton Engineering, Inc. Source and multiple loads regulator
US20120194073 28 Apr 2011 2 Aug 2012 Jing-Chyi Wang Driving circuit capable of enhancing energy conversion efficiency and driving method thereof
US20120201025 8 Mar 2011 9 Aug 2012 Cree, Inc. Lighting apparatus providing increased luminous flux while maintaining color point and cri
US20120306375 3 Jun 2011 6 Dec 2012 Cree, Inc. Systems and methods for controlling solid state lighting devices and lighting apparatus incorporating such systems and/or methods
US20130026923 27 Jan 2012 31 Jan 2013 Cree, Inc. Solid state lighting apparatus and methods of forming
US20130077299 7 Nov 2012 28 Mar 2013 Cree, Inc. High voltage array light emitting diode (led) devices, fixtures and methods
US20130207559 17 Dec 2012 15 Aug 2013 Lumenetix, Inc. Linear bypass electrical circuit for driving led strings
US20130278157 16 Dec 2011 24 Oct 2013 Koninklijke Philips Electronics N.V. Device and method for controlling current to solid state lighting circuit
USD188916 5 Oct 1959 27 Sep 1960 Mcgraw Luminaire
USD207867 21 Jul 1966 6 Jun 1967 General Electric Company Light fixture
USD384430 7 Aug 1996 30 Sep 1997 light projector
USD400280 3 Oct 1997 27 Oct 1998 Mercury vapor light
USD418620 9 Sep 1998 4 Jan 2000 Regent Lighting Corporation Outdoor light
USD425024 10 Sep 1998 16 May 2000 Dal Partnership Compact fluorescent bulb socket
USD437439 30 Apr 1999 6 Feb 2001 Shih-Chuan Tang Floodlight
USD490181 20 Aug 2002 18 May 2004 Zumtobel Staff Gmbh & Co. Kg Ceiling lighting fixture
USD544979 25 Sep 2006 19 Jun 2007 Itc Incorporated Light fixture
USD557853 10 Feb 2007 18 Dec 2007 Eml Technologies Llc Yard light with dark sky shade
USD558374 10 Feb 2007 25 Dec 2007 Eml Technologies Llc Yard light
USD610291 21 Nov 2008 16 Feb 2010 Toshiba Lighting & Technology Corporation Recessed lighting fixture
USD618376 29 Jun 2009 22 Jun 2010 Zumtobel Staff Gmbh & Co. Kg Lighting fixture
USD625038 15 Jan 2009 5 Oct 2010 Fawoo Technology Co., Ltd. Explosion-resistant street light
USD627502 23 Nov 2009 16 Nov 2010 Foxconn Technology Co., Ltd. LED lamp
USD627911 7 Dec 2009 23 Nov 2010 Foxconn Technology Co., Ltd. LED lamp
USD633099 25 Sep 2009 22 Feb 2011 Cree, Inc. Light engine for a lighting device
USD636921 15 Jan 2010 26 Apr 2011 Cree, Inc. Lighting device
USD636922 25 Feb 2010 26 Apr 2011 Toshiba Lighting & Technology Corporation Recessed lighting fixture
USD638160 25 Sep 2009 17 May 2011 Cree, Inc. Lighting device
USD646011 27 Jul 2010 27 Sep 2011 Hamid Rashidi LED light with baffle trim
CN1575623A 16 Oct 2002 2 Feb 2005 皇家飞利浦电子股份有限公司 LED control apparatus
CN101137261A 29 Aug 2007 5 Mar 2008 安华高科技Ecbu Ip（新加坡）私人有限公司 Device and method for driving LED
CN101668373A 29 Sep 2009 10 Mar 2010 李云霄 LED light source driving circuit supplied by AC power
CN101772245A 12 Mar 2010 7 Jul 2010 陈林 LED lighting device capable of automatically adapting to power supply voltage
CN101827481A 22 May 2010 8 Sep 2010 李云霄 Alternating-current power supply LED light source drive circuit with segmented conversion input
EP1594348A2 19 Apr 2005 9 Nov 2005 Nec Corporation Light source controlling circuit and portable electronic apparatus
EP1881259A1 3 Jul 2007 23 Jan 2008 Liquidleds Lighting Co., Ltd. High power LED lamp with heat dissipation enhancement
JP2003273404A Title not available
JP2006103404A Title not available
JP2006332022A Title not available
JP2008125339A Title not available
JP2008205357A Title not available
JP2008544569A Title not available
JP2009016280A Title not available
JP2009049010A Title not available
JP2010092776A Title not available
JPH04196359A Title not available
JPS59113768A Title not available
WO2006007388A1 16 Jun 2005 19 Jan 2006 3M Innovative Properties Company Solid state light device
WO2006018604A1 8 Aug 2005 23 Feb 2006 E-Light Limited Lighting system power adaptor
WO2008036873A2 21 Sep 2007 27 Mar 2008 Cree Led Lighting Solutions, Inc. Lighting assemblies, methods of installing same, and methods of replacing lights
WO2008051957A2 23 Oct 2007 2 May 2008 Cree Led Lighting Solutions, Inc. Lighting devices and methods of installing light engine housings and/or trim elements in lighting device housings
WO2008129504A1 21 Apr 2008 30 Oct 2008 Philips Intellectual Property & Standards Gmbh Led string driver with shift register and level shifter
WO2009013676A2 16 Jul 2008 29 Jan 2009 Nxp B.V. Led arrangement with bypass driving
1 "Assist Recommends . . . LED Life for General Lighting: Definition of Life", vol. 1, Issue 1, Feb. 2005.
2 "Bright Tomorrow Lighting Competition (L Prize(TM) )", May 28, 2008, Document No. 08NT006643.
3 "Bright Tomorrow Lighting Competition (L Prize™ )", May 28, 2008, Document No. 08NT006643.
4 "Energy Star® Program Requirements for Solid State Lighting Luminaires, Eligibility Criteria-Version 1.1", Final: Dec. 19, 2008.
5 Application Note: CLD-APO6.006, entitled Cree® XLamp® XR Family & 4550 LED Reliability, published at cree,com/xlamp, Sep. 2008.
6 Bulborama, Lighting Terms Reference and Glossary, http://www.bulborama.com/store/lightingreferenceglossary-13.html, 6 pages.
7 Chinese First Office Action Corresponding to Chinese Patent Application No. 201180022813.5; Date of Issuance: Feb. 25, 2014; Foreign Text, 16 Pages; English Translation Thereof, 5 Pages.
8 Chinese Office Action Corresponding to Chinese Patent Application No. 2010-80053242.7;. Date of Issue: Nov. 27, 2013, Foreign Text, 16 Pages, English Translation, 35 Pages.
9 Chinese Office Action Corresponding to Chinese Patent Application No. 201280044038.8; Date of Notification: Dec. 12, 2014; Foreign Text, 16 Pages, English Translation Thereof, 7 Pages.
10 Chinese Office Action Corresponding to Chinese Patent Application No. 201280054168.X; Date of Notification: Feb. 12, 2015; Foreign Text, 8 Pages, English Translation Thereof, 8 Pages.
11 Chinese Office Action issues on Nov. 3, 2014 for corresponding Application No. 201180004266.8 (4 pages).
12 Chinese Second Office Action Corresponding to Chinese Patent Application No. 201080053889.X; Date of Issue: Dec. 17, 2014; Foreign Text, 9 Pages, English Translation Thereof, 15 Pages.
13 DuPont "DuPontSMDiffuse Light Reflector", Publication K-20044, May 2008, 2 pages.
14 European Extended Search Report Corresponding to European Application No. 11777867; Dated; May 13, 2014; 7 Pages.
15 EXM020, Multi-Channel 160W LED Driver, Rev. 2,0 Nov. 2010, 13 pages, www.exclara.com.
16 EXM055, 14.8W Dimmable LED Ballast, Rev. 0.7, Mar. 11, 2011, 10 pages, www.exclara.com.
17 EXM057, 14.5W Dimmable LED Ballast, Rev. 0.5, Mar. 11, 2011, 8 pages, www.exclara.com.
18 Extended European Search Report corresponding to European Patent Application No. 10819249; Date of Mailing: Mar. 27, 2014, 8 pages.
19 Furukawa Electric Co., Ltd,, Data Sheet, "New Material for Illuminated Panels Microcellular Reflective Sheet MCPET", updated Apr. 8, 2008, 2 pages.
20 Global Patent Literature Text Search Corresponding to PCT Application No. PCT/US2011/38995; Date of Search: Sep. 8, 2011; 7 pages.
21 Illuminating Engineering Society Standard Lm-80-08, entitled "IES Approved Method for Measuring Lumen Maintenance of LED Light Sources", Sep. 22, 2008, ISBN No, 978-0-87995-227-3.
22 International Preliminary Report Corresponding to International Application No. PCT/US2011/033736; Date of Mailing: Nov. 22, 2012; 8 Pages.
23 International Preliminary Report on Patentability corresponding to International Application No, PCT/US2010/029897; Date of Mailing: Apr. 27, 2011; 14 pages.
24 International Preliminary Report on Patentability Corresponding to International Application No. PCT/US2010/048567; Date of Mailing: Apr. 5, 2012; 10 pages.
25 International Search Report and The Written Opinion of the International Searching Authority Corresponding to International Application No. PCT/US2011/033736; Date of Mailing: Jul. 7, 2011; 10 pages.
26 International Search Report And The Written Opinion of the International Searching Authority Corresponding to International Application No. PCT/US2011/038995; Date of Mailing: Sep. 16, 2011; 9 pages.
27 International Search Report and Written Opinion for Corresponding PCT Application No. PCT/US2014/068534, mailed Mar. 4, 2015 (22 pages).
28 International Search Report and Written Opinion for PCT Application No. PCT/US12/47643 mailed on Oct. 25, 2012,.
29 International Search Report Corresponding to International Application No, PCT/US12/54869; Date of Mailing: Nov. 23, 2012; 10 Pages.
30 International Search Report Corresponding to International Application No. PCT/US11/54846; Date of Mailing: Jan. 23, 2012; 13 pages.
31 International Search Report Corresponding to International Application No. PCT/US12/54888; Date of Mailing: Nov. 23, 2012; 12 Pages.
32 International Search Report Corresponding to International Application No. PCT/US2010/048567; Dated: Oct. 29, 2010.
33 International Search Report Corresponding to International Application No. PCT/US2010/049581; Date of Mailing: Nov. 23, 2010; 3 pages.
34 Japanese Office Action Corresponding to Japanese Patent Application No. 2012-530920; Mailing Date: May 28, 2014; Foreign Text, 3 Pages; English Translation Thereof, 2 Pages.
35 Japanese Office Action Corresponding to Japanese Patent Application No. 2013-509109; Dispatch Date: Sep. 17, 2013; Foreign Text, 2 Pages, English Translation, 3 Pages.
36 Kim et al. "Strongly Enhanced Phosphor Efficiency in GaInN White Light-Emitting Diodes Using Remote Phosphor Configuration and Diffuse Reflector Cup" Japanese Journal of Applied Physics 44(21):L649-L651 (2005).
37 LEDs Magazine, Press Release May 23, 2007, "Furukawa America Debuts MCPET Reflective Sheets to Improve Clarity, Efficiency of Lighting Fixtures", downloaded Jun. 25, 2009 from http://www.ledsmagazine.com/press/15145, 2 pages.
38 MCPET-Microcellular Reflective Sheet Properties, http://www.trocellen.com, downloaded Jun. 25, 2009, 2 pages.
39 Notification Concerning Transmittal of Copy of International Preliminary Report on Patentability, Application No. PCT/US2010/048225, Date of Mailing: Feb. 27, 2014, 9 Pages.
40 Notification Concerning Transmittal of Copy of International Preliminary Report on Patentability, Application No. PCT/US2012/054869, Date of Mailing: Mar. 27, 2014, 8 pages.
41 Notification Concerning Transmittal of Copy of International Preliminary Report on Patentability, Application No. PCT/US2012/054888, Date of Mailing: Mar. 27, 2014, 10 pages.
42 Notification of transmittal of the international search report and the written opinion of the international searching authority, or declaration, PCT/US2010/029897, Jun. 23, 2010.
43 Notification of Transmittal of the International Search Report and the Written Opinion of the International Searching Authority, or the Declaration, International Search Report, and Written Opinion of the International Searching Authority, PCT International Application No.PCT/US2006/011820, Aug. 7, 2006.
44 Notification of Transmittal of the International Search Report and the Written Opinion of the International Searching Authority, or the Declaration; International Search Report; and Written Opinion of the International Searching Authority, PCT Application No. PCT/US2010/037608, Jul. 30, 2010.
45 Notification of Transmittal of the International Search Report and the Written Opinion of the International Searching Authority, or the Declaration; International Search Report; Written Opinion of the International Searching Authority; Corresponding to International Application No. PCT/US2010/048225; Dated: Nov. 4, 2010; 11 pages.
46 Philips Lumileds, Technology White Paper: "Understanding power LED lifetime analysis", downloaded from http://www.philipslumileds.com/pdfs/WP12.pdf, Document No. WP12, Last Modified May 22, 2007.
47 Search Report issued for corresponding Taiwan Patent Application No. 099131743, mailed May 20, 2015, 1 page.
48 Sutardja, P., "Design for High Quality and Low Cost SSL with Power Factor Correction", Marvell Semiconductor Inc. Jul. 2011, 16 pages.
49 U.S. Appl. No. 11/854,744, filed Sep. 13, 2007, Myers.
50 U.S. Appl. No. 12/328,115, filed Dec. 4, 2008, Chobot.
51 U.S. Appl. No. 12/328,144, filed Dec. 4, 2008, Chobot.
52 U.S. Appl. No. 60/844,325, filed Sep. 13, 2006, Myers.
US9474118 * 20 Nov 2014 18 Oct 2016 Microchip Technology Inc. Cascode-type dimming switch using a bipolar junction transistor for driving a string of light emitting diodes
US9763296 * 15 Jun 2016 12 Sep 2017 Infineon Technologies Ag Multifunction DC to DC driver
US20150145425 * 20 Nov 2014 28 May 2015 Microchip Technology Inc. Cascode-type Dimming Switch Using A Bipolar Junction Transistor For Driving A String Of Light Emitting Diodes
International Classification H05B39/00, H05B37/00, H05B33/08, H05B41/00
Cooperative Classification H05B33/0818, H05B33/0896, H05B33/0821, H05B33/0809, H05B33/083