Source: https://patents.google.com/patent/US8358644B2/en
Timestamp: 2020-04-04 06:49:24
Document Index: 733008959

Matched Legal Cases: ['art 900', 'art 900', 'art 1000', 'art 1000', 'art 1100', 'art 1100', 'art 1200', 'art 1300']

US8358644B2 - Methods and apparatus for generating synchronization/pilot sequences for embedding in wireless signals - Google Patents
Methods and apparatus for generating synchronization/pilot sequences for embedding in wireless signals Download PDF
US8358644B2
US8358644B2 US12/051,535 US5153508A US8358644B2 US 8358644 B2 US8358644 B2 US 8358644B2 US 5153508 A US5153508 A US 5153508A US 8358644 B2 US8358644 B2 US 8358644B2
US12/051,535
US20090003308A1 (en
2008-03-19 Application filed by General Dynamics C4 Systems Inc filed Critical General Dynamics C4 Systems Inc
2008-03-19 Priority to US12/051,535 priority patent/US8358644B2/en
2008-03-19 Assigned to GENERAL DYNAMICS C4 SYSTEMS, INC. reassignment GENERAL DYNAMICS C4 SYSTEMS, INC. ASSIGNMENT OF ASSIGNORS INTEREST (SEE DOCUMENT FOR DETAILS). Assignors: BAXLEY, ROBERT JOHN, KLEIDER, JOHN ERIC
2009-01-01 Publication of US20090003308A1 publication Critical patent/US20090003308A1/en
2010-03-17 Priority claimed from US12/725,985 external-priority patent/US8379752B2/en
2013-01-22 Publication of US8358644B2 publication Critical patent/US8358644B2/en
238000000819 phase cycle Methods 0 claims description 14
The inventive subject matter generally relates to methods and apparatus for wirelessly communicating signals, and more particularly to methods and apparatus for generating synchronization/pilot sequences and wirelessly communicating signals in which the synchronization/pilot sequences are embedded.
Orthogonal frequency division multiplexing (OFDM) is a modulation method used in traditional, high-speed wireless networks. However, waveforms generated using traditional OFDM techniques exhibit noise-like properties, and thus OFDM waveforms tend to suffer from relatively large peak-to-average ratios (PARs), which in turn may lead to significant distortion noise and low power efficiency in peak-limited channels. In addition, under relatively harsh channel conditions, transmitted OFDM signals tend to incur significant timing offsets and carrier frequency offsets. Because traditional OFDM techniques tend not to be robust under harsh channel conditions, significant timing offsets may result in inter-block interference, and significant carrier frequency offsets may result in inter-carrier interference. Both of these forms of interference are detrimental to the bit error rates of received signals.
In addition, the channel estimate naturally has some error, when compared with actual channel conditions. Traditional OFDM transmission methods may experience an increase in channel estimation errors on the receiver side, which may result from non-linear amplification, by a power amplifier device on the transmitter side, of transmit information sequences having higher than desired PARs. Such non-linear transmission may cause significant out-of-band interference (i.e., interference outside the signal bandwidth, such as in the adjacent channels and/or other user channels), and also may induce undesired in-band interference, which adds distortion the transmitted information bits and also to the channel training information. Furthermore, improper synthesis of the channel training information may lead to further channel estimation errors at the receiver. Thus, non-linear amplification of high peak-to-average power ratio signals and improper channel training information design may, in the receiver, result in unacceptably high channel estimation errors and excessively high bit error rates.
In some OFDM systems, prior to transmission, an information-bearing OFDM sequence is combined with a synchronization/pilot sequence, which provides spectral efficiency improvements over preamble-based synchronization approaches. Traditional sequences include, for example, Pseudorandom Number (PN) sequences, Gold codes, Kasami codes, and m-sequences. Although traditional synchronization/pilot sequences are appropriate for some situations, they do not provide for adequate system performance in other situations. For example, although traditional sequences are designed to perform relatively well for synchronization purposes, they are not designed to provide low PAR or flat frequency response in conjunction with optimal channel estimation by the receiver. Essentially, in an OFDM system, traditional synchronization/pilot sequences do not provide for adequate system performance in channel environments in which significant timing offsets, carrier frequency offsets, and multi-path fading effects simultaneously are present.
Accordingly, what are needed are methods and apparatus for generating synchronization/pilot sequences that provide for adequate system performance even under harsh channel conditions, and methods and apparatus for generating and transmitting OFDM waveforms in which these synchronization/pilot sequences are embedded. More particularly, what are needed are methods and apparatus for generating synchronization/pilot sequences that have synchronization properties that are at least as good as the synchronization properties of traditional sequences, and that also exhibit low-PAR properties and a flat frequency response. Other features and characteristics of the inventive subject matter will become apparent from the subsequent detailed description and the appended claims, taken in conjunction with the accompanying drawings and this background.
FIG. 6 is a flowchart of a method for generating a set of synchronization/pilot sequences (SPS), in accordance with an example embodiment;
FIG. 7 is a flowchart of a method for generating and transmitting wireless signals that include embedded SPS, in accordance with an example embodiment;
FIG. 8 is a flowchart of a method for receiving and processing wireless signals that include embedded SPS, in accordance with an example embodiment;
FIG. 9 is a chart plotting relationships between an embedding factor representing a ratio of SPS power to signal power, and a ratio of pilot power to the total SPS power for various ratios of pilot to data subcarriers using three example embodiments;
FIG. 10 is a chart plotting bit error rate (BER) performance that may be achieved using two example embodiments;
FIG. 11 is a chart indicating acquisition performance for a system employing an embodiment versus a system that uses traditional binary sequences;
FIG. 12 is a chart indicating channel estimation performance for a system using SPS generated according to an embodiment and a system that uses traditional binary sequences;
FIG. 13 is a chart illustrating a complementary cumulative distribution function (CCDF) of the peak-to-average ratio (PAR) for signals produced using two different embedding sequences and various embedding factors, in accordance with example embodiments;
FIG. 14 is a plot of the convergence characteristics of the PAR for five lowest-PAR SPS, which were generated according to an embodiment; and
FIG. 15 is a scatter plot of the PAR and mean square error (MSE) after 600 iterations of an embodiment of a method for SPS generation for 150 different initial conditions.
Embodiments include methods and apparatus for wirelessly communicating orthogonal frequency division multiplexing (OFDM) signals between wireless communication devices. A signal communicated according to an embodiment includes an embedded synchronization/pilot sequence (SPS) selected from a set of SPS that is accessible to a transmitter and a receiver, as will be described in detail below. As used herein, the acronym SPS may indicate a single synchronization/pilot sequence or multiple synchronization/pilot sequences. For example, the term “set of SPS” means a set of multiple synchronization/pilot sequences.
Embodiments include methods for generating SPS that may have one or more significant advantages over traditional techniques. More particularly, embodiments of methods for generating sets of SPS that may jointly provide for robust synchronization, low peak-to-average ratios (PARs), and accurate channel estimation, among other things. SPS generated according to various embodiments may have synchronization properties (e.g., compensation for timing offsets and frequency offsets) that are comparable to and potentially better than for synchronization/pilot sequences generated using traditional techniques. In addition, low PARs may be achieved because embodiments may enable a transmitter's power amplifier to be operated more efficiently. Improved channel estimation may be achieved because SPS may be generated, according to various embodiments, using arbitrary frequency domain profiles while achieving a relatively flat frequency response over all frequencies of interest to the signal. In addition to the above advantages, embodiments may result in increased link ranges, because signals may be transmitted using lower power, and correspondingly may be less susceptible to detection. Conversely, embodiments may result in higher link margins, as it may be possible to transmit higher-power signals using a given power amplifier, when compared to traditional techniques that utilize non-constant envelope transmissions. In addition, for battery-powered apparatus, improved battery life may be achieved, because the power amplifier may be operated at a higher efficiency than using traditional techniques. Embodiments may lead to higher power amplifier efficiency, as a signal that includes an SPS generated according to an embodiment may require substantially less back-off than a system that utilizes traditional synchronization/pilot sequences.
Embodiments may be utilized in various types of systems. For example, embodiments may be utilized in multi-carrier communication systems, single-carrier communication systems, and spread spectrum communication systems. Although embodiments discussed in detail below may pertain to a multi-carrier communication system, it is to be understood that other embodiments may apply to other types of systems, as well. Further, as will be explained in more detail below, embodiments include methods for generating a single SPS or for generating a set of SPS. Embodiments include embedded synchronization methods and apparatus that are employed in a selected mapping (SLM) system, and accordingly such embodiments may be referred to herein as SPS-SLM. It is to be understood that other embodiments may apply to systems in which selected mapping techniques are not employed.
FIG. 1 is a simplified block diagram of a multi-carrier communication system 100 that includes multiple wireless communication devices 102, 104 that communicate over a wireless communication channel 106, in accordance with an example embodiment. Multi-carrier communication system 100 may be, for example but not by way of limitation, a currently existing or future multi-carrier based, ultra-wideband system, an OFDM multiple access system, a multi-carrier code division multiple access (MC-CDMA) system, a digital video broadcasting system, a WiMax (long range broadband wireless) system, a wireless local area network (WiLAN) system (e.g., an 802.11a system), and/or a number of other types of multi-carrier communication systems.
Functionality of transmitter 110 and receiver 114, are described only briefly in conjunction with the description of FIG. 1. More detailed descriptions of the details of various transmitter and receiver embodiments are included later, in conjunction with FIGS. 3-6. Briefly, transmitter 110 is adapted to apply multiple phase shifts to an input data symbol 118, and to combine a plurality of SPS, which are selected from a set of SPS accessible to transmitter 110, with the phase shifted input data in order to produce a plurality of candidate signals. Embodiments of methods for generating SPS will be described in more detail later in conjunction with FIG. 6. First and second scaling factors may be applied to the input data symbol and to the plurality of SPS, respectively, prior to combining the phase shifted input data and the plurality of SPS. As will be discussed in detail later, the scaling factors affect the relative signal power allocated to the phase shifted input data and the SPS with which they are combined. Transmitter 110 also is adapted to determine PARs for at least some of the candidate signals, and to identify a selected candidate signal based on the PARs (e.g., the selected candidate signal may be the candidate signal with the lowest PAR). Transmitter 110 also is adapted to transmit a wireless signal 120 representing the selected candidate signal over the wireless communication channel 106.
where fPA(·) represents the power amplifier input-to-output characteristic, which may be assumed to be time-invariant (although the input-to-output characteristic may be time-variant, in other embodiments), h[τ] represents multi-path fading component 202, y[n−n0] represents a transmitted signal, y[n], subjected to a TO component 204, e−j2πε/N represents a CFO component 206, η[n] represents an additive noise component 208, and * is the convolution operator.
ρ = ∑ k ⁢  S ⁡ [ k ]  2 ∑ k ⁢  Y ⁡ [ k ]  2 .
Each of the plurality of phase shifters 304 includes computational apparatus adapted to apply a different phase shift 326,
ⅇ jϕ k ( d ) ,
to the scaled input data symbol 324, in order to produce a plurality of phase shifted input data signals
1 - ρ ⁢ X k ( d ) ⁢ ⅇ jϕ k ( d ) ,
where D is a value referred to herein as a candidate number quantity, d is an index referred to herein as a relational index, and d∈ε{1, 2, . . . , D}. The candidate number quantity, D, may be selected as any integer number from 1 to 16, in an embodiment, although the candidate number quantity may be a larger number, in other embodiments. In a particular embodiment, the candidate number quantity is selected as an integer number between 3 and 10. In an embodiment, the number of phase shifted input data signals 328 produced equals the candidate number quantity D, although the number of phase shifted input data signals 328 may be different, in other embodiments. The different phase shifts 326 may be represented within entries of a table of phase shift values, in an embodiment, and the relational index, d, may be used an index into the phase shift value table, among other things. Accordingly, the phase shift value table may have D entries, in an embodiment, although the phase shift value table may have more or fewer entries in other embodiments. Transmitter 300 also is adapted to obtain a plurality of SPS 332, Sk (d), each of which represents a unique synchronization/pilot sequence. In an embodiment, the plurality of SPS 332 may be obtained from a table of SPS, which is accessible to or stored in transmitter 300, and which includes one or more sets of pre-generated SPS, each of which may be referenced by a unique index (referred to below as an SLM index). Each SPS 332 in the transmitter's SPS table is represented in the frequency domain, in an embodiment. Embodiments of methods for generating sets of SPS will be described in more detail later in conjunction with FIG. 6.
Y k ( d ) = ρ ⁢ S k ( d ) + 1 - ρ ⁢ X k ( d ) ⁢ ⅇ j ⁢ ⁢ ϕ k ( d ) . ( Equation ⁢ ⁢ 2 )
where x(d)[n]=IDFT{Xkejφ h (d) }, s(d)[n]=IDFT{Sk (d)}, and n∈{0, 1, . . . , N−1}. In an embodiment, an efficient algorithm for computing the inverse discrete Fourier transform (IDFT) may be implemented, such as an inverse fast Fourier transform (IFFT), for example.
FIG. 4 is an example of a frequency-domain representation of a transmit signal 400, in accordance with an example embodiment. Axis 402 represents frequency, and axis 404 represents signal power (e.g., in dB). An embodiment is implemented in a pilot symbol assisted modulation (PSAM) OFDM system with null edge sub-carriers. Embodiments may, alternatively, be implemented in other types of systems, although such other systems are not discussed in detail herein. Within frequency band 406, the transmit signal 400 includes a data component 408 and an SPS component 410, which are modulated onto a plurality, N, of sub-carriers. More particularly, the subcarriers occupied by the data component 408, Xk, of the transmit signal 400, may be decomposed into several non-overlapping parts: 1) data-bearing subcarriers 412, which may be denoted by a set of indices Kd; pilot subcarriers 414, which may be denoted by a set of indices Kp; and null edge subcarriers 416, which may be denoted by the set of indices Kn. In an embodiment, Xk∉K d =0, so that the data component 408 of the transmit signal 400 only contains energy in data-bearing subcarriers 412. Null edge subcarriers 416 may be constrained, in an embodiment, to zero to limit the amount of spectral regrowth that may encroach on neighboring channels. Pilot signals 420 may be defined as part of the SPS (e.g., SPS 332, FIG. 3 and SPS 538, FIG. 5). The subcarriers occupied by the SPS component 410 of the transmit signal 400, may be decomposed into the same non-overlapping parts as the data component 408, or more particularly: 1) synchronization subcarriers 412, Kd; pilot subcarriers 414, Kp; and null edge subcarriers 416, Kn. These signal segmentations may be summarized as Table 1, below:
TABLE 1 k ε Kd k ε Kp k ε Kn Xk ≠0 =0 =0 Sk ≠0 ≠0 =0 Yk ≠0 ≠0 =0
Although fifty-one total sub-carriers (e.g., N=51), thirty-eight data-bearing subcarriers 412, five pilot subcarriers 414, and eight null edge sub-carriers 416 are illustrated in FIG. 4, these numbers are used for example purposes only, and more or fewer total sub-carriers, data-bearing subcarriers 412, pilot subcarriers 414, and/or null edge sub-carriers 416 may be utilized, in other embodiments.
SPS component 410 includes synchronization sequence information 422 conveyed within synchronization subcarriers 412 (e.g., data-bearing subcarriers 412), and a plurality of pilot signals 420 conveyed within pilot subcarriers 414, in an embodiment. Because at least some of the synchronization subcarriers 412 occupied by the SPS component 410 are the same as the data-bearing subcarriers 412 occupied by the data component 408, the synchronization sequence information 422 (and thus the SPS component 410) may be considered to be “embedded” within the data component 408. Embodiments of methods for generating SPS components 410 will be described in more detail later in conjunction with FIG. 6.
Pilot signals 420 (or pilot subcarriers 414) have constant power and are evenly spaced (e.g., a same number of data-bearing subcarriers 414 exist between consecutive pilot subcarriers 414), in an embodiment. In alternate embodiments, the positioning and spacing of pilot signals 420 may be different from that illustrated in FIG. 4. In a particular embodiment, the pilot subcarrier 414 spacing is less than the number of null edge subcarriers (e.g., N/|Kp|>|Kn|), which may result in a relatively small channel estimation mean square error (MSE). The amount of power in pilot subcarriers 414 may be quantified according to the equation:
β = ∑ k ∈ K p ⁢  S ⁡ [ k ]  2 ∑ k ∈ K p ⋃ K d ⁢  S ⁡ [ k ]  2 , ( Equation ⁢ ⁢ 6 )
where fPA(·) represents the power amplifier input-to-output characteristic, which may be assumed to be time-invariant (although the input-to-output characteristic may be time-variant, in other embodiments), h[τ] represents a multi-path fading component of the channel, y({tilde over (d)})[n−n0] represents the transmitted signal, y({tilde over (d)})[n], subjected to a TO component, e−j2πε/N represents a CFO component, η[n] represents an additive noise component, * is the convolution operator, and {tilde over (d)} is the SLM index. It is to be noted that any carrier phase shift present between the transmitter and receiver is assumed to be included in the phase of the channel at the receiver.
In an embodiment, the candidate synchronization sequences 538 include time-domain versions of the same synchronization/pilot sequences (e.g., SPS 332, FIG. 3) as were combined by the transmitter (e.g., transmitter 300, FIG. 3) with the phase shifted input data (e.g., phase shifted input data 328, FIG. 3). As mentioned previously, both the transmitter (e.g., transmitter 300) and the receiver 500 each may have knowledge of the candidate SPS by each having access to substantively identical tables of SPS, although the transmitter's SPS table may include SPS represented in the frequency domain, and the receiver's SPS table may include the same SPS represented in the time domain, in an embodiment.
where (·)* is the conjugate operation.
d ~ ^ = arg ⁢ ⁢ max d ⁢  r ( d ) ⁡ [ u ]  . ( Equation ⁢ ⁢ ( 11 )
Accordingly, the SLM index estimate 540 corresponds to the conjugate correlation output 536 that represents a highest correlation peak. Unlike traditional methods, embodiments include blind phase sequence detection criterion (e.g., no side information representing the SLM index is transmitted) in order to determine the SLM index estimate 540, and the SLM index estimate 540 is determined based on the conjugate correlations between the received signal 534 and the candidate synchronization sequences 538. Correct detection of {tilde over (d)} may depend on the magnitude of the peaks of |r(d)[u]| for d≠{tilde over (d)}, also referred to herein as “spurious correlation peaks.” When the spurious correlation peaks all are less than the peak in |r(d)[u]|, {tilde over (d)} may be correctly detected (e.g., {tilde over ({circumflex over (d)}={tilde over (d)}). In an embodiment, and as will be described in more detail later, the candidate SPS 538 are designed so that the spurious correlation peaks are low. In a particular embodiment, the candidate SPS 538 are designed so that:
└maxCC{s (d) [n],s (d) [n−u]}┘<th self, (Equation 12)
Accordingly, the coarse timing offset estimate 542 is determined based on the maximum of the {tilde over ({circumflex over (d)} th conjugate correlation output. Assuming that {tilde over ({circumflex over (d)}={tilde over (d)}, the coarse timing offset estimate should be determined (or “detected”) correctly as long as |r({tilde over (d)})[n0]|>r({tilde over (d)})[n] form n≠n0.
ɛ ^ = angle ( r ( d ~ ^ ) ⁡ [ n ^ 0 ] ) . ( Equation ⁢ ⁢ 14 )
Once the coarse timing and carrier frequency offsets are removed, the coarsely-corrected signal 550 may be transformed to the frequency domain by time domain-to-frequency domain (TD-to-FD) transformer 514, which includes computational apparatus adapted to perform a time domain-to-frequency domain transformation on the corrected signal 550, in order to produce a frequency-domain, coarsely-corrected signal 553. The time domain-to-frequency domain transformation may include a Fourier transform (FT) or, more particularly, a fast Fourier transform (FFT), in various embodiments, although other types of time domain-to-frequency domain transformations may be performed in other embodiments.
E ⁡ [  X ^ k - X k  2 ❘ H k ] ≈ σ 2  H k  2 · ( ( 1 - β ) ⁢  K p  β ⁡ ( 1 - ρ ) ⁢  K d  +  K p  β ⁢ ⁢ ρ ⁢  K d  + 1 1 - ρ ) . ( Equation ⁢ ⁢ 19 )
As Equation 19 indicates, the MSE is dependent on the ratio of pilot to data subcarriers |Kp|/|Kd|. Also, the minimizing the pilot subcarrier power is achieved by setting β=1 when perfect synchronization is assumed. However, in an embodiment, β is selected such that β<1, in order to achieve desired synchronization performance.
SPS removal element 518 includes computational apparatus adapted to receive the equalized combined signal 554, and to remove the scaled SPS 562 corresponding to the SLM index estimate 540 from the equalized combined signal 554 (e.g., to combine −√{square root over (ρ)}sk ({tilde over ({circumflex over (d)}) with the equalized combined signal 554) in order to produce an estimated, phase shifted data signal 564. In an embodiment, the scaled SPS 562 may be obtained by retrieving the SPS sk ({tilde over ({circumflex over (d)}) corresponding to the SLM index estimate 540 from a table of SPS, which is accessible to or stored in receiver 500, and by applying the scaling factor √{square root over (ρ)} to the retrieved SPS. The SPS table includes one or more pre-generated sets of SPS, where each SPS in a set may be referenced by an SLM index. Each SPS in the receiver's SPS table is represented in the frequency domain, in an embodiment. Embodiments of methods for generating sets of SPS will be described in more detail later in conjunction with FIG. 6. Scaling element 520 is adapted to apply a scaling factor to the estimated, phase shifted data signal 564, in order to produce a scaled, phase shifted data signal 566, which has a peak amplitude approximately equal to that of the original input data, X[n]. Phase shift element 522 includes computational apparatus adapted to phase shift the scaled, phase shifted data signal 566 by a phase shift value 568 corresponding to the SLM index estimate 540 (e.g., to shift the scaled, phase shifted data signal 566 by
ⅇ - j ⁢ ⁢ ϕ ( d ~ ^ ) ⁢
The remaining signal is demodulated in order to produce the output data symbol 580, {circumflex over (X)}k[n]. When the SLM index estimate 540 represents a correctly-detected SLM index (e.g., an SLM index corresponding to the selected signal 346, FIG. 3, identified at the transmitter 300), then blind phase sequence detection has been robustly performed by receiver 500, and the output data symbol 580 reflects an accurate estimate of the input data symbol (e.g., input data symbol 320, FIG. 3).
As discussed in detail above, both a transmitter (e.g., transmitter 300, FIG. 3) and a receiver (e.g., receiver 500, FIG. 5) have access to at least one set of pre-generated SPS. Embodiments include methods for generating sets of SPS that result in significant PAR reductions and that have comparable synchronization and channel estimation properties, when compared with traditional methods.
According to an embodiment, generating an SPS that results in significant PAR reductions are achieved when IDFT{Sk (d)}=sd[n] has low PAR. In this case, the combined sequence y(d)[n]=√{square root over (ρ)}s(d)[n]+√{square root over (1−ρ)}x(d)[n] may, on average, have lower PAR than x(d)[n]. The magnitude of PAR reduction depends on the value of the embedding factor, ρ, where larger PAR reductions may be achieved when ρ has a relatively large value (e.g., when ρ>0.6), and smaller PAR reductions may be achieved when ρ has a relatively small value (e.g., when ρ<0.6). In an embodiment, a value for ρ may derived assuming perfect acquisition by minimizing the maximum symbol estimate in E└|xd|2┘, or MSEx, where MSEx=E[|{circumflex over (X)}d−Xd|2]. Using various embodiments, an SPS may be generated to have a PAR <0.5 dB, although embodiments may be implemented in which an SPS has a PAR ≧0.5 dB, as well.
In addition, SPS generated according to an embodiment may have excellent synchronization properties. As discussed previously, synchronization includes estimating the SLM index, {tilde over (d)}, for the transmitted signal, estimating a coarse timing offset, n0, and estimating a coarse CFO, {circumflex over (ε)}. An estimation of which phase sequence index, {tilde over (d)}, was transmitted may be made via criterion specified in Equation 11, above. From Equation 11, it is apparent that correct estimation of {tilde over (d)} depends on the peaks of |r(d)[u]| for d≠{tilde over (d)} (i.e., spurious correlation peaks). When the spurious correlation peaks all are less than the peak in |r(d)[u]|, {tilde over (d)} will be correctly detected. Accordingly, in an embodiment, sets of SPS are generated so that spurious correlation peaks are low, when compared with the peak in |r(d)[u]|.
Assuming that x(d)[n] is independent of s(d)[n], the peaks in |r(d)[u]| when d≠{tilde over (d)} are dictated by the peaks of the conjugate correlation CC{sd[n],sq[n]} for d≠q. In an embodiment, a set of SPS is generated so that maxu,d≠qCC{sd[n],sq[n−u]} is minimized using an optimization procedure. In an alternate embodiment, a set of SPS may be generated more simply according to the following equation:
[ max u , d ≠ q ⁢ C ⁢ ⁢ C ⁢ { s ( d ) ⁡ [ n ] , s ( q ) ⁡ [ n - u ] } < th cross , ( Equation ⁢ ⁢ 21 )
As discussed previously, once {tilde over (d)} is detected, a coarse timing offset estimate (e.g., coarse timing offset estimate 542), {circumflex over (n)}0, may be determined according to Equation 13, above. As Equation 13 indicates, the coarse timing offset estimate is determined based on the maximum of the {tilde over ({circumflex over (d)} th conjugate correlation output. Although the channel estimator (e.g., channel estimator/corrector 516, FIG. 5) may compensate for differences |n0−{circumflex over (n)}0|≦Lcp−Lh+1, where Lh is the length of the channel and Lcp is the length of the cyclic prefix, the SPS are generated, in an embodiment, to minimize this difference. According to Equation 13, above, no is determined based on the maximum of the {tilde over ({circumflex over (d)} th conjugate correlation output, and it may be assumed that {tilde over ({circumflex over (d)}={tilde over (d)}, n0 may be detected correctly as long as |r({tilde over (d)})[n0]|>r({tilde over (d)})[n] for all n≠n0. In an embodiment, a set of SPS is generated so that maxd,u≠n 0 CC{s(d)[n],s(d)[n−u]} is minimized. In an alternate embodiment, a set of SPS may be generated more simply according to the equation:
[ max d , u ≠ n 0 ⁢ C ⁢ ⁢ C ⁢ { s ( d ) ⁡ [ n ] , s ( d ) ⁡ [ n - u ] } < th self , ( Equation ⁢ ⁢ 22 )
FIG. 6 is a flowchart of a method for generating a set of SPS, according to an example embodiment. The set of SPS may be used, for example, as a set of pre-generated SPS that are accessed by a transmitter (e.g., transmitter 300, FIG. 3) and a receiver (e.g., receiver 500, FIG. 5), as discussed previously in conjunction with FIGS. 3 and 5. A set of SPS may be represented, for example, as {s(d)[n]}d=1 D, where D is the number of SPS in the set, and d is a relational index that may be correlated, for example, with an SLM index or an SLM index estimate (e.g., SLM index estimate 540, FIG. 5). In an embodiment, the number of SPS in a set, D, is an integer having a value between 2 and 10, although a set of SPS may have more SPS, in other embodiments.
In an embodiment, each SPS in the set is generated by performing multiple iterations of a time-frequency projection (e.g., a Projection onto Convex Sets (POCS) algorithm), or an iterative convergence process based on PAR results and/or mean square error properties. In an embodiment, the number of iterations, I, is an integer having a value between about 100 and 300, although a smaller or larger number of iterations may be performed, in alternate embodiments. The flowchart of FIG. 6 includes an inner loop, which represents an iteration of a time-frequency projection (e.g., the inner loop is performed I times) in order to generate a single SPS, and an outer loop, which is performed S times in order to generate a set of S candidate SPS. Further steps of the method reduce the number of candidate SPS to a set of D SPS (e.g., D<S), as will be explained in detail below.
The method may begin, in block 602, by initializing an inner loop counter, i, and an outer loop counter, s. Inner loop counter, i, indicates a current time-frequency projection iteration being performed for the SPS being generated, and accordingly may be referred to as an iteration counter. In an embodiment, the inner loop counter is initialized to a value of 1 and is incremented by 1 for each iteration being performed up to a value of I, although the inner loop counter may be initialized to some other value, and/or may be incremented differently, or may be a decrementing counter, in alternate embodiments.
The group of blocks 603 are executed in order to generate a single candidate SPS. As mentioned previously, generation of a candidate SPS includes using an iterative time-frequency projection algorithm (e.g., a POCS algorithm). For each candidate SPS, the algorithm is initialized using different initial conditions (e.g., a different random phase) from the other candidate SPS that are generated. Accordingly, generation of a candidate SPS may begin, in block 604, by initializing the algorithm by generating an initial, random phase, constant modulus phase sequence, to which a pre-determined power profile is applied. In an embodiment, the random phase is determined by choosing a uniformly generated random phase between 0 and 2π radians or between −π and π radians. In an embodiment, the actual generation of the phase may be performed using a uniform random number generator between 0 and 1 inclusive (e.g., denoting as ru), and applying the randomly generated number to a complex phasor of form exp(j2πru). The power profile that is applied to the phase sequence is determined by computing a desired amplitude for each subcarrier, where the amplitudes for the subcarriers are computed to provide a lowest symbol MSE performance at the receiver, in an embodiment. The power profile is applied by multiplying the desired amplitudes by the subcarrier value generated in the inner loop of FIG. 6 (e.g., in block 604), in order to produce a power-adjusted phase sequence. In an embodiment, the applied power profile is the same for all SPS generated in the set. The length of the frequency-domain sequence is in a range of 32 to 124 values, in an embodiment, although shorter or longer sequences may be generated, in alternate embodiments.
In block 606, a time-domain transformation is performed on the initial, power-adjusted phase sequence to produce a time-domain sequence. The time domain-to-frequency domain transformation may include a Fourier transform or, more particularly, a discrete Fourier transform (DFT), in various embodiments, although other types of time domain-to-frequency domain transformations may be performed in other embodiments.
In block 608, amplitudes of the time-domain sequence are set to unity while maintaining phases of the time-domain sequence to produce an amplitude-adjusted time-domain sequence. More particularly, given that the time domain version may not be unity in amplitude, the sequence is converted to magnitude and phase (i.e., polar form). The magnitude of the converted sequence is set so that the amplitude is unity, while the original phase is retained. The converted sequence is then converted back to real and imaginary (i.e., rectangular form) to produce the amplitude-adjusted time-domain sequence.
In block 610, a frequency-domain transformation is performed on the amplitude-adjusted time-domain sequence to produce an adjusted frequency-domain sequence. The frequency domain-to-time domain transformation may include an inverse Fourier transform or, more particularly, an inverse discrete Fourier transform, in various embodiments, although other types of frequency domain-to-time domain transformations may be performed in other embodiments. In block 612, the power profile is applied to the adjusted frequency-domain sequence while maintaining phases of the adjusted frequency-domain sequence in order to produce an adjusted candidate sequence.
In block 614, a determination is made whether the last iteration has been performed for the candidate SPS being generated (e.g., whether i=I). If not, then the inner loop counter is incremented (e.g., by 1), in block 616, and the method iterates as shown by repeating blocks 606-614 at least an additional time using the adjusted candidate sequence.
When the last iteration has been performed, then the then-current adjusted candidate sequence represents a completed version of a candidate SPS. A determination may then be made, in block 618, whether the last candidate SPS has been generated in the set of candidate SPS (e.g., whether s=S). If not, then the outer loop counter is incremented (e.g., by 1), in block 620, and the method iterates as shown by repeating blocks 604-618 until the last candidate SPS has been generated.
When the last SPS has been generated, a subset of D candidate SPS may be selected, via blocks 622, 624, 626, and 628, which will represent the set of SPS being generated according to the method of FIG. 6. In block 622, certain candidate SPS that were generated via blocks 604-620 may be eliminated from the set of candidate SPS. In an embodiment, candidate SPS are eliminated that do not meet a PAR selection criteria (e.g., a selection criteria based on PAR). For example, in a particular embodiment, the PAR selection criteria may be a PAR threshold, thPAR, and those candidate SPS having a PAR value that is greater than (or is equal to or greater than) the PAR threshold may be eliminated from the set of candidate SPS. In other words, when (max|s(s)[n]|)>thPAR for a candidate SPS, the candidate SPS may be eliminated. A PAR threshold may have a value in a range between about 0 dB and about 2.0 dB, in an embodiment, although the PAR threshold may be smaller or greater than the values within the above-given range, in other embodiments. In other embodiments, an inclusion process (rather than an exclusion process) may be performed, in which those candidate SPS having a PAR value that is less than a PAR threshold may be allowed to remain within the set of candidate SPS. In still another embodiment, block 622 may be excluded altogether from the SPS set generation method.
In block 624, a plurality of correlations are performed among the candidate SPS (e.g., the candidate SPS that remain after block 622) to generate a plurality of correlation values. In a particular embodiment, performing the correlations includes performing a plurality of cross-correlations among the candidate SPS to generate a plurality of cross-correlation results, and also performing a plurality of auto-correlations among the candidate SPS to generate a plurality of auto-correlation results. In an embodiment, P2 cross-correlations are performed, where P is a number of candidate SPS being correlated. In other words, each candidate SPS is correlated with each other candidate SPS in order to generate P2 cross-correlation results. Each cross-correlation result represents a maximum peak for the cross-correlation, and may be represented by
max τ ≠ 0 ⁢  s ( s ) ⁡ [ n ] ⊗ s ( s ) ⁡ [ n + τ ]  .
In block 626, which may be performed earlier in other embodiments, a plurality of permutations of sets of candidate SPS are determined. In an embodiment, each permutation includes a different combination of D SPS selected from the set of candidate SPS. Permutations may be determined for each possible combination of SPS, although in other embodiments, a smaller number of permutations may be determined.
In block 628, a permutation is identified, from the plurality of permutations, as a selected set of SPS (e.g., the end result of the SPS set generation method). In a particular embodiment, the identified permutation corresponds to the permutation having a smallest maximum max-correlation value (e.g., the set that gives the smallest maximum cross-correlations within the set and/or the smallest secondary peak in the auto-correlations in the set). In an embodiment, identifying the selected permutation from the plurality of permutations includes identifying a permutation that corresponds to a maximum cross correlation threshold and/or a maximum secondary peak of the auto-correlations. In general, a low secondary peak indicates a more definitive result for each auto-correlation, and the same is true for a maximum cross-correlation (e.g., one would desire the maximum peak of the cross-correlation to be as small as possible). After identifying the permutation, the method may then end.
FIG. 7 is a flowchart of a method for generating and transmitting wireless signals that include embedded SPS, in accordance with an example embodiment. Embodiments of the method are only briefly discussed in conjunction with FIG. 7, as various details and alternate embodiments were discussed in more detail above.
Referring also to FIG. 3, the method may begin, in block 702, when a transmitter (e.g., transmitter 300) receives (e.g., by data/scaling factor combiner 302) an input data symbol (e.g., input data symbol 320). In block 704, a first scaling factor (e.g., first scaling factor 322) may be applied to the input data symbol, in order to produce a scaled input data symbol (e.g., scaled input data symbol 324). As discussed previously, the first scaling factor may have a value of √{square root over (1−ρ)}, where ρ is an embedding factor having an absolute value between 0 and 1. In other embodiments, the first scaling factor may have a different value. In block 706, various different phase shifts (e.g., phase shifts 326) are applied (e.g., by phase shifters 304) to the scaled input data symbol, in order to produce a plurality of phase shifted input data signals (e.g., phase shifted input data signals 328).
In block 708, a plurality of SPS (e.g., SPS 332) are obtained (e.g., a plurality of SPS generated according to an embodiment), and a second scaling factor (e.g., second scaling factor 330) is applied to the plurality of SPS in order to produce a plurality of scaled SPS (e.g., scaled SPS 334). As discussed previously, the second scaling factor may have a value of √{square root over (ρ)}, in an embodiment, although the second scaling factor may have a different value, in other embodiments. Preferably, but not essentially, the second scaling factor has an inverse mathematical relationship with the first scaling factor (e.g., by varying the value of the embedding factor, as the second scaling factor value increases, the first scaling factor value decreases, and vice versa).
In block 710, each one of the plurality of phase shifted input data signals is combined (e.g., by data/SPS combiners 308) with one of the scaled SPS in order to produce a plurality of combined signals (e.g., combined signals 340). In block 712, a frequency domain-to-time domain transformation is performed (e.g., by FD-to-TD transformers 310) on each of the combined signals, in order to produce a plurality of candidate signals (e.g., candidate signals 342).
In block 714, peak-to-average ratios (PARs) are determined (e.g., by signal selector 312) for some or all of the candidate signals, and based on the peak-to-average ratios, a selected signal (e.g., selected signal 346) is identified from the candidate signals. As discussed previously, the selected signal may be identified as the candidate signal with the lowest PAR, in an embodiment. In block 716, the selected signal is up-converted (e.g., by up-converter 314), amplified (e.g., by power amplifier 316), and transmitted over the channel (e.g., channel 106, FIG. 1). Although not illustrated or discussed herein, those of skill in the art would realize that various other processes for conditioning, filtering, and/or processing the various signals prior to transmission also may be performed at various stages within the process of generating and transmitting the selected signal. Upon transmitting the selected signal, the method may then end.
FIG. 8 is a flowchart of a method for receiving and processing wireless signals that include embedded SPS, in accordance with an example embodiment. Embodiments of the method are only briefly discussed in conjunction with FIG. 8, as various details and alternate embodiments were discussed in more detail above.
Referring also to FIG. 5, the method may begin, in block 802, when a receiver (e.g., receiver 500) receives (e.g., via antenna 502) a wireless RF signal (e.g., RF signal 530) from the channel. The received RF signal includes a channel-affected version of a data signal combined with an SPS, as discussed in conjunction with the description of embodiments of the transmitter (e.g., transmitter 300, FIG. 3), and embodiments of the method for generating and transmitting the wireless RF signal (e.g., FIG. 7). In block 804, the received RF signal is down-converted and digitized (e.g., by down-converter 532), in order to produce an IF or baseband received signal (e.g., received signal 534).
In block 806, the received signal is correlated (e.g., by correlators 506) with a plurality of SPS (e.g., SPS 538 generated according to an embodiment) to produce a plurality of conjugate correlation outputs (e.g., conjugate correlation outputs 536). In block 808, an SLM index estimate (e.g., SLM index estimate 540) is determined (e.g., by peak detector 508), based on the conjugate correlation outputs.
In block 810, coarse offset estimates (e.g., coarse TO and coarse CFO) may be determined (e.g., by coarse offset estimator 510) based on the conjugate correlation output corresponding to the SLM index estimate. In block 812, corrections are made (e.g., by offset corrector 512) for the coarse timing and carrier frequency offsets in the received signal, in order to produce a coarsely-corrected signal (e.g., coarsely-corrected signal 550). In block 814, fine estimated offsets (e.g., fine CFO, fine TO, and/or phase offset) may be determined (e.g., by fine offset estimator 515) based on the coarsely-corrected signal, and in block 816, additional corrections may be made (e.g., by offset corrector 512 in the time domain or by a frequency-domain offset corrector), in order to produce a finely-corrected signal (e.g., finely-corrected signal 551).
In block 818, channel effects are estimated (e.g., by channel estimator/corrector 516) from a frequency-domain version of the finely-corrected signal. The finely-corrected signal is then equalized based on the estimated channel effects, in order to produce an equalized combined signal (e.g., equalized combined signal 554).
In block 820, a scaled SPS (e.g., scaled SPS 562) corresponding to the SLM index estimate is removed (e.g., by SPS removal element 518) from the equalized combined signal, in order to produce an estimated, phase shifted data signal (e.g., estimated, phase shifted data signal 564), which may be scaled (e.g., by scaling element 520). A phase shift operation is performed (e.g., by phase shift element 522), in block 822, which includes phase shifting the scaled, phase shifted data signal by a phase shift value corresponding to the SLM index estimate. This operation results in the production of an output data symbol (e.g., output data symbol 580), which reflects and estimate of the input data symbol (e.g., input data symbol 320, FIG. 3). The method may then end.
FIGS. 9-15 indicate potential simulated results for systems that employ various example embodiments. For example, FIG. 9 is a chart 900 plotting relationships between an embedding factor representing a ratio, ρ, of synchronization/pilot sequence power to signal power, and a ratio, β, of pilot power to the total synchronization/pilot sequence power for various ratios of pilot to data subcarriers, or |Kp|/|Kd| using three example SPS-SLM embodiments. Chart 900 includes a β axis 902 and a ρ axis 904. As described above in conjunction with Equation 19, the MSE is dependent on the ratio of pilot to data subcarriers. Accordingly, adjusting the ratio of pilot to data subcarriers affects the MSE.
Plot 910 represents simulated results when the relationship between ρ and β is |Kp|/|Kd|=0.15. Plot 911 represents simulated results when the relationship between ρ and β is |Kp|/|Kp|=0.10. Finally, plot 912 represents simulated results when the relationship between ρ and β is |Kp|/|Kd|=0.05.
FIG. 10 is a chart 1000 plotting bit error rate (BER) performance that may be achieved using two example SPS-SLM embodiments. Chart 1000 includes a signal-to-noise ratio (SNR) axis 1002 and a BER axis 1004. The SPS used to generate plot 1000 were generated using a convex optimization algorithm, according to an embodiment. In addition, an ideal soft limiter channel was used with an input backoff of 3 dB. The CFO was set to a constant (ε=0.2, where ε is a function of the subcarrier spacing, 1/Ts, and number of subcarriers, N, where Ts=NT, and T is the baseband sampling period: Thus, an ε=0.2 represents a carrier frequency offset of 20% of the subcarrier spacing), the multi-path channel was set to length 16 with an exponential delay spread such that
A ⁢ ∑ τ = 0 14 ⁢ ⅇ - τ .
Also, N=256, with |Kp|=16, |Kd|=240, and |Kn|=0. The pilot tones were evenly spaced with equal power, and the embedding factors were chosen to be β=0.25 and ρ=0.35.
Trace 1010 plots BER performance for a system in which embedded synchronization was used without any PAR reduction considerations (e.g., the candidate number quantity, D=1, and Sk (1) was generated with a prescribed power profile but random phases). In contrast, trace 1012 plots BER performance for a system in which PAR reduction was achieved using SPS generated in accordance with an embodiment, and with a candidate number quantity, D=1 (no selective mapping). A comparison between plots 1010 and 1012 indicates that significantly improved BER performance may be achieved when PAR reduction is used, in accordance with an example embodiment. Trace 1014 plots BER performance for a system in which SPS-SLM was used with PAR reduction, in accordance with various embodiments, and with a candidate number quantity, D=8. A further comparison between plots 1012 and 1014 indicates that even further improved BER performance may be achieved when the candidate number quantity, D, is increased.
FIG. 11 is a chart 1100 indicating acquisition performance for a system employing an embodiment versus a system that uses traditional binary sequences. Acquisition performance was measured as a plot of the probability of false detection, Pf (measured along axis 1102) versus the probability of missed detection, Pm (measured along axis 1104). To generate the chart 1100, an acquisition circuit was simulated, and three different synchronization/pilot sequences having N=64, Kn=8, Kd=49, Kp=7, and a signal-to-noise ratio of about −7 dB. Plot 1110 (square data points) corresponds to a pseudo-noise code derived from an m-sequence with a 31-chip code. To make the bandwidth consistent between codes, the m-sequence chips were repeated to provide two samples per chip. Plot 1111 (circular data points) corresponds to an SPS generated according to an embodiment, where β=1, and plot 1112 (diamond data points) corresponds to an SPS generated according to an embodiment, where β=1/7. A comparison between plots 1110-1112 indicates that comparable acquisition performance may be achieved using SPS generated according to an embodiment when compared with traditional m-sequences.
FIG. 12 is a chart 1200 indicating channel estimation performance for a system using SPS generated according to an embodiment and a system that uses traditional binary sequences. Channel estimation performance was measured as a plot of the embedding factor, ρ (measured along axis 1202) versus the mean square error (measured along axis 1204). Plot 1210 corresponds to channel estimation performance for a signal in which channel estimation pilots and a pseudo-noise code derived from an m-sequence were embedded according to traditional techniques. Plot 1211 corresponds to channel estimation performance for a signal in which an SPS generated according to an embodiment were embedded. A comparison between plots 1210 and 1211 indicates that low power portions of the m-sequence, which are non-flat frequency response, results in higher mean square error, when compared with an SPS generated according to an embodiment, which has a relatively flat frequency response. Accordingly, the use of SPS generated according to an embodiment may result in significantly improved channel estimation performance.
FIG. 13 is a chart 1300 illustrating a complementary cumulative distribution function (CCDF) of the PAR for signals produced using two different embedding sequences and various embedding factors, ρ, in accordance with various example embodiments. A first group 1310 of traces, which includes traces 1301, 1302, 1303, and 1304, plot CCDF curves that correspond to SPS that are constant modulus with a 0 dB PAR, and for embedding factors of ρ=0, ρ=0.3, ρ=0.5, and ρ=0.7, respectively. A second group 1312 of traces, which includes traces 1305, 1306, and 1307, plot CCDF curves that correspond to a time-domain impulse SPS with a 9 dB PAR for embedding factors of ρ=0.3, ρ=0.5, and ρ=0.7, respectively. A comparison between the two groups 1310, 1312 of traces indicates that, for a given embedding factor, significant PAR reductions may be achieved using constant modulus SPS, in accordance with various embodiments. For example, a PAR reduction at the 10−3 probability level is about 2 dB from the ρ=0.3, constant modulus embedded SPS case (trace 1302) to the ρ=0.3, 9 dB PAR embedded SPS case (trace 1305).
However, it has been found that, even though SPS generated using a POCS algorithm, in accordance with various embodiments, are not constant modulus, these SPS do have a low enough PAR to realize a large OFDM PAR reduction. When a lowest-PAR SPS found after 300 iterations and 150 initial conditions is embedded into an OFDM signal, a plot (not illustrated) of the CCDF of an OFDM PAR is virtually indistinguishable from the CCDF curves that correspond to SPS that are constant modulus with a 0 dB PAR (e.g., traces 1301-1304.
FIG. 14 is a plot 1400 of the convergence characteristics of the PAR for the five lowest-PAR SPS (out of 150 different initial conditions), which were generated according to an embodiment. The parameters used to generate the five traces within plot 1400 were N=64, Kn=4, Kp=7, and the amount of power in the non-pilot subcarriers was 20 percent of the power in the pilot subcarriers. From plot 1400, it is apparent that the PAR is well below 1 dB for the given initial conditions after 300 iterations. Although the convergence characteristics of the SPS corresponding to the remaining 145 different initial conditions is not plotted in FIG. 14, the PAR was below 1 dB for the vast majority of those remaining SPS.
FIG. 15 is a scatter plot 1500 of the PAR and mean square error (MSE) after 600 iterations of the SPS generation methods discussed previously (e.g., blocks 604-616, FIG. 6) for 150 different initial conditions. The parameters used to generate the scatter plot 1500 were the same as for the plot 1400 illustrated in FIG. 14. In scatter plot 1500, MSE corresponds to the mean squared difference between Ŝ[k] and √{square root over (P[k])}, and the PAR is the PAR of ŝ[n]. As scatter plot 1500 indicates, the PAR and MSE are highly dependent on the initial conditions of the SPS generation algorithm.
Embodiments of methods and apparatus for generating SPS for embedding in wireless signals have now been described above. The foregoing detailed description is merely exemplary in nature and is not intended to limit the inventive subject matter or the application and uses of the inventive subject matter to the described embodiments. Furthermore, there is no intention to be bound by any theory presented in the preceding background or detailed description.
An embodiment includes a method for generating a set of synchronization/pilot sequences. The method embodiment includes the steps of generating a candidate synchronization/pilot sequence using initial conditions, and repeating the generating step a first number of iterations using different initial conditions for each iteration to generate a plurality of candidate synchronization/pilot sequences. The method embodiment also includes performing a plurality of correlations among the candidate synchronization/pilot sequences to generate a plurality of correlation values, determining a plurality of permutations of the candidate synchronization/pilot sequences, wherein each permutation of the plurality of permutations includes a different set of candidate synchronization/pilot sequences, and wherein each permutation includes a number, D, of candidate synchronization/pilot sequences, and identifying a selected permutation from the plurality of permutations, wherein the selected permutation corresponds to the set of synchronization/pilot sequences being generated.
An embodiment of a method for generating a wireless signal in which a synchronization/pilot sequence is embedded includes the steps of combining each synchronization/pilot sequence of a set of synchronization/pilot sequences with phase shifted input data to produce a plurality of combined signals, determining peak-to-average power ratios for at least some of the combined signals, identifying a selected combined signal based on the peak-to-average power ratios, and transmitting the selected combined signal over a wireless communication channel. In an embodiment, the set of synchronization/pilot sequences includes synchronization/pilot sequences generated by generating a plurality of candidate synchronization/pilot sequences, wherein each candidate synchronization/pilot sequence of the plurality is generated using a different random phase, performing a plurality of correlations among the candidate synchronization/pilot sequences to generate a plurality of correlation values, determining a plurality of permutations of the candidate synchronization/pilot sequences, wherein each permutation of the plurality of permutations includes a different set of candidate synchronization/pilot sequences, and wherein each permutation includes a number, D, of candidate synchronization/pilot sequences, and identifying a selected permutation from the plurality of permutations, wherein the selected permutation corresponds to the set of synchronization/pilot sequences being generated.
generating a candidate synchronization/pilot sequence using initial conditions;
repeating the generating step a first number of iterations using different initial conditions for each iteration to generate a plurality of candidate synchronization/pilot sequences;
performing a plurality of correlations among the candidate synchronization/pilot sequences to generate a plurality of correlation values;
determining a plurality of permutations of the candidate synchronization/pilot sequences, wherein each permutation of the plurality of permutations includes a different set of multiple candidate synchronization/pilot sequences, and wherein each permutation includes a number, D, of candidate synchronization/pilot sequences; and
identifying a selected permutation from the plurality of permutations based on the correlations among the candidate synchronization/pilot sequences in the selected permutation, wherein the selected permutation corresponds to the set of synchronization/pilot sequences being generated.
2. A method for generating a set of synchronization/pilot sequences, the method comprising the steps of:
generating a candidate synchronization/pilot sequence using initial conditions by
generating a constant modulus phase sequence having the initial conditions corresponding to a random phase,
applying a power profile to the constant modulus phase sequence to produce a power-adjusted phase sequence,
performing a time-domain transformation of the power-adjusted phase sequence to produce a time-domain sequence,
setting amplitudes of the time-domain sequence to unity while maintaining phases of the time-domain sequence to produce an amplitude-adjusted time-domain sequence,
performing a frequency-domain transformation of the amplitude-adjusted time-domain sequence to produce a modified frequency-domain sequence,
applying the power profile to the modified frequency-domain sequence while maintaining phases of the modified frequency-domain sequence to produce a power-adjusted frequency-domain sequence, and
repeating, at least an additional time for the power-adjusted frequency-domain sequence, the steps of performing the time-domain transformation, setting the amplitudes, performing the frequency-domain transformation, and applying the power profile to produce the pilot sequence;
determining a plurality of permutations of the candidate synchronization/pilot sequences, wherein each permutation of the plurality of permutations includes a different set of candidate synchronization/pilot sequences, and wherein each permutation includes a number, D, of candidate synchronization/pilot sequences; and
identifying a selected permutation from the plurality of permutations, wherein the selected permutation corresponds to the set of synchronization/pilot sequences being generated.
3. The method of claim 1, wherein performing the plurality of correlations comprises the steps of:
performing a plurality of cross-correlations among the candidate synchronization/pilot sequences to generate a plurality of maximum cross-correlation values;
performing a plurality of auto-correlations among the candidate synchronization/pilot sequences to generate a plurality of secondary maximum auto-correlation values; and
including the maximum cross-correlation values and the secondary maximum auto-correlation values in the plurality of correlation values.
4. A method for generating a set of synchronization/pilot sequences, the method comprising the steps of:
performing a plurality of correlations among the candidate synchronization/pilot sequences to generate a plurality of correlation values, wherein performing the plurality of correlations comprises performing P2 cross-correlations, and P auto-correlations, where P is a number of candidate synchronization/pilot sequences being correlated;
eliminating, from the candidate synchronization/pilot sequences, any candidate synchronization/pilot sequences that do not meet a peak-to-average power ratio selection criteria.
6. The method of claim 5, wherein the step of eliminating uses a peak-to-average power ratio threshold value as the peak-to-average power ratio selection criteria, and comprises eliminating, from the candidate synchronization/pilot sequences, any candidate synchronization/pilot sequence that exceeds the peak-to-average power ratio threshold.
combining each synchronization/pilot sequence of the set of synchronization/pilot sequences with phase shifted input data to produce a plurality of combined signals;
determining peak-to-average power ratios for at least some of the combined signals;
identifying a selected combined signal based on the peak-to-average power ratios; and
transmitting the selected combined signal over a wireless communication channel.
10. The method of claim 9, wherein identifying the selected combined signal comprises the step of:
identifying the selected combined signal as the combined signal of the plurality of combined signals that has a lowest peak-to-average power ratio of the peak-to-average power ratios.
receiving a received combined signal from the wireless communication channel, wherein the received combined signal represents a channel-affected version of the selected combined signal;
determining estimated channel errors within the received combined signal based on the set of synchronization/pilot sequences; and
applying corrections to the received combined signal, based on the estimated channel errors, to produce output data.
12. A method for generating a wireless signal in which a synchronization/pilot sequence is embedded, the method comprising the steps of:
combining each synchronization/pilot sequence of a set of synchronization/pilot sequences with phase shifted input data to produce a plurality of combined signals, wherein the set of synchronization/pilot sequences includes synchronization/pilot sequences generated by generating a plurality of candidate synchronization/pilot sequences, wherein each candidate synchronization/pilot sequence of the plurality is generated using a different random phase, performing a plurality of correlations among the candidate synchronization/pilot sequences to generate a plurality of correlation values, determining a plurality of permutations of the candidate synchronization/pilot sequences, wherein each permutation of the plurality of permutations includes a different set of multiple candidate synchronization/pilot sequences, and wherein each permutation includes a number, D, of candidate synchronization/pilot sequences, and identifying a selected permutation from the plurality of permutations based on the correlations among the candidate synchronization/pilot sequences in the selected permutation, wherein the selected permutation corresponds to the set of synchronization/pilot sequences being generated;
13. The method of claim 12, wherein identifying the selected combined signal comprises the step of:
a transmitter adapted to generate and transmit a wireless signal in which a synchronization/pilot sequence is embedded by
combining each synchronization/pilot sequence of a set of synchronization/pilot sequences with phase shifted input data to produce a plurality of combined signals, wherein the set of synchronization/pilot sequences includes synchronization/pilot sequences generated by generating a plurality of candidate synchronization/pilot sequences, wherein each candidate synchronization/pilot sequence of the plurality is generated using a different random phase, performing a plurality of correlations among the candidate synchronization/pilot sequences to generate a plurality of correlation values, determining a plurality of permutations of the candidate synchronization/pilot sequences, wherein each permutation of the plurality of permutations includes a different set of multiple candidate synchronization/pilot sequences, and wherein each permutation includes a number, D, of candidate synchronization/pilot sequences, and identifying a selected permutation from the plurality of permutations based on the correlations among the candidate synchronization/pilot sequences in the selected permutation, wherein the selected permutation corresponds to the set of synchronization/pilot sequences being generated,
determining peak-to-average power ratios for at least some of the combined signals,
identifying a selected combined signal based on the peak-to-average power ratios, and
16. The system of claim 15, wherein the transmitter is a wireless communication device selected from a group that includes a cellular telephone, a radio, an unmanned autonomous vehicle, a one-way pager, a two-way pager, a personal data assistant, a computer, a base station, a wireless transmitter, and a wireless transceiver.
a receiver adapted to receive a received combined signal from the wireless communication channel, wherein the received combined signal represents a channel-affected version of the selected combined signal, determine estimated channel errors within the received combined signal based on the set of synchronization/pilot sequences, and apply corrections to the received combined signal, based on the estimated channel errors, to produce output data.
19. The system of claim 18, wherein the receiver is a wireless communication device selected from a group that includes a cellular telephone, a radio, an unmanned autonomous vehicle, a one-way pager, a two-way pager, a personal data assistant, a computer, a base station, a wireless transmitter, and a wireless transceiver.
20. The system of claim 15, wherein the system is a multi-carrier communication system selected from a group that includes a multi-carrier based, ultra-wideband system, an orthogonal frequency division multiplexing multiple access system, a multi-carrier code division multiple access system, a digital video broadcasting system, a WiMax system, a long range broadband wireless system, a wireless local area network system, and an 802.11a system.
US12/051,535 2007-04-13 2008-03-19 Methods and apparatus for generating synchronization/pilot sequences for embedding in wireless signals Active 2030-08-08 US8358644B2 (en)
US12/051,535 US8358644B2 (en) 2007-04-13 2008-03-19 Methods and apparatus for generating synchronization/pilot sequences for embedding in wireless signals
US12/725,985 US8379752B2 (en) 2008-03-19 2010-03-17 Methods and apparatus for multiple-antenna communication of wireless signals with embedded synchronization/pilot sequences
US12/725,985 Continuation-In-Part US8379752B2 (en) 2007-04-13 2010-03-17 Methods and apparatus for multiple-antenna communication of wireless signals with embedded synchronization/pilot sequences
US20090003308A1 US20090003308A1 (en) 2009-01-01
US8358644B2 true US8358644B2 (en) 2013-01-22
US8457226B2 (en) * 2008-10-10 2013-06-04 Powerwave Technologies, Inc. Crest factor reduction for OFDM communications systems by transmitting phase shifted resource blocks
CN102362448B (en) * 2008-11-26 2015-11-25 安德鲁无线系统有限责任公司 For the single-input single-output repeater of relaying MIMO signal
FR3038800A1 (en) * 2015-07-09 2017-01-13 Stmicroelectronics (Rousset) Sas Method for processing a signal from a transmission channel, particularly an in-line carrier current vehicle signal, and especially channel estimation, and corresponding receiver
WO2017151027A1 (en) * 2016-03-02 2017-09-08 Telefonaktiebolaget Lm Ericsson (Publ) Methods and devices operating with fine timing reference signals transmitted occasionally
EP3479536A1 (en) * 2016-06-29 2019-05-08 Telefonaktiebolaget LM Ericsson (PUBL) Adaptive selection of signal-detection mode
US20040196921A1 (en) 2003-04-02 2004-10-07 Stratex Networks, Inc. Adaptive broadband post-distortion receiver for digital radio communication system
US20080089437A1 (en) 2006-10-11 2008-04-17 Nokia Corporation Method and apparatus for reducing peak-to-average power ratio of a signal
US20090011722A1 (en) 2007-04-13 2009-01-08 General Dynamics C4 Systems, Inc. Methods and apparatus for wirelessly communicating signals that include embedded synchronization/pilot sequences
US20090207936A1 (en) 2008-02-14 2009-08-20 Broadcom Corporation Real and complex spectral shaping for spectral masks improvements
US20090316826A1 (en) 2008-06-21 2009-12-24 Vyycore Corporation Predistortion and post-distortion correction of both a receiver and transmitter during calibration
US20100002784A1 (en) * 2006-02-06 2010-01-07 Ondrej Hlinka Method for Reducing Peak-To-Average Power Ratio in an Ofdm Transmission System
US20100118990A1 (en) 2008-11-13 2010-05-13 Hwang-Soo Lee Apparatus for synchronizing ofdm signal using open-loop frequency synchronization method and frequency offset estimation scheme using the apparatus
US7764593B2 (en) 2003-10-24 2010-07-27 Electronics And Telecommunications Research Institute Downlink signal configuring method and device in mobile communication system, and synchronization and cell searching method and device using the same
US20120140838A1 (en) 2005-08-22 2012-06-07 Qualcomm Incorporated Method and apparatus for antenna diversity in multi-input multi-output communication systems
Anderson, K., et al. "Two dimensional diversity enhancement for tactical wireless networks using multi-carrier cooperative networking," in proceedings of SDR Forum Technical Conference 2006.
Cao, Z., et al. "Frequency synchronization fro generalized OFDMA uplink," in Proc. IEEE Globecomm, 2004.
Choi, J., et al. "Carrier frequency offset compensation for uplink of OFDM-FDAM systems," IEEE Trans. Commun., 2005.
Giannakis, G., et al. Space-Time Coding for Broadband Wireless Communications, John Wiley and Sons, Hoboken, NJ, 2007. www.researchandmarkets.com/reports/449857.
Huang, D., et al. "An Interference cancellation scheme for carrier frequency offsets correction in OFDMA systems," IEEE Trans. Commun., 2005.
M. Morelli et al., "A Comparison of Pilot-Aided Channel Estimation Methods for OFDM Systems," IEEE Trans. on Signaling Processing, vol. 49, No. 12, pp. 3065-3073, 2001.
Ma, Q. et al. "Differential space-time-frequency coded OFDM with maximum multipath diversity," IEEE Trans. Wireless Commun., vol. 4, No. 5, pp. 2232-2243, Sep. 2005.
Notice of Allowance issued Apr. 3, 2012 in U.S. Appl. No. 12/649,672.
Notice of Allowance issued Oct. 27, 2011 in U.S. Appl. No. 12/038,983.
S. Chennakeshu et al., "Error Rates for Rayleigh Fading Multichannel Reception of MPSK Signals," IEEE Transactions on Communications, vol. 43, pp. 338-346, February, March, Apr. 1995.
Schmidl, T.M., et al. "Robust frequency and timing synchronization for OFDM," IEEE Trans on Commun., vol. 45, No. 12, pp. 1613-1621, Dec. 1997.
Tonello, A. "Multiuser detection and turbo multiuser decoding for asynchronous multitone multiple access," in Proc. IEEE Veh. Techn. Conf., 2002.
U.S. Office Action issued Jun. 24, 2011 in U.S. Appl. No. 12/038,983.
USPTO "Final Office Action" mailed Aug. 28, 2012; U.S. Appl. No. 12/567,509, filed Sep. 25, 2009.
USPTO "Non-Final Office Action" mailed Jun. 22, 2012; U.S. Appl. No. 12/567,509, filed Sep. 25, 2009.
USPTO "Non-Final Office Action" mailed May 25, 2012; U.S. Appl. No. 12/567,505, filed Sep. 25, 2009.
USPTO "Notice of Allowance" mailed Aug. 21, 2012; U.S. Appl. No. 12/567,505, filed Sep. 25, 2009.
USPTO "Notice of Allowance" mailed Jul. 30, 2012; U.S. Appl. No. 13/359,205, filed Jan. 26, 2012.
USPTO "Notice of Allowance" mailed Jun. 19, 2912; U.S. Appl. No. 12/649,672, filed Dec. 30, 2009.
USPTO "Notice of Allownce" mailed Oct. 11, 2012; U.S. Appl. No. 12/725,985, filed Mar. 17, 2010.
USPTO Non-Final Office Action mailed May 3, 2012; U.S. Appl. No. 13/359,205, filed Jan. 26, 2012.
US8131218B2 (en) 2012-03-06
US8619841B1 (en) 2013-12-31 Transceiver with carrier frequency offset based parameter adjustment
CN100542158C (en) 2009-09-16 In communication system, estimate the apparatus and method of interference and noise
KR100929470B1 (en) 2009-12-02 Wireless transmitting device and wireless transmitting method
Lee et al. 2006 SNR analysis of OFDM systems in the presence of carrier frequency offset for fading channels
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:BAXLEY, ROBERT JOHN;KLEIDER, JOHN ERIC;REEL/FRAME:020707/0983