Source: http://www.google.com/patents/US7106246?ie=ISO-8859-1&dq=6,249,089
Timestamp: 2014-11-28 13:19:16
Document Index: 595852232

Matched Legal Cases: ['art 1', 'art 1', 'arts 1', 'arts 1', 'art 1', 'art 1', 'art 1', 'art 1', 'art 34', 'art 34', 'art 34', 'art 34', 'art 28', 'art 28', 'art 3', 'art 3', 'art 3', 'art 3', 'art 3', 'art 3', 'art 3', 'art 3']

Patent US7106246 - Oscillator coupled to an antenna and an application - Google PatentsSearch Images Maps Play YouTube News Gmail Drive More »Sign inAdvanced Patent SearchPatentsA radio frequency device has an antenna for capturing an incoming signal for processing by the device or for radiating an outgoing signal from the device and a signal processor having one or more synchronous oscillators responsive to an input signal for providing an amplified output signal without using...http://www.google.com/patents/US7106246?utm_source=gb-gplus-sharePatent US7106246 - Oscillator coupled to an antenna and an applicationAdvanced Patent SearchPublication numberUS7106246 B1Publication typeGrantApplication numberUS 10/771,944Publication dateSep 12, 2006Filing dateFeb 4, 2004Priority dateFeb 5, 2003Fee statusPaidPublication number10771944, 771944, US 7106246 B1, US 7106246B1, US-B1-7106246, US7106246 B1, US7106246B1InventorsKevin W LindellOriginal AssigneeKevin W LindellExport CitationBiBTeX, EndNote, RefManPatent Citations (9), Non-Patent Citations (8), Referenced by (19), Classifications (10), Legal Events (3) External Links: USPTO, USPTO Assignment, EspacenetOscillator coupled to an antenna and an applicationUS 7106246 B1Abstract A radio frequency device has an antenna for capturing an incoming signal for processing by the device or for radiating an outgoing signal from the device and a signal processor having one or more synchronous oscillators responsive to an input signal for providing an amplified output signal without using much power. An application is a radio frequency (RF) transponder (tag) for receiving an RF signal from an interrogator includes a tag antenna for receiving the RF signal from the interrogator and a receiver section connected to the tag antenna wherein the receiver consumes a significantly lower amount of power than conventional receiver technologies by using one or more synchronous oscillators.
The invention claimed is: 1. Portable radio frequency device comprising a power supply and a power control for controlling power provided to said device during a limited period so as to consume a correspondingly limited amount of power from said power supply,
CROSS REFERENCE TO RELATED APPLICATION This application claims priority from U.S. provisional application Ser. No. 60/445,338 filed Feb. 5, 2003
TECHNICAL FIELD The present invention relates to a low power radio frequency device that is either a receiver that responds to an incoming radio frequency interrogation signal by providing an output signal or a transmitter that provides an outgoing radio frequency signal in response to an input signal, or a combination of both a receiver and a transmitter.
BACKGROUND OF THE INVENTION A device of this type would include a radio frequency transponder. Such a device transmits a reply signal upon reception of an incoming signal. It usually includes a receiver, responsive to the incoming signal, for providing an amplified intermediate signal and a transmitter, responsive to the amplified intermediate signal, for transmitting the reply signal.
Synchronous oscillators (SOs) are known generally for instance from US 2003/0011438 A1 published Jan. 16, 2003 by Vasil Uzunoglu There, a modification of the synchronous oscillator is described, having regenerative positive feedback. The circuit includes an amplifier, a high-Q tank circuit, and a conventional synchronous oscillator feedback network. An additional feedback path provides a negative impedance conversion effect, according to Uzunoglu. There are various articles and US Patents by the same inventor Vasil Uzunoglu, (U.S. Pat. Nos. 4,335,404, 4,274,067, 4,356,456) relating to SOs (see the list at page 15 of US 2003/0011438). Various applications of the modified SO are shown. A further important characteristic of the synchronous oscillator is its energy efficiency. According to Uzunoglu, the regenerative feedback results in very little power dissipation, enabling the circuit to operate highly effectively with very low power supply requirements, for example, approximately 2�3 volts.
BRIEF SUMMARY OF THE INVENTION An object of the invention is to provide a portable radar or radio frequency device with improved power efficiency. This will allow the power source to last for a longer period of time for a given sensitivity. This can also allow the power source to become smaller.
BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWING FIG. 1 shows an application as part of a device also including an antenna and a synchronous oscillator.
DETAILED DESCRIPTION OF THE INVENTION FIG. 1 shows a radio frequency device, according to the present invention, comprising an application 1 cd coupled to an antenna 1 a and to an oscillator which is preferably but not necessarily a synchronous oscillator (S.O.) 1 b. The antenna and the S.O. work together with the application 1 cd to carry out the ends of the application which can be various. According to the present invention, the application takes advantage of the signal processing properties of the S.O. The application 1 cd is shown comprising two parts, one part 1 c coupled between the antenna 1 a and the S.O. and the other part 1 d coupled to the S.O. on the other side of the S.O., but it should be realized that the application may comprise one or both parts 1 c, 1 d. Either or both application parts 1 c, 1 d can exist in a given embodiment and their functions can be separate or overlapping but nonetheless together constitute an application. The application therefore interfaces with an input, an output, or both the input and the output of the S.O. If the application part 1 c is interfaced with the input to an S.O. 1 b, then the application part 1 d interfaces with the output of an S.O. 1 b. In a similar fashion, if the application part 1 c is interfaced with the output of an S.O. 1 b then the application part 1 d interfaces to the input of the S.O. 1 b. As shown below, signals can be received, transmitted, or both by the synchronous oscillator in these configurations.
The power control 1 h, 2 h, 3 h of FIGS. 1�3 may take many different forms. For instance, it could be a pulse generator that closes the switch 1 k, 2 k, 3 k according to the timing and duration of its pulsed output. It could be a time-of-day clock that closes the switch for a timed period one or more times per day. It could be a manual control that allows a user to turn the device on or off with the push of a button. A low cost example that only uses a few microamps would be a fifteen stage (15 flip flops) counter that counts clocks output by a clock oscillator connected to a crystal oscillator providing a 32768 Hz continuous wave output. Every time the counter reaches full count to 32768 (every second), it outputs a pulse that drives another counter driven by the clock oscillator that may count 333 clocks to provide an on time pulse of about 10 milliseconds every second. The on time pulse is used to close the switch 1 k, 2 k, or 3 k. Therefore, to conserve power, according to the teachings hereof, a power control circuit can be used to deliver power with a specific duty cycle where there is a period of time where the device is powered and available to operate, and there will be a period of time that the device is powered down and not available. This conserves power from a power source at the cost of availability. Using the above-mentioned parameters, the device can be made to be available within a 5 second period to reduce power consumption as compared to a device that is continuously powered. The device is placed in a state of minimized power consumption to conserve the limited capacity of the power supply. During the state of zero or minimal power consumption, the device is off or not fully operational and will not perform to specification. A periodic duration of applied power allows the device to perform to desired specification while consuming power.
The devices of FIGS. 1�3 are put to use by means of an application having some purpose. Each is therefore a device that transforms a signal into a useful output having some end purpose. Stated otherwise, the purpose of the device is achieved by the cooperation of its parts including the application part. The purpose may vary widely and may include those shown below such as RF identification (in a transponder application), ranging, vehicle tire air pressure sensing, controlling a switch, conveying information, quadrature modulation for communications circuits, and many others not shown.
However, the signal processing of the S.O. block 2 b can be affected by a second application part 34 e. Second application part 34 e accepts signals from one or more synchronous oscillators 22, 24 . . . 26 within the S.O. block 2 b on a line 34 be. Likewise, second application part 34 e can inject signals into one or more synchronous oscillators 22, 24 . . . 26 within the S.O. block 2 b on a line 34 eb. The second application part 34 e can be viewed as part of an application also including first application part 28 c so as to be consistent with the previous description of FIG. 2 and the examples of FIGS. 4�7. In this case, the first application part 28 c is merely a node.
In the context of the above-discussed receiver and transmitter, the S.O. 2 b, 3 b of FIG. 12( a) in effect performs double-duty, performing both the role of the receiver S.O. 2 b of FIGS. 2 and the transmitter S.O. 3 b of FIG. 3. Alternatively, the S.O. 22 can be viewed as the S.O. 2 b of FIG. 2 while the S.O. 26 may be viewed as the S.O. 3 b of FIG. 3. The receiver and transmitter boundaries of FIG. 12 would then change accordingly to show the receiver and transmitter as non-overlapping, e.g., with S.O. 24 and any other intermediate S.O.s �coupling� distinct receiver and transmitter parts of the transponder. Or, S.O. 24 could be shown within the receiver and any other S.O.s within the transmitter. It should be evident that a cascaded chain of S.O.s can be viewed as partly in the receiver and partly in the transmitter.
The skirt selectivity of the filtering characteristic defines the attenuation of signals outside the passband of the filter. The skirt selectivity of the synchronous oscillator has been shown to be better than a conventional three-section filter, as shown in �The Synchronous Oscillator: A synchronization and Tracking Network� by Vasil Uzunoglu et al, IEEE Journal of Solid-State Circuits, vol. SC-20, No. 6, Dec. 1985.
Furthermore, the synchronous oscillator can be modified to produce frequency translation, according to the invention. If a radio transponder is required to operate with an output frequency that differs from the input frequency, the frequency converting stage can take advantage of the ability of the synchronous oscillator to lock to integer and noninteger frequency multiples as described in U.S. Pat. No. 4,356,456, in �The Synchronous Oscillator: A synchronization and Tracking Network� by Vasil Uzunoglu et al, IEEE Journal of Solid-State Circuits, vol. SC-20, No. 6, Dec. 1985, in �Some important properties of Synchronous Oscillators�, by Vasil Uzunoglu et al, Proceeding of the IEEE, Vol. 74, No. 3, March 1986, in �Synchronous and Coherent Phase-Locked Synchronous Oscillators: New Techniques in Synchronization and Tracking� by Vasil Uzunoglu et al, IEEE Transactions on Circuits and Systems, Vol. 36, No. 7, July 1989, and in �Theoretical Analysis of a Coherent Phase Synchronous Oscillator� by Marion Tam et al, IEEE Transactions on Circuits and Systems, Vol. 39, No. 1, Jan. 1992.
The base-emitter voltage (Vb-e) across Q1 is shown in FIG. 14. When the junction is forward biased, placing it in the linear operating region, the oscillator output is phase modulated by the input signal on the line 18 through the transistor Q2. The Colpitts oscillator includes the transistor Q1, a RF coupling capacitor C1, the coil L and a pair of capacitors C2 and C3. The combination of devices L-C2�C3 form a tank circuit located above Q1 that sets the free-running frequency of the S.O. 22. The path created by C1, the connection between the emitter of Q1 and the node between C2 and C3, as well as the transistor Q2, introduce a set of positive and negative feedback paths that allow oscillations to exist. The Colpitts oscillator is isolated by a radio frequency choke Lc. The choke also could be another type of isolation device, such as a resistor or in some instances may not be required. The transistors are direct current biased by a bias network to be in the linear region.
FIG. 15 shows how the various gain factors of the three-stage S.O. (22, 24, 26) example of FIG. 12( a) become additive in amplifying the input signal on the line 28 cb. The first stage 22 is responsive to the input signal occupying a relatively narrow input bandwidth, as shown. It amplifies the input signal level to a higher level using the Stage 1 gain factor. The output of Stage 22 is provided as an input to Stage 24, which has its own Stage 2 gain. This Stage 2 gain adds to the Stage 1 gain and has a wider input bandwidth which must respond, due to the properties of the synchronous oscillator of Stage 1. Similarly, the second Stage 24 provides an output signal as an input to the third Stage 26, which has a still wider frequency bandwidth presented thereto. Stage 3 adds its gain to the total gain applied to the input signal on the line 28 cb, so that the intermediate signal on the line 28 bd has been amplified considerably, as shown by the relative input level increase of FIG. 15. A range of frequencies as determined by the input bandwidth of the first stage is therefore made available to the transmitter for use in providing the output signal on the line 28 da. If frequency translation is required and the previously mentioned methods utilizing an non-modified S.O. are undesirable because of the limited number of input/output frequency combinations, there is another solution that requires a small modification to a S.O. according to the present invention. Instead of the S.O. locking to a frequency multiple or submultiple, the creation of a new non-related frequency can be made, and mixed with the injection frequency of the S.O. To accomplish this, a mixing process is performed within the S.O. stage. The synchronous oscillator operates with non-linearities by placing the Colpitts oscillator into a non linear operating region as shown in FIG. 14. The input of the S.O. is also switched to accept signals during the linear operating region of the Colpitts oscillator which is a portion of the cycle at the rate of oscillation determined by its resonant network. These characteristics can be utilized to produce frequency mixing. This mixing function will create new output frequencies based on the input frequencies. Mixed products of the input frequencies may be used to lock the S.O. The non-modified S.O. can only synchronize to input signals that are periodically related to the S.O. output frequency. Input signals to the S.O. that do not cause the S.O. to properly synchronize, will frequency beat or heterodyne and create mixed products determined by the operating frequency of the S.O. and the injected input frequency. To frequency heterodyne two or more signals, one of the frequencies injected into the S.O. can be used as the signal that is within the lock bandwidth of the S.O., while the other signal is outside the lock bandwidth. Frequencies at the output of the S.O. are products of the input signals, where one input signal is effectively frequency translated by the other input signal. This technique of translating carrier frequencies is known as superheterodyning, usually abbreviated to �superhet.� The purpose of this translation is to allow further amplification of the desired signal at a carrier frequency convenient for circuit design. Typically, superhet receivers provide improved rejection of unwanted signals. In the transponder application here, the translation would allow a different reply frequency than the injected input frequency to the transponder.
A radio frequency tag is a device that is attached to something to identify, classify, or label it. The tag information is conveyed to an interrogator unit by wireless radio frequency means. The transponder of FIG. 12, can be applied as a radio frequency tag, and will now be described in detail in connection with FIG. 12( a), with three stages of synchronous amplifiers 22, 24, 26 acting in different modes. Although the transponder can be configured with more or less stages, an example will be useful to show different uses for the synchronous amplifier stages. The stages in the transponder can be compared to the stages in a conventional radio receiver because the signals in the stages of a conventional radio receiver are similar to signals in the stages of a transponder. A conventional radio receiver that produces an audible output contains radio frequency stages that have specific functions. An example of a common radio receiver contains four different stages to receive a radio signal and produce an audio output. The function of the first stage is to amplify the receive signal captured from an antenna. The function of the second stage is radio tuning with a local oscillator and a heterodyne mixer circuit. The second stage frequency translates the amplified receive signal from stage one to a specific frequency that a third stage operates at. The function of the third stage is to filter undesired signals and is optimized to work at a specific frequency that is output from the second stage. The function of the fourth stage is to detect the audio signal present on the frequency converted and filtered radio frequency signal and produce an audible output. The transponder described does not produce an audio output so a stage performing the function of the fourth stage as described above is not required. The filtered frequency output from the third stage described above will be used as the transmit signal in the transponder. In the transponder, S.O. stage 22 will be used as the first stage and will function as an amplifier for the received signals captured from the receive antenna. FIG. 13 shows the S.O. 22 in detail. This is similar to the first stage 22 shown in FIG. 21 which is shown to be an amplifier for the frequency F1. S.O. stage 24 of FIGS. 12( a), 16, and 21 will be used as the second stage which will frequency translate the signal from stage one to a desired transponder transmit frequency via the use of a local oscillator and heterodyning mixer. S.O. 22 as shown in FIG. 13 has one input 18. In order to heterodyne two or more signals by an S.O. stage, it is required to inject the two or more signals into the stage. If one of the mixing signals is generated locally in the S.O. stage, the other heterodyne mix signal is injected on the input to the S.O. stage requiring only one input. The signal generated locally and the signal externally injected would heterodyne and produce mixing products on the output of the S.O. stage. If no signals are generated locally to the S.O. stage, the signals that are to be heterodyned are injected externally to the S.O. stage as shown in FIG. 16 requiring an input for each different signal. This will require the S.O. stage to have available more than one input to accommodate the different injected signals. FIG. 16 shows the additional input and a resonant network 50 of S.O. 24 as compared to S.O. 22 of FIG. 13. The S.O. 24 is a modified version of S.O. 22 adding inputs 80,81 to make the S.O. 24 more flexible in use for inputting signals used for modulation and frequency conversion. S.O. stage 24 also has the capability of generating a signal locally through the use of a resonant network 50. This stage is similar in operation to the stage 24 shown in FIG. 21. By using a crystal resonator 50 in the feedback path (such as in FIGS. 16 and 18) on the input transistor Q2, the output will contain F1, the mix frequencies F1+Fxtal and F1−Fxtal. The LC network L-C2�C3 will still be tuned to F1. The resonator in S.O. 24 can be implemented with other devices besides a quartz crystal, as suggested by FIG. 19. A simple L-C network can be used as the resonator. Electrically tunable elements can be incorporated for the purpose of modulation which is not used in this example. S.O. stage 26 will be used as the third transponder stage and will filter any undesired signals from the second stage, S.O. 24, and output the filtered signal to the transmit antenna. S.O. 26 has the same structure as S.O. 22 of FIG. 13. As shown in FIG. 21, to filter the unwanted frequency components the final output stage 26 will be used as a single sideband filter by tuning the LC network L-C2�C3 to F1+Fxtal. Referring back to FIG. 12( a), the transponder is built by taking the configuration of FIG. 21 and adding antennas 28 a, 30 a and applications 28 c,28 d (either or one of which could be a mere node). The transponder will track input frequency F1 captured by antenna 28 a and will reply on the line 28 bd and on the line 28 da with F1+Fxtal. The signal on line 28 da is injected into the transmit antenna 30 a. The antenna 30 a in turn provides the outgoing radio frequency signal on the line 12. If the transponder application required modulation to be introduced, this would be possible through the use of application 34 e on a line 34 eb. Placement of a modulator in the transponder radio frequency chain is dependent on several factors. One factor is the type of modulation and another is its bandwidth. The transponder can be made to reply with Amplitude modulated (AM), FM, PM or spread spectrum signals.
The purpose of utilizing transmitter configurations as shown in FIG. 3 is to support a specific application such as a tire pressure measurement and reporting application shown in FIG. 30. The application shown in FIG. 30 is one where the reporting signal being transmitted by the antenna 3 a is a sensed signal on a line 3 e representing air pressure. An air pressure sensor located for instance inside a tire measures air pressure as indicated on the line 3 e and provides an output signal having a magnitude indicative thereof. The application of measuring an air pressure signal 3 e is accomplished by injecting into the S.O. a usable signal on a line 3 cb from an application part 3 c 5 that includes the sensor. An air pressure sensor that has the characteristic of changing its capacitive value with air pressure may be used as a resonant element in an oscillator (OSC) also included the application part 3 c 5 to produce such a useable signal 3 cb as shown in FIG. 30. As the capacitance value changes, the resonant frequency of the oscillator (OSC) also changes. The oscillator output signal on the line 3 cb is injected from the application part 3 c 5 into the S.O. 3 b. The application part 3 c 5 may also inject a controlled frequency reference signal on the line 3 cb into the S.O. 3 b to lock the S.O. frequency but this is not necessary if the stability of a free running S.O. is acceptable. The S.O. frequency mixes with the oscillator signal from the application part 3 c 5 and creates frequency sidebands. The output signal from the S.O. on line 3 bd is injected into the antenna 3 a by signal line 3 da 5 where an application part 3 d 5 is merely an electrical node with the signal 3 bd as an �input� and the signal 3 da 5 as an �output.� If a linear change in the resonant frequency of the oscillator operating from the air pressure sensor occurs from a comparable linear change in air pressure, the receiver capturing the transmitted reporting signal can determine the air pressure value by means of linear interpolation by measuring the frequency of the sideband signal. A simpler form of circuitry would involve integrating the air pressure sensor application part 3 c 5 into the S.O. 3 b, eliminating the need for a separate oscillator (OSC). For instance, a capacitive air pressure sensor could replace a capacitor such as the capacitor C4 in network 50 shown in one of the S.O. examples previously (see FIG. 15, for instance). As mentioned above, if the frequency stability of a free running S.O. is acceptable, the application part 3 c 5 does not need to inject a controlled frequency reference signal into the S.O. 3 b to cause frequency lock. As also previously described, the air pressure may be determined in the receiver by linear interpolation.
An application combining a receiver and transmitter section: a transponder will comprise a S.O. receive and transmit section used to measure distance between an interrogator and a transponder. To measure distance, the number of wavelengths of the radio frequency signal between the interrogator and the transponder will be used as a �yardstick� or measurement tool. The wavelength is the distance a wave travels in the time required to complete one cycle. The wavelength of the signal is equal to the frequency of the signal in cycles per unit time divided by the speed of propagation in a distance traveled expressed in length units per unit time leaving wavelength with the dimensions of cycles per unit length. Since the frequency of the interrogator, the frequency of the transponder reply signal, and the speed of propagation are known values, the number of wavelengths needed to traverse the distance between the interrogator and the transponder is readily calculated. If the distance from the interrogator to the transponder and back to the interrogator is less than one wavelength, the distance is unambiguously calculated by comparing the time or phase shift of the transmit and receive signal in the interrogator. A transmitted signal 14 out of an interrogator travels a distance and is received by the transponder 10 by an antenna 28 a as shown in FIG. 12( a). As the signal 14 propagates over an increased distance, the relative phase of the signal as compared to the signal originated in the interrogator increases. In the transponder 10 the signal captured by the antenna is injected on a line 28 cb to the S.O. chain 2 b. If the cascaded S.O. chain 2 b simply amplifies the input signal on the line 28 cb, then it will produce a signal on a line 28 bd that represents the phase and frequency of the signal injected on the line 28 cb. The signal on the line 28 bd is input to an antenna 30 a for transmission as the signal 12 back over the return distance from the transponder to the originating interrogator unit. The signal path from the transponder back to the originating interrogator will add an additional phase shift. The total phase shift as seen by the interrogator unit is the phase shift produced by the signal path from the interrogator to the transponder 10 plus the phase shift of the signal path from the transponder 10 back to the interrogator plus any phase shift introduced by the transponder. The distance or range between the interrogator and transponder is calculated by the observed phase shift of the receive signal phase relative to the transmit signal phase in the interrogator. The phase shift in the transponder 10 can be characterized and possibly discarded in the range calculation since this introduced phase shift may be a small portion of the total phase shift or not large enough to affect the accuracy of the range measurement. A phase shift is of course a time difference between two periodic signals measured between reference points on each signal and usually expressed in the angular displacement of degrees. Phase is a fraction of the period of a periodic waveform such as the wavelength of a signal. If only a phase measurement is made, and the distance between the interrogator and the transponder and back to the interrogator is longer than one wavelength, an ambiguity in the measurement exists as to how many additional wavelengths are present in the path between the interrogator and the transponder. Different frequencies have different wavelengths, so a different number of cycles between the interrogator and transponder will exist for a fixed distance for different frequencies. By using more than one known frequency of the signal in the system to determine the range, and by making a phase measurement at each frequency, the ambiguity can be resolved and a distance calculated. The two or more frequencies used must be chosen for a maximum range to be measured to ensure a non ambiguous measurement. The nature of removing phase ambiguities for range measurements is not the subject of the present invention so is not treated with any detail herein.
The invention is shown applied to a quadrature (IQ) modulator in FIG. 31. An IQ modulator controls both the amplitude and phase of a signal. It includes an input power divider called a quadrature divider 90 which divides the input signal (which might come from an antenna and/or an application) into two paths each having an amplitude and/or phase control element 92, 94, followed by a summer 96 which sums the outputs of the elements 92, 94 and provides an output signal (which might go to an antenna and/or an application). According to the present invention, synchronous oscillators are used for the elements 92, 94 instead of the typical components used in the prior art, i.e., PIN diode, Schottky diode or FET devices. The S.O.s 92, 94 of FIG. 31 may also be responsive to respective I and Q input signals as shown provided for example for mixing or control purposes, as the case may be. Any of the specialized S.O. circuits shown above with two inputs may be used for the S.O.s 92, 94, depending on the application. If the device is portable and supplied by a limited power supply, it may also employ a power control to the S.O.s 92, 94 such as the power controls shown in FIGS. 1�3.
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