Source: http://www.google.com/patents/US5847895?dq=6008737
Timestamp: 2014-11-26 10:05:25
Document Index: 599378190

Matched Legal Cases: ['art 218', 'art 226', 'art 230', 'art 230', 'art 234', 'art 238', 'art 238']

Patent US5847895 - To control a read/write head actuator in magnetic disk drive storage system - Google PatentsSearch Images Maps Play YouTube News Gmail Drive More »Sign inAdvanced Patent SearchPatentsA sliding mode controller is disclosed for controlling a read/write head actuator in a magnetic disk drive storage system wherein a voice coil motor connected to the actuator operates to adjust the position of the read/write head over a selected data track recorded on a magnetic medium. The sliding mode...http://www.google.com/patents/US5847895?utm_source=gb-gplus-sharePatent US5847895 - To control a read/write head actuator in magnetic disk drive storage systemAdvanced Patent SearchPublication numberUS5847895 APublication typeGrantApplication numberUS 08/943,997Publication dateDec 8, 1998Filing dateOct 3, 1997Priority dateMar 7, 1995Fee statusPaidPublication number08943997, 943997, US 5847895 A, US 5847895A, US-A-5847895, US5847895 A, US5847895AInventorsPaul M. Romano, Louis SupinoOriginal AssigneeCirrus Logic, Inc.Export CitationBiBTeX, EndNote, RefManPatent Citations (7), Non-Patent Citations (32), Referenced by (25), Classifications (12), Legal Events (4) External Links: USPTO, USPTO Assignment, EspacenetTo control a read/write head actuator in magnetic disk drive storage systemUS 5847895 AAbstract A sliding mode controller is disclosed for controlling a read/write head actuator in a magnetic disk drive storage system wherein a voice coil motor connected to the actuator operates to adjust the position of the read/write head over a selected data track recorded on a magnetic medium. The sliding mode controller comprises an improved technique for reducing chatter associated with the inherent operation of such a controller--rapid switching between gains to force the observable phase states to follow a predetermined phase state trajectory. The sliding mode controller generates a motor control command by multiplying an actuator position error and an actuator position error velocity by respective switching gains. The gains are switched according to a predetermined relationship σ between the phase states and a phase state trajectory. In order to reduce the switching frequency (i.e., chatter), a boundary layer around the phase state trajectory results in hysteresis, causing the gain blocks to switch only after the phase states cross over the boundary line. Further chatter reduction is achieved by generating the motor control signal proportional to an integral of sgn(σ). The cost of the sliding mode controller is reduced by implementing the phase state trajectory σ as a lookup table indexed by a single phase state.
We claim: 1. A sliding mode controller for controlling a read/write head actuator in a magnetic disk drive storage system wherein a motor connected to the actuator operates to adjust the position of the read/write head over a selected data track recorded on a magnetic medium, the sliding mode controller comprising:(a) a first input connected to receive an actuator position error signal X1, indicative of a difference between an estimated actuator position and a desired actuator position; (b) a first switching gain block for selectively multiplying a first phase state signal, responsive to the actuator position error signal X1, by a first gain or by a second gain according to a first predetermined relationship σ between the first phase state signal and a first phase plane trajectory, to generate a first proportional phase state signal; and (c) a multiplier for multiplying a computed control signal, responsive to the first proportional phase state signal, by a phase plane trajectory signal, responsive to the first predetermined relationship σ, to generate a motor control signal applied to the motor, wherein the multiplier attenuates high frequency components of the motor control signal. 2. The sliding mode controller as recited in claim 1, wherein a sign of the predetermined relationship σ is integrated to generate the phase plane trajectory signal.
3. The sliding mode controller as recited in claim 1, wherein the first phase state signal is the actuator position error signal X1.
4. The sliding mode controller as recited in claim 1, wherein the first phase state signal is an actuator position error velocity signal X2 generated by differentiating the actuator position error signal X1.
5. The sliding mode controller as recited in claim 1, further comprising:(a) a second input connected to receive an actuator velocity error signal Xv; (b) a second switching gain block for selectively multiplying a second phase state signal, proportional to the actuator velocity error signal Xv, by a third gain or by a fourth gain according to a second predetermined relationship σ between the second phase state signal and a second phase plane trajectory to generate a second proportional phase state signal; (c) a third input connected to receive an actuator acceleration signal Xα; and (d) a third switching gain block for selectively multiplying a third phase state signal, proportional to the actuator acceleration signal Xα, by a fifth gain or by a sixth gain according to a third predetermined relationship σ between the third phase state signal and the second phase plane trajectory to generate a third proportional phase state signal added with the first proportional phase state signal and the second proportional phase state signal to generate the computed control signal. 6. The sliding mode controller as recited in claim 1, further comprising an integrator for integrating the actuator position error signal X1 to generate an integrated signal added to the first proportional phase state signal to generate the computed control signal.
7. A sliding mode controller for controlling a read/write head actuator in a magnetic disk drive storage system wherein a motor connected to the actuator operates to adjust the position of the read/write head over a selected data track recorded on a magnetic medium, comprising:(a) a first switching gain block for selectively multiplying a phase state signal by a first gain or by a second gain according to a predetermined relationship σ between the phase state signal and a phase plane trajectory, to generate a motor control signal; (b) a predetermined boundary layer around the phase plane trajectory; (c) a means for computing the predetermined relationship σ relative to the boundary layer; and (d) an integrator for integrating a sign of the predetermined relationship σ to generate a phase plane trajectory signal,wherein the motor control signal is further responsive to the phase plane trajectory signal and the integrator attenuates high frequency components of the motor control signal. 8. A method for controlling a read/write head actuator in a magnetic disk drive storage system wherein a motor connected to the actuator operates to adjust the position of the read/write head over a selected data track recorded on a magnetic medium, the method comprising the steps of:(a) selectively multiplying a phase state by a first gain or by a second gain according to a predetermined relationship σ between the phase state and a phase plane trajectory, to generate a proportional phase state signal; and (b) multiplying a computed control signal, responsive to the proportional phase state signal, by a phase plane trajectory signal, responsive to the first predetermined relationship σ, to generate a motor control signal applied to the motor, wherein the step of multiplying attenuates high frequency components of the motor control signal. 9. The method for controlling a read/write head actuator as recited in claim 8, further comprising the step of integrating a sign of the predetermined relationship σ to generate the phase plane trajectory signal.
10. The method for controlling a read/write head actuator as recited in claim 8, wherein the phase state is an actuator position error X1.
11. The method for controlling a read/write head actuator as recited in claim 8, wherein the first and second gains are programmable.
12. A method for controlling a read/write head actuator in a magnetic disk drive storage system wherein a motor connected to the actuator operates to adjust the position of the read/write head over a selected data track recorded on a magnetic medium, comprising the step of selectively multiplying a phase state by a first gain or by a second gain according to a predetermined relationship σ between the phase state and a phase plane trajectory, wherein:(a) the phase plane trajectory has a predetermined boundary layer; (b) the predetermined relationship σ is computed relative to the boundary layer; and (c) integrating a sign of the predetermined relationship σ to generate a phase plane trajectory signal,wherein the motor control signal is further responsive to the phase plane trajectory signal and the step of integrating attenuates high frequency components of the motor control signal. Description
RELATED APPLICATION This application is a continuation, of application Ser. No. 08/507,621, filed Jul. 26, 1995, now abandoned, which is a continuation-in-part of U.S. patent application Ser. No. 08/438,937 filed May 10, 1995, now U.S. Pat. No. 5,699,207 which is a continuation of U.S. patent application Ser. No. 08/399,679 filed Mar. 7, 1995, now abandoned.
BACKGROUND OF THE INVENTION In magnetic disk storage systems, a transducing head writes digital data onto the surface of a magnetic storage medium. The digital data serves to modulate the current in a read/write head coil so that a sequence of corresponding magnetic flux transitions are written onto the magnetic medium in a series of concentric data tracks. To read this recorded data, the magnetic medium again passes under the read/write head which transduces the magnetic transitions into pulses in an analog signal. These pulses are then decoded by read channel circuitry to reproduce the digital data.
Conventional servo systems are typically linear controllers employing Proportional-Integral-Derivative (PID) feedback or state estimators. The problem with these types of linear controllers, however, is they are sensitive to parametric variations in the VCM control system and to external load disturbances. Conventional adaptive linear controllers overcome these sensitivity problems by continuously re-programming the controller to compensate for the parameter variations and load disturbances. Although adequate, adaptive control systems can be overly complex and expensive to implement. Further, adaptive linear controllers require notch filters to compensate for mechanical resonances. What is needed is a low cost, less complex solution to the sensitivity and resonance problems inherent in the control of a disk drive actuator.
By switching between positive and negative feedback gains, the sliding mode controller operates to drive the phase states toward a predetermined phase plane trajectory. The sliding mode controller is, therefore, a function of the phase states rather than the physical characteristics of the disk drive, and the control system is substantially insensitive to parametric variations and external load disturbances. Further, the notch filters in conventional linear controllers used to compensate for mechanical resonances are not necessary in the sliding mode controller of the present invention. It is relatively simple and inexpensive to implement, and there is a well defined method for proving global stability.
The present invention achieves further improvements in chatter reduction by defining a converging boundary layer around the phase plane trajectory σ. Still better performance is achieved by generating the control signal proportional to an integral of sgn(σ). In effect, the control signal is smoothed to attenuate the high frequency components that can generate electromagnetic and/or acoustic emissions.
FIG. 9 illustrates the converging boundary layer around the sliding mode phase plane trajectory.
FIG. 11 is the preferred lookup table embodiment of the a processing block.
The state space equation in positive feedback is: ##EQU4##
The solution to equation (5) is: ##EQU5##
Combining equations (5) and (6), ##EQU6## The phase plane plot of equation (7) is a set of hyperbolas with two asymptotes as shown in FIG. 3B.
The two individual phase state trajectories of equations (4) and (7) result in an unstable system since the phase states never reach the origin. It is possible, however, to reach the origin by driving the phase states along a third phase trajectory defined at the intersection of the negative and positive feedback trajectories. This is achieved by switching between the positive and negative gains in response to the current phase state values so that the phase states follow the predetermined third phase state trajectory.
The switching operation is understood with reference to FIG. 3C where the predetermined third phase state trajectory is shown as a linear segment 60. When a new track is selected, the initial actuator position error is at point A, and the control system is initially switched to select the positive gain (i.e., negative feedback). As the actuator begins to accelerate toward the selected track, the phase states follow the arc trajectory 64 of the negative feedback mode. When the phase states reach the beginning of the third phase state trajectory 60 at the intersection point B, the sliding mode controller switches to the negative gain and the phase states begin to follow the hyperbola trajectory 66 of the positive feedback mode. When the phase states cross the third phase state trajectory 60 at point C, the controller switches back to the positive gain to drive the phase states along arc 68 back toward the third phase state trajectory 60. This switching action is repeated so that the phase states slide along the linear segment 60 toward the origin of the phase plane. When the phase states are within a predetermined minimum distance from the origin of the phase plane, the system switches to a tracking mode where the sliding mode controller 26 repeatedly switches between positive and negative feedback in order to keep the phase states near the origin of the phase plane, thereby keeping the read/write head 6 aligned over the centerline of the selected track.
X1(t)=X1(t1)e-C (t-t1)                           (8)
where the constant C is the slope of the linear segment 60 (sliding line). By observing the phase states, the sliding mode controller switches the gains so that:
The sliding mode controller switches to the positive gain when σ�X1>0 and to the negative gain when σ�X1<0 in order to drive the phase states toward the linear phase state trajectory.
The overall response of the system is made faster by increasing the slope of the sliding line (i.e., increasing C). However, an important limitation in sliding mode control is that the third phase state trajectory must be constrained to a region in the phase plane where the positive and negative feedback phase state trajectories intersect in opposite directions. From FIG. 3C it follows that the slope of the sliding line must be constrained to 0<C<√K. A further relationship derived from this constraint is: ##EQU7## Equation (11) is known as the existence equation and is used to determine values for the positive and negative gains.
The linear sliding line trajectory 60 of FIG. 3C has the disadvantage in that it initially operates in a linear feedback mode, and the initial arc trajectory 64 may drift due to parameter variation and external load disturbance. This problem is reduced by extending the sliding mode region of operation. For instance, a phase trajectory adjustor can continuously adjust the slope of the linear segment as shown in FIG. 3D. After the phase states reach the first sliding line 65 and follow it for a predetermined amount of time, the phase trajectory adjustor increases the slope to sliding line 67 by increasing the constant C. The system operates in a linear mode (non-sliding mode) only during the inter-segment transitions. Eventually, the slope is increased to a predetermine maximum at sliding line 69 at which point the phase states slide along line 69 toward the origin of the phase plane.
&#963;2=X2-X2I; and                                       (12)
σ3=X2+C2�X1; where:
C2=the slope of the first segment;
The optimum phase plane trajectory, and the preferred embodiment of the present invention, is illustrated in FIG. 4. This trajectory comprises a substantially parabolic acceleration segment σ80, a linear constant velocity segment σ2 82, a second substantially parabolic deceleration segment σ3 84, and a linear deceleration segment σ4 86:
&#963;4=X2+C3�X1; where:                         (16)
Hardware Description FIG. 6 is a detailed diagram of the disk drive actuator sliding mode control system of the present invention. The actuator position error X1 22 is input into the sliding mode controller 26, and a differentiator 102 differentiates the actuator position error X1 22 to generate the actuator position error velocity signal X2 100. In an alternative embodiment not shown, the state estimator 14 generates the position error velocity X2. Two switching gain circuits 104 and 106 multiply the position error �X1 130 and error velocity �X2 132 control signals, respectively. Multipliers 108 and 110, responsive to the phase states �X1 and X2 and the current trajectory segment σi, control the switching operation of the gain circuits. The sign of the resulting multiplication determines the state of the switch so as to drive the phase states X1 and X2 toward the predetermined sliding line trajectory shown in FIG. 4. A σ processing block 112, responsive to the phase states X1 and X2, implements the trajectory segment switching logic to determine which segment σi of the phase plane trajectory the phase states are to follow. The operation of the σ processing block 112, the integrator 116, the reference error velocity generator 114, and multiplexers 118, 120, and 122, are discussed in detail bellow.
The gain values σi, βi, γi and ζi in switching gain blocks 104 and 106 are programmably set to appropriate values according to the current trajectory segment being followed by the phase states. Also, the gain values are programmed to predetermined values depending on whether the controller is executing a forward or reverse seek. Using the existence equation (11) and the phase state trajectory equations (13), (14), (15) and (16), the gain values for each segment of the phase state trajectory shown in FIG. 4 can now be computed.
For σ=σ2 (seek at constant velocity), differentiating equation (14) with respect to time and multiplying by equation (14) obtains: ##EQU15##
From equation (17): ##EQU16##
From equation (19): ##EQU17##
In order to satisfy existence equation (11) (i.e., equation (23) is negative for any X1 and X2), the gain constants must satisfy the following inequalities: ##EQU18##
For σ=σ3 (seek decelerate), differentiating equation (15) with respect to time and multiplying by equation (15) obtains: ##EQU19## From equation (17) and factoring σ1 �X2 obtains: ##EQU20## From equations (18) and (19), and ignoring term ##EQU21## as insignificantly small: ##EQU22## In order to satisfy existence equation (11) (i.e., equation (25) is negative for any X1 and X2), the gain constants must satisfy the following inequalities: ##EQU23##
For σ=σ4 (tracking), differentiating equation (16) with respect to time and multiplying by equation (16) obtains: ##EQU24## From equation (17): ##EQU25## From equation (19): ##EQU26## In order to satisfy existence equation (11) (i.e., equation (26) is negative for any X1 and X2), the gain constants must satisfy the following inequalities: ##EQU27##
Preferring now to FIG. 7B, at the beginning of SEEK ACCELERATE (σ=σ1) 208, the sliding mode controller initializes various parameters 210. The gain constants in blocks 104 and 106 of FIG. 6 are updated to the values corresponding to the acceleration trajectory σ=σ1 80. In order to reduce switching noise during a seek operation, the position error phase state X1 22 is switched out of the sliding mode control. The σ processing block 112 selects, over line 126, the ground plane as the output of multiplexer 122. As a result, �X1 130 is set to zero in order to disable the switching action of multiplier 110 and to remove the contribution of ψ1 from the computation of the VCM command U 28 at the output of adder 103. Because the position error phase state �X1 130 is disabled, the velocity phase state �X2 132 is initialized to a predetermined value to ensure the actuator begins moving in the desired direction (i.e., moving in reverse toward the selected track). To accomplish this, the σ processing block 112 selects, over line 124, X2Ref 114 as the output of multiplexer 120. The σ processing block 112 also selects, over line 126, the predetermined constant C 134 as the output of multiplexer 118 (the third input ψ3 109 to adder 103). The function of the predetermined constant C 134 and the integrator 116 are discussed in further detail bellow.
The σ processing block 112 continuously checks the location of the phase states with respect to the acceleration trajectory σi 80 to determine when to switch to the next trajectory segment. The next trajectory segment will either be the constant velocity segment σ2 82 or, if the seek distance is sufficiently short, the deceleration segment σ3 84. By comparing the σ values, the σ processing block 112 determines when to switch to the next trajectory. If σ1≦σ3? 220 is YES, then the σ processing block 112 switches to the deceleration trajectory σ3 84. Else if σ1≦σ2? 222 is YES, then the a processing block 112 switches to the constant velocity trajectory σ2 82. If both 220 and 222 are NO, then the sliding mode controller 26 loops around and computes the next VCM command U 28 according to flow chart 218.
Referring now to the constant velocity flow chart 226 shown in FIG. 7C, first the gain constants for switching gain blocks 104 and 106 are updated 228 to values corresponding to the constant velocity trajectory σ2 82 of FIG. 4. Then, in flow chart 230, the σ processing block 112 updates σ2 and σ3 according to equations (14) and (15), respectively. The output σi 128 of σ processing block 112 is assigned to σ2. Again, in response to σi and X2, multiplier 108 sets the state of switching gain block 104 in order to drive X1 and X2 toward the σ2 82 phase trajectory. The next command U 28 is generated and applied to the VCM 10 to continue moving the actuator 8 toward the selected track.
The σ processing block 112 continuously checks the location of the phase states with respect to the constant velocity trajectory σ2 82 to determine when to switch to the deceleration trajectory segment σ3 84. If σ2≦σ3? 232 is YES, then the a processing block 112 switches to the deceleration trajectory σ3 84. Otherwise, the sliding mode controller 26 loops around and computes the next VCM command U 28 according to flow chart 230.
Continuing now to the deceleration flow chart 234, first the gain constants for switching gain blocks 104 and 106 are updated 236 to values corresponding to the deceleration trajectory σ3 84 of FIG. 4. Then, in flow chart 238, the σ processing block 112 updates σ3 and σ4 according to equations (15) and (16), respectively. The output σi 128 of σ processing block 112 is assigned to σ3. Again, in response to σi and X2, multiplier 108 sets the state of switching gain block 104 in order to drive X1 and X2 toward the σ3 84 phase trajectory. The next command U 28 is generated and applied to the VCM 10 to decelerate the actuator 8 toward the selected track.
The σ processing block 112 continuously checks the location of the phase states with respect to the deceleration trajectory σ3 84 to determine when to switch to the tracking trajectory segment σ4 86. If σ4≦σ3?240 is YES, then the σ processing block 112 switches to the tracking trajectory σ4 86. Otherwise, the sliding mode controller 26 loops around and computes the next VCM command U 28 according to flow chart 238.
For reverse seeks, the sliding mode controller 26 operates as described in the flow charts of FIGS. 7A, 7B, and 7C except that the inequalities are reversed. The α processing block 112 can also adjust the slope of the linear phase trajectory segment as shown in FIG. 3D. An alternative embodiment of the σ processing block 112 would be to compare the position error and velocity phase states to values stored in a look up table where the stored values represent the phase plane trajectory shown in FIG. 4.
During seek accelerate and seek decelerate, a reference velocity Vref is generated as a function of the position error X1 corresponding to the velocity profiles σ1 80 and σ3 84 shown in FIG. 4. The reference velocity generator can be implemented with a lookup table or with polynomial equations. An actuator velocity error phase state Xv is generated by subtracting an estimated actuator velocity -X2 from the reference velocity Vref. An actuator acceleration phase state Xα is generated by taking the second derivative of the position error X1. Phase states Xv and Xα are multiplied by respective switching gain blocks to generate control signals ψ2 and ψ4. As discussed with reference to FIGS. 6 and 7, control signal ψ1 is disabled during seeks and ψ3 is insignificantly small. Therefore, the motor control signal U is a function of ψ2 and ψ4 during seek accelerate and seek decelerate. During seek at constant velocity (σ=σ2) and tracking (σ=σ4), Vref is set to zero such that Xv=X2, and σ4 is disabled by setting the gains δ and θ in the switching gain block to zero. In this manner, the sliding mode controller of FIG. 8 operates as described in FIGS. 6 and 7 during seek at constant velocity and tracking.
&#963;4=X2+C3�X1; where
σ1 to σ2 when |X21|>X2I;
σ3 to σ4 when |X1|< predetermined track acquire threshold; and
σ1 to σ3 when (X1*Xv<0) and (|X2|<|V2I|).
Lookup Table The σ processing block 112 of the present invention can be implemented using a lookup table rather than switching between trajectory segments. As mentioned previously, the phase states can be used as an index into a lookup table in order to implement the phase plane trajectory σ. To reduce the size of the table, the phase plane trajectory is redefined according to the following derivation:
distance=1/2�Acc�t2 implies
X1-X1I=1/2�Acc�t2 where:            (27)
Acc=a predetermined acceleration/deceleration constant; and
Solving for t in equation (27) provides:
t=(2�(X1-X1I)/Acc)1/2.                       (28)
Since velocity=-X2=Acc�t, then after substituting t from equation (28) provides:
X2(Ideal)=-(2�(X1-X1I)�Acc)1/2.     (29)
During seeks, the phase plane trajectory σ is defined as:
&#963;=X2+(2�(X1-X1I)�Acc)1/2 =0   (30)
Boundary Layer The sliding mode controller of the present invention provides further improvements in chatter reduction by defining a boundary layer around the phase state trajectory σ. This is illustrated in FIG. 9 which shows a linear segment for the phase plane trajectory σhaving a boundary layer defined as an offset �ε added to σ. The boundary layer reduces chatter by reducing the amount of switching in the system. Without the boundary layer, the switching gain blocks 104 and 106 of FIG. 6 will switch every time the phase states cross the sliding line (i.e., every time σ changes sign). The boundary layer results in hysteresis which causes the gain blocks to switch only after the phase states cross over the boundary line.
The boundary layer offset�ε added to the phase plane trajectory σ is a predetermined constant until the phase states reach a predetermined value (X1C, X2C) at which time the offset�ε is computed as the sum of the phase states XI and X2 so that the boundary layer converges to the origin of the phase plane, as shown in FIG. 9, in order to prevent oscillations around the origin. The σ processing block 112 of FIG. 6 computes σi as follows:
if switching gain blocks {104,106} are set to select gains {γi,αi} then
&#963;i=&#963;+&#949;;
else if switching gain blocks {104,106} are set to select gains
{ζi,βi} then
&#963;i=&#963;-&#949;;
&#963;=X2+C�X1;
&#949;=constant for X1&gt;X1C and X2&gt;X2C ;
&#949;=|X1|+|X2| for X1&#8806;X1C and X2&#8806;X2C.
In an alternative embodiment, rather than compute ε as the sum of X1 and X2 when X1≦X1C and X2≦X2C, the slope of the sliding line is changed (i.e., σ=X2+C1 �X1; or σ=X2+C2 �X1; depending on the current state of the switching gain blocks {104,106}).
Smoothing Function The sliding mode controller of the present invention achieves still better chatter reduction by generating the VCM control signal U proportional to an integral of sgn(σ). In effect, the control signal is smoothed to attenuate the high frequency components that can generate electromagnetic and/or acoustic emissions.
In an embodiment of the present invention, as shown in FIG. 10,σi from σ processing block 112 is input into an integrating block 101 which computes the following function: ##EQU28## The output 128 of the integrating block 101 controls the state of switching gain blocks {104, 106} and is also input into an absolute value function 111. The control signal at the output of adder 103 is then attenuated by the absolute value of the integrated sgn(σi) through a multiplier 113 to generate the smoothed VCM control signal U.
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