Source: http://www.google.com/patents/US5657356?dq=5,973,252
Timestamp: 2016-12-10 03:13:11
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Patent US5657356 - Control signal detection method with calibration error and subscriber unit ... - Google PatentsSearch Images Maps Play YouTube News Gmail Drive More »Sign inPatentsIn a control signal detection method, a calibration error value is obtained by obtaining the mean value of the received continuous (4×n) data, and the calibration error is compensated with respect to the received data by subtracting the obtained calibration error value from the received data, and the...http://www.google.com/patents/US5657356?utm_source=gb-gplus-sharePatent US5657356 - Control signal detection method with calibration error and subscriber unit therewithAdvanced Patent SearchTry the new Google Patents, with machine-classified Google Scholar results, and Japanese and South Korean patents.Publication numberUS5657356 APublication typeGrantApplication numberUS 08/670,748Publication dateAug 12, 1997Filing dateJun 21, 1996Priority dateJun 27, 1995Fee statusPaidAlso published asCN1098604C, CN1147187A, DE69633425D1, DE69633425T2, EP0751644A2, EP0751644A3, EP0751644B1, US5933465Publication number08670748, 670748, US 5657356 A, US 5657356A, US-A-5657356, US5657356 A, US5657356AInventorsShinji OzakiOriginal AssigneeSony CorporationExport CitationBiBTeX, EndNote, RefManPatent Citations (11), Referenced by (8), Classifications (14), Legal Events (4) External Links: USPTO, USPTO Assignment, EspacenetControl signal detection method with calibration error and subscriber unit therewith
US 5657356 AAbstract
In a control signal detection method, a calibration error value is obtained by obtaining the mean value of the received continuous (4×n) data, and the calibration error is compensated with respect to the received data by subtracting the obtained calibration error value from the received data, and the correlation value is obtained on the basis of the corrected received data, so that the control signal is detected. Therefore, it is able to compensate the calibration error with ease and to detect the control signal efficiently, with a simple construction.
1. A control signal detection method for detecting the control signal in received continuous (4×n) data, signal points of said control signal being rotated by 90° in the same direction on the same circle on a complex plane, comprising the steps of:obtaining a mean value of said received continuous (4×n) data existing on said complex plane, where n is a natural number, for obtaining a calibration error value for data demodulation; error-calibrating said received continuous data on the basis of the obtained calibration error value for compensating an error of said received continuous data; and obtaining a correlation value on the basis of the error-compensation received continuous data, thereby detecting said control signal. 2. The control signal detection method according to claim 1, whereinsaid step of obtaining a calibration error value includes obtaining a mean value of four data of said received continuous data. 3. The control signal detection method according to claim 1, whereinsaid step of obtaining a mean value includes using even-numbered data or odd-numbered data out of said received continuous (4×n) data. 4. A subscriber unit for receiving data transmitted from a base station and for detecting a control signal in received continuous (4×n) data, in which signal points in said control signal are rotated by 90° in the same direction on the same circle on a complex plane, comprising:calibration error value operation means for obtaining a mean value of said received continuous (4×n) data existing on said complex plane, where n is a natural number, for obtaining a calibration error value for data demodulation; error compensation means for error-calibrating with respect to said received continuous data on the basis of the calibration error value from said calibration error value operation means for compensating an error of said received continuous data; and control signal detection means for obtaining a correlation value on the basis of the error-calibrated received data, so that said control signal is detected. 5. The subscriber unit according to claim 4, whereinsaid calibration error value operation means includes means for obtaining the mean value of four data of said received continuous data. 6. The subscriber unit according to claim 4, whereinsaid calibration error value operation means includes means for obtaining the mean value by the use of even-numbered data or odd-numbered data out of said received continuous (4×n) data. 7. The subscriber unit according to claim 4, further comprisingmeans for performing timebase adjustment with respect to said subscriber unit on the timing of said control signal which has been detected by said control signal detection means. 8. The subscriber unit according to claim 4, further comprisingmeans for compensating the oscillation frequency of a local oscillator provided in said subscriber unit on the basis of the received data which has been error-corrected by said error correction means. Description
This invention relates to a control signal detection method with calibration error and a subscriber unit for use therewith, and is applicable to, for instance, the digital cellular telephone system which is referred to as the GSM (Groupe Speciale Mobile) cellular system and that is standardized in Europe.
Heretofore, in the GSM cellular system, a circuit is connected between a base station and mobile terminal equipment using the time division multiple access (TDMA) system, and voice data and others which have been coded are transmitted and received.
In the GSM cellular system, 124 broadcast channels are prepared as physical channels. Each broadcast channel is time-shared into 8 channels by the TDMA system. Logical channels are roughly separated into 2 channels, that is an information channel and a control channel. The information channel is, used to transmit coded voice data, etc., and the control channel is used to transmit various control signals.
The control channel includes a broadcast control channel (BCCH), a frequency correction channel (FCCH), a synchronization channel (SCH), a paging channel (PCH), a random access channel (RACH), attendant control channels, namely a slow associated control channel (SACCH), and a fast associated control channel (FACCH), etc.
In the GSM cellular system, the Gaussian filtered minimum shift keying (GMSK) is used as a modulating system, and each data is exposed to GMSK modulation and then transmitted. An example of the GMSK modulation has been disclosed in U.S. Pat. No. 5,131,008, assigned to Motorola Inc. In this patent, compensation of an in-phase signal and a quadrature signal is performed based on a coherent carrier signal.
By the way, to perform connection of a circuit by means of the time division multiple access, synchronization must be established between a base station and mobile terminal equipment. Therefore, after the electric power source has been energized, the mobile terminal equipment first detects the FCCH in the control channel which is transmitted by the base station, and then roughly realizes the initial synchronization on the basis of the detected FCCH and corrects the oscillating frequency of the local oscillator (hereinafter, this is referred to as the local oscillation frequency). However, the fine synchronism is established by the means of the SCH.
As shown in FIG. 1, one period of the control channel is comprised of 51 frames, and each frame is comprised of 8 slots. On the GSM cellular system, burst data in each slot are transmitted and received. On the control channel which has a constitution like this, the FCCH is inserted once (1 slot) within 10 frames. The FCCH is a control signal which is composed of continuous [0]s of the stated bits, and the data are not varied to [0] or to [1], as contrasted to the other burst data. Therefore, when the received data of the case where the FCCH has been normally received is represented on a complex plane, the signal points are respectively rotated by 90° in the same direction, as shown in FIG. 2. This is due to the fact that when the transmitted data are the series of a same values, the signal points are respectively rotated by 90° in the same direction on the same circle on a complex plane, in the case of GMSK modulation.
To detect such a FCCH, the mobile terminal equipment performs a calculation which is represented by the following expression: ##EQU1## with respect to the I,Q data (ik, qk) which is the received data, and obtains the correlation value. More specifically, the mobile terminal equipment is provided with a correlation value calculating circuit 1 shown in FIG. 3, so that the correlation value is obtained by this.
The received I,Q data (ik, qk) is first fed to a multiplier 2, and multiplied, by e-jp which is outputted from a numeric value generator 3 (where, P=πk/2). The result of this multiplication is fed to an N-stage shift register 4, and shifted sequentially. N pieces of data which are output from the shift register 4 are respectively fed to an adder 5, and added. The result of the addition is fed into an absolute value circuit 6, so that the correlation value is obtained.
This obtained correlation value becomes a large value in the case where the received data is FCCH, and becomes a small value in the other cases, as shown in FIG. 4. Therefore, if the magnitude of the correlation value is continuously examined, the FCCH can be detected on the basis of such a fact that the correlation values have reached to the maximum value.
By the way, to obtain the correlation value, at first, standard adjustment (hereinafter, this is referred to as calibration) must be performed in such a manner that the center of the I,Q data comes to the position of the origin (0, 0) on the complex plane, prior to the start of receiving. Because, some signals are occasionally output owing to the characteristics of the circuits of the receiving system, even though it is in the non-receiving status. In order to compensate this, it is necessary that the center of the I,Q data be adjusted to the position of the origin on the complex plane. When such calibration is performed, heretofore, an analog calibration circuit shown in FIG. 5 is utilized.
In this calibration circuit 10, in the first place, at non-receiving time, switches SW1 and SW2 are brought into OFF state, and a switch SW3 is brought into ON state. By this, the difference of the I component and the -I component is charged in a capacitor C1, through a differential amplifier 11.
On the other hand, when receiving has been started, in the calibration circuit 10, the switches SW1 and SW2 are brought into ON state, and a switch SW3 is brought into OFF state. By this, compensation is performed with respect to the I component and the -I component, by the difference of non-receiving time, by means of current sources A1 to A4 and a differential amplifier 12, according to the electric charge which has been charged in the capacitor C1. In this connection, compensation is also performed in a similar manner with respect to the Q component.
By the way, in the GSM cellular system, only one slot of FCCH exists in 10 frames as stated above; therefore, continuous receiving operation of a long time (about 50 [mS] which is corresponding to about 11 frames) is needed, until an FCCH is detected. In this case, such a phenomenon occurs that the compensation comes to be not performed correctly, because the electric charge that has been charged in the capacitor C1 of the abovementioned calibration circuit 10 is discharged. As a result, so-called calibration error occurs, which is such a phenomenon that the calibration which has been performed prior to the start of reception drifts in proportion as reception advances, as shown in FIG. 6.
Such a calibration error has bad influences; the probability of detection of FCCH is lowered, an error occurs in the estimate which is estimated when the drift of the local oscillation frequency (hereinafter, this is referred to as local oscillation frequency offset) would be calibrated, and others. Explaining it concretely, because a correlation value is used in order to detect FCCH as stated above, if a calibration error exists then the correlation value becomes small, therefore the probability of detection of FCCH is lowered.
When the local oscillation frequency offset is estimated, it is estimated on the basis of the quantity of phase drift (Δθ) of the I,Q data, as shown in FIG. 7A. If the calibration has drifted, then a phase error (=Δθ'-Δθ) occurs as shown in FIG. 7B, so that an error occurs in the estimate of the local oscillation frequency offset as well.
In view of the foregoing, an object of this invention is to provide a control signal detection method which is able to detect the control signal efficiently and to compensate a calibration error easily with a simple construction, and a subscriber unit which utilizes the method.
The foregoing object and other objects of this invention have been achieved by the provision of a control signal detecting method in which a calibration error is obtained by obtaining the mean value of the received continuous (4×n) data, the calibration error is compensated toward the received data by subtract the obtained calibration error from the received data, and the correlation value is obtained on the basis of the corrected received data, so that the control signal is detected.
In the case where the received data are rotated by 90° in the same direction on a complex plane, and where there is no calibration error, the mean value of the received continuous (4×n) data becomes zero. In other words, the center of the received continuous (4×n) data lies at the origin of the complex plane. However, if there is a calibration error, the mean value of the received continuous (4×n) data does not become zero. In other words, the center of the received continuous (4×n) data lies at a position which is shifted from the origin of the complex plane. This discrepancy with the origin of the complex plane represents the calibration error. Therefore, the calibration error can be obtained by obtaining the mean value of the received continuous (4×n) data.
By subtracting the calibration error which has been thus obtained from the received data, the calibration error can be easily compensated with reference to the received data. When the correlation value is obtained on the basis of the received data which have been thus corrected, it can be avoided that the correlation value is diminished as usual due to a calibration error, and so the control signal can be detected efficiently and certainly.
FIG. 1 is a schematic diagram illustrating the structure of the control channel;
FIG. 2 is a schematic diagram showing the result of reception of the FCCH;
FIG. 3 is a block diagram illustrating the correlation value calculating circuit;
FIG. 4 is a schematic diagram showing the change of the correlation value;
FIG. 5 is a connection diagram illustrating the calibration circuit;
FIG. 6 is a schematic diagram showing the calibration error;
FIGS. 7A and 7B are schematic diagrams showing the influence of the calibration error;
FIG. 8 is a block diagram illustrating mobile terminal equipment of the GSM cellular system according to an embodiment of this invention; and
FIG. 9 is a block diagram illustrating the calibration error compensating circuit.
In FIG. 8, the reference numeral 20 generally designates mobile terminal equipment of the GSM cellular system. The mobile terminal equipment 20 receives through an antenna 21 the transmission signal which has been issued from the base station, and inputs the resulted received-signal to a duplexer 23 of a radio frequency (RF) block 22. The duplexer 23 is an antenna sharing device which enables transmitting and receiving operations to share the antenna 21, and outputs the inputted received-signal to a low-noise amplifier 24.
The low-noise amplifier 24 amplifies the inputted received-signal and outputs it to a filter 25. The filter 25 limits the band of the received signal to eliminate the unnecessary components, and then outputs the received signal that has been band limited to a mixer 26. Using a local-oscillation signal which has been generated in an oscillator 28 and supplied through a buffer 27, the mixer 26 performs frequency conversion of the received signal to change it into a reception intermediate-frequency signal. This means that the received signal of 900 [MHz] of RF frequency which has entered at the antenna 21 is exposed to down-conversion into the intermediate-frequency signal.
In this connection, in the mobile terminal equipment 20, the desired broadcast channel can be received selectively, by switching the frequency of the local-oscillation signal which is generated in the oscillator 28.
The reception intermediate-frequency signal which has been obtained with the mixer 26 is exposed to band-limitation at a filter 29, and then amplified by an amplifier 30, and inputted to a demodulator 31. The demodulator 31 demodulates the reception intermediate-frequency signal using an oscillated signal which has been generated by an oscillator 32, so that the I-signal (IR) and the Q-signal (QR) which are respective orthogonal signal components are obtained. The I-signal and the Q-signal are inputted to amplifiers 34 and 35 of a base-band block 33, respectively.
The amplifier 34 and the amplifier 35 amplify respectively the I-signal and the Q-signal which have been inputted respectively, and output the amplified I-signal and the amplified Q-signal to a filter 36 and a filter 37 respectively. The filter 36 and the filter 37 eliminate unnecessary components, leaving behind only the modulated wave components of the I-signal and the Q-signal respectively, and output the resulted I-signal and Q-signal to analog-to-digital (A/D) converters 38 and 39 respectively. The A/D converters 38 and 39 perform analog-to-digital conversion of the inputted I-signal and Q-signal respectively, on the basis of the stated sampling clock, and deliver respectively the resulting I data (i) and Q data (q) to an equalizer 41 of a digital signal processor (DSP) block 40.
The equalizer 41 performs correction of distortion with respect to the I data and the Q data to eliminate influences of multipath, etc., and restores the data which have been transmitted by the I data and the Q data, and then outputs it as the transmitted data. A channel decoder 42 and a voice decoder 43 extract the voice data from the transmitted data which are outputted from the equalizer 41, and output the voice data to a digital-to-analog (D/A) converter 44. In this connection, processing of each block in the DSP block 40 is performed basically by software installed in the equipment.
By converting the voice data into an analog voice signal by means of the D/A converter 44 in this way, the voice which is corresponding to the voice signal is outputted from a speaker 45.
On the other hand, in the case of transmission of, for instance, a voice signal in the mobile terminal equipment 20 the voice signal is exposed to orthogonal modulation through an analog-to-digital converter, a voice encoder, a channel encoder and so on none of which are shown in FIG. 8, to be converted into the I data and the Q data, and then these are converted into analog signals by digital-to-analog converters not shown, so that the I signal (IT) and the Q signal (QT) are generated. Then, the generated I signal and the Q signal are inputted to a modulator 46, and the oscillated signal, which is a carrier wave and which has been produced by oscillators 28 and 47, a buffer 48 and a mixer 49, is also inputted to the modulator 46. The modulator 46 performs modulation with respect to the oscillated signal on the basis of the inputted I signal and the Q signal, and outputs the resulted transmission signal to a filter 50.
The filter 50 limits the band of the transmission signal and eliminates the unnecessary components. The transmission signal which is outputted from the filter 50 is amplified by a power amplifier 51, and then introduced into the antenna 21 by the duplexer 23, and radiated in the air.
In this connection, the operation of the power amplifier 51 can be controlled by a power-amplifier controller 52, so that switching of the transmission output can be controlled according to the base station which is used, and transmission can be stopped at a time other than transmission timing.
In addition, by switching the frequency of the local oscillation signal which is generated in the oscillator 28, selective transmission in the desired broadcast channel is enabled.
By the way, the mobile terminal equipment 20 communicates with the base station using the TDMA system, therefore it must be synchronized with the base station at the time of various operations such as transmission or reception. Therefore, after the power has been turned on, the mobile terminal equipment 20 detects, at first, the FCCH in the control channel which is transmitted by the base station, and then roughly realizes the initial synchronization on the basis of the detected FCCH, and also compensates the local oscillation frequency offset.
More specifically, after the power has been turned on, the mobile terminal equipment 20 receives, at first, the control channel which is transmitted by the base station. Then, on the basis of the resulting I data and the Q data, the equipment 20 detects the FCCH by an FCCH detecting part 53. Upon detecting the FCCH in the control channel, the FCCH detecting part 53 outputs the result of the detection to a system timebase part 54, and also outputs the quantity of phase drift see FIGS. 7A and 7B) which is obtained from the I data and the Q data to a local-oscillation-frequency offset estimate/compensate part 55.
The system timebase part 54 is one for performing time management of the operation of the entire system; and the part 54 resets a timer, etc., according to the result of the detection which is outputted by the FCCH detecting part 53, in other words, the timing of the FCCH, and performs time management of the entire system, and synchronizes various operations with the base station.
The local-oscillation-frequency offset estimate/compensate part 55 estimates a drift of the local oscillation frequency toward the base station on the basis of the quantity of phase drift which is outputted from the FCCH detecting part 53, and compensates the drift of the local oscillation frequency in accordance with the estimated value.
By the way, in the FCCH detecting part 53, by performing the calculation shown in equation (1) toward the I data and the Q data as usual, the correlation value is obtained and the FCCH is detected, and the quantity of phase drift is obtained on the basis of the I data and the Q data. At this time, the FCCH detecting part 53 digitally processes the occurred calibration error, and eliminates it from the I data and the Q data. By this, bad influences of calibration error can be avoided, in the mobile terminal equipment 20.
A correcting method for eliminating a calibration error in a digitalized manner is explained hereinafter.
As stated above, the FCCH is a control signal which is composed of continuous [0]s of the stated bits, therefore the received I,Q data are rotated by 90° in the same direction on the same circle on a complex plane (see FIG. 2). Accordingly, when the continuous 4 samples of I,Q data are designated as (ik, qk) (where k=0, 1, 2, 3), and if there is no calibration error, each sum becomes [0] as shown in the following equation: ##EQU2##
In other words, the center of the continuous 4 samples of I,Q data is located at the origin (0, 0) on the complex plane. Utilizing this point, a calibration error is eliminated in a digitalized manner, in the case of this embodiment.
First, the I,Q data which include a calibration error are designated as (in, qn), and sum of the continuous 4 samples of I,Q data is obtained, and then the obtained sum is divided by the number of the samples (in other words, the mean value of the continuous 4 samples is found), as shown in the following equation: ##EQU3##
Then, as shown in the following equation:
(i'n, q'n)=(in, qn)-(&#916;in, &#916;qn)(4)
the mean value (Δin, Δqn) of the continuous 4 samples which has been obtained in the abovementioned equation (3) is subtracted from the I,Q data (in, qn) which include a calibration error, so that the I, Q data (in, qn) are converted into the I, Q data (i'n, q'n).
In the I,Q data (i'n, q'n) which have been converted at this time, a calibration error is not included. Hereinafter, this point is explained.
First, the I,Q data which do not include a calibration error are designated as (i"n, q"n), and the calibration error is designated as (ei, eq). Where, it is assumed that the value of the calibration error is same, with respect to a short period. On such an assumption, the I,Q data (in, qn) which include a calibration error appear as the following equation:
(in, qn)=(i"n +ei, q"n +eq)  (5)
Substituting this equation (5) for the abovementioned equation (3), we have the following equation: ##EQU4##
This equation (6) can be reduced into the following equation:
(&#916;in, &#916;qn)=(ei, eq)        (7)
on the basis of the abovementioned equation (2).
In other words, the mean value of the continuous 4 samples becomes the calibration error itself, as will be seen from this equation (7). Therefore, it is known that the I,Q data (i'n, q'n) which have been obtained by the abovementioned equation (4) is the data which do not include a calibration error, as shown in the following equation:
(i'n, q'n)=(i"n +ei, q"n +eq)-(ei, eq)=(i"n, q"n)                             (8)
In this way, in the FCCH detecting part 53, the mean value of the continuous 4 samples of the I,Q data is obtained to obtain the calibration error, and then the obtained calibration error is subtracted from the I,Q data, so that the calibration error is eliminated. By this, it is able to avoid bad influences based on a calibration error, in the mobile terminal equipment 20.
In this connection, as to the data other than FCCH, it is not rotated by 90° in the same direction, therefore the calibration error can not be eliminated with such a correcting method; however, it is not a problem, because detection of the FCCH is the object in the case of this embodiment.
On this subject, a calibration error compensating circuit shown in FIG. 9 is provided, in practice, in the FCCH detecting part 53, so as to realize the abovementioned equation (4). As shown in this FIG. 9, in the calibration error compensating circuit 60, the I data (i) which has been issued from the A/D converter 38 of the baseband block 33 is inputted to a 4-stage shift-register 61, and shifted sequentially at this place. Therefore, in the respective registers of the shift register 61, the continuous 4 samples of the I data (in+3, in+2, in+1 and in) are arranged.
This continuous 4 samples of the I data are inputted to a mean value calculation part 62, respectively. The mean value calculation part 62 finds the sum of the inputted four I data, and then divides the found sum by the number of the samples, so as to obtain the mean value of the continuous 4 samples of the I data (that is, the calibration error), and then outputs the obtained mean value to an adder 63.
The adder 63 subtracts the mean value which is outputted from the mean value calculation part 62 from the I data (in) which is outputted from the shift register 61, and thereby obtains the I data (i') which does not include a calibration error, and outputs it.
Similarly, in the calibration error compensating circuit 60, the Q data (q) which has been issued from the A/D converter 39 of the baseband block 33 is inputted to a 4-stage shift-register 64, and shifted sequentially at this place. Therefore, in the respective registers of the shift register 64, the continuous 4 samples of the Q data (qn+3, qn+2, qn+1 and qn) are arranged.
This continuous 4 samples of the Q data are inputted to a mean value calculation part 65, respectively. The mean value calculation part 65 finds the sum of the inputted four Q data, and then divides the found sum by the number of the samples, so as to obtain the mean value of the continuous 4 samples of the Q data (that is, the calibration error), and then outputs the obtained mean value to an adder 66.
The adder 66 subtracts the mean value which is outputted from the mean value calculation part 65 from the Q data (qn) which is outputted from the shift register 64, and thereby obtains the Q data (q') which does not include a calibration error, and outputs it.
So, in the FCCH detecting part 53, on the basis of the I data (i') and the Q data (q') of which calibration errors have been eliminated and which have been obtained by means of the calibration compensating circuit 60, the correlation value is found to detect the FCCH, and the quantity of phase drift which is used at the time of compensation of the local-oscillation-frequency offset is also found.
In the above construction, on the mobile terminal equipment 20, in order to realize synchronization with the base station, after the electric power source has been energized, a control channel which is transmitted by the base station is first received, and the FCCH in the control channel is detected by the use of the FCCH detecting part 53. At that time, the FCCH detecting part 53 finds the calibration error by finding the mean values of the I data and the Q data of the continuous 4 samples, and subtracts the found calibration error from the received I data and the received Q data to correct the I data and the Q data. Then, the FCCH detecting part 53 finds the correlation value on the basis of the corrected I data and the Q data so as to detect the FCCH, and also finds the quantity of phase drift which is used at the time of compensation of the local-oscillation-frequency offset.
Hereby, in the mobile terminal equipment 20, it can be avoided that the correlation value is lowered due to a calibration error and the provability of detecting the FCCH is hereby lowered as in the past, and the FCCH can be detected efficiently. Besides, in the mobile terminal equipment 20, it can be avoided that an error is generated in the quantity of phase drift due to a calibration error as in the past, and the local oscillation frequency offset can be compensated surely.
In this connection, taking a calibration utilizing an analog calibration circuit (see FIG. 5) has been usually performed in the past; but it has been very difficult to maintain such a state for a long time, after the calibration has been taken. However, in the case of this embodiment, even though such a state that the calibration has been taken can not be maintained for a long time, the FCCH can be detected efficiently owing to the fact that compensation is performed in a digitalized manner. Besides, if a calibration error is compensated ultimately in a digitalized manner like this embodiment, then an analog calibration can be omitted from the beginning.
According to the above construction, the FCCH detecting part 53 is provided which finds the calibration error on the basis of the mean values of the I data and the Q data of the continuous 4 samples, corrects the I data and the Q data using the found calibration error, and detects the FCCH on the basis of the corrected I data and the Q data. As a result, the calibration error can be easily compensated with a simple construction and the FCCH can be detected efficiently.
In the aforementioned embodiment, the calibration error has been obtained on the basis of the mean value of the continuous 4 samples of the I,Q data, however, the same effects as the above can be also obtained in the case where a calibration error is found on the basis of the mean value of the continuous (4×n) samples of the I,Q data, where n is a natural number (that is, n =0, 1, 2, . . . ), as the other embodiment. Besides, in this embodiment a calibration error may be obtained on the basis of the mean value of only the even-numbered I,Q data or only the odd-numbered I,Q data out of the continuous (4×n) samples. In brief, a calibration error can be obtained also by obtaining a calibration error using I,Q data which are located symmetrically with respect to the origin (0, 0) on the complex plane.
In the aforementioned embodiments, the quantity of phase drift is found by the FCCH detecting part 53, and the drift of the local oscillation frequency is estimated and compensated by the local-oscillation-frequency offset estimate and compensate part 55, however, this invention is not limited to this but the same effects as the above can be also obtained in the case where the quantity of phase drift is found and also the drift of the local oscillation frequency is estimated, on the basis of the found quantity of phase drift, and compensated, in the local-oscillation-frequency offset estimate and compensate part 55, on the basis of the I,Q data which has been corrected by the FCCH detecting part 53.
Further, in the aforementioned embodiments, this invention has been applied to detection of the FCCH in the GSM cellular system, however, this invention is not limited to this but is widely applicable to detection of a control signal wherein the received data are rotated by 90° in the same direction on the same circle on a complex plane, at the time of reception.
Patent CitationsCited PatentFiling datePublication dateApplicantTitleUS3680056 *Oct 8, 1970Jul 25, 1972Bell Telephone Labor IncUse equalization on closed loop message block transmission systemsUS4590519 *May 4, 1983May 20, 1986Regency Electronics, Inc.Television signal scrambling/descrambling systemUS4590524 *Nov 18, 1983May 20, 1986Hitachi, Ltd.Multitrack PCM reproducing apparatusUS4606051 *Nov 10, 1983Aug 12, 1986Universal Data Systems, Inc.QPSK demodulator with I and Q post-detection data correctionUS4819646 *Aug 18, 1986Apr 11, 1989Physio-Control CorporationFeedback-controlled method and apparatus for processing signals used in oximetryUS5245794 *Apr 9, 1992Sep 21, 1993Advanced Micro Devices, Inc.Audio end point detector for chemical-mechanical polishing and method thereforUS5442655 *May 27, 1993Aug 15, 1995Fujitsu LimitedDC cancellation and restoration in receiving apparatusUS5528382 *Nov 14, 1994Jun 18, 1996Canon Kabushiki KaishaReproduction apparatus for video signals accompanied by control informationUS5550759 *Aug 7, 1995Aug 27, 1996The United States Of America As Represented By The Secretary Of The NavyAdaptive parameter kernel processorUS5579351 *Oct 19, 1995Nov 26, 1996Lg Information & Communications, Ltd.Jitter suppression circuitUS5594756 *Jun 20, 1995Jan 14, 1997Hitachi, Ltd.Decision feedback equalization circuit* Cited by examinerReferenced byCiting PatentFiling datePublication dateApplicantTitleUS5909433 *Aug 30, 1996Jun 1, 1999Telefonaktiebolaget L M Ericsson (Publ)Method and apparatus for acquiring low duty-cycle reference signals in a mobile communications environmentUS5933465 *Mar 28, 1997Aug 3, 1999Sony CorporationControl signal detection method with calibration error and subscriber unit therewithUS5940439 *Feb 26, 1997Aug 17, 1999Motorola Inc.Method and apparatus for adaptive rate communication systemUS6038270 *May 21, 1997Mar 14, 2000Sony CorporationRadio receiver apparatus and radio receiving methodUS6192087Nov 15, 1996Feb 20, 2001Conexant Systems, Inc.Method and apparatus for spectral shaping in signal-point limited transmission systemsUS6278744Nov 27, 1996Aug 21, 2001Conexant Systems, Inc.System for controlling and shaping the spectrum and redundancy of signal-point limited transmissionUS6765974 *Jul 19, 2000Jul 20, 2004Radwin Ltd.Rach starting time vicinity estimationUS20060066431 *Oct 5, 2004Mar 30, 2006Anand Seema BAdjustable differential inductor* Cited by examinerClassifications U.S. Classification375/346, 375/349International ClassificationH04L25/06, H04L27/20, H04L1/24, H04L27/14, H04B7/26, H04B7/005Cooperative ClassificationH04L27/20, H04L1/24, H04L25/062European ClassificationH04L25/06A1, H04L1/24, H04L27/20Legal EventsDateCodeEventDescriptionNov 6, 1996ASAssignmentOwner name: SONY CORPORATION, JAPANFree format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:OZAKI, SHINJI;REEL/FRAME:008370/0123Effective date: 19961021Feb 9, 2001FPAYFee paymentYear of fee payment: 4Feb 14, 2005FPAYFee paymentYear of fee payment: 8Sep 30, 2008FPAYFee paymentYear of fee payment: 12RotateOriginal ImageGoogle Home - Sitemap - USPTO Bulk Downloads - Privacy Policy - Terms of Service - About Google Patents - Send FeedbackData provided by IFI CLAIMS Patent Services