Source: https://patents.google.com/patent/US9711852B2/en
Timestamp: 2018-10-20 05:30:55
Document Index: 688912021

Matched Legal Cases: ['Application No. 61', 'Application No. 61', 'Application No. 61', 'Application No. 62', 'Application No. 2013027842', 'Application No. 2011314378', 'art 2', 'Application No. 2013119332', 'Application No. 201180055705', 'Application No. 201180055705']

US9711852B2 - Modulation patterns for surface scattering antennas - Google Patents
Modulation patterns for surface scattering antennas Download PDF
US9711852B2
US9711852B2 US14549928 US201414549928A US9711852B2 US 9711852 B2 US9711852 B2 US 9711852B2 US 14549928 US14549928 US 14549928 US 201414549928 A US201414549928 A US 201414549928A US 9711852 B2 US9711852 B2 US 9711852B2
US14549928
US20150372389A1 (en )
Pai-Yen Chen
Siamak Ebadi
John Desmond Hunt
Nathan Ingle Landy
Melroy Machado
Milton Perque, JR.
H01Q13/20—Non-resonant leaky-waveguide or transmission-line antennas; Equivalent structures causing radiation along the transmission path of a guided wave
Modulation patterns for surface scattering antennas provide desired antenna pattern attributes such as reduced side lobes and reduced grating lobes.
The present application constitutes a continuation-in-part of U.S. patent application Ser. No. 14/510,947, entitled MODULATION PATTERNS FOR SURFACE SCATTERING ANTENNAS, naming Pai-Yen Chen, Tom Driscoll, Siamak Ebadi, John Desmond Hunt, Nathan Ingle Landy, Melroy Machado, Milton Perque, Jr., David R. Smith, and Yaroslav Urzhumov as inventors, filed 9, Oct. 2014, which is currently co-pending or is an application of which a currently co-pending application is entitled to the benefit of the filing date.
U.S. Patent Application No. 61/455,171, entitled SURFACE SCATTERING ANTENNAS, naming NATHAN KUNDTZ ET AL. as inventors, filed Oct. 15, 2010, is related to the present application.
U.S. patent application Ser. No. 13/317,338, entitled SURFACE SCATTERING ANTENNAS, naming ADAM BILY, ANNA K. BOARDMAN, RUSSELL J. HANNIGAN, JOHN HUNT, NATHAN KUNDTZ, DAVID R. NASH, RYAN ALLAN STEVENSON, AND PHILIP A. SULLIVAN as inventors, filed Oct. 14, 2011, is related to the present application.
U.S. patent application Ser. No. 13/838,934, entitled SURFACE SCATTERING ANTENNA IMPROVEMENTS, naming ADAM BILY, JEFF DALLAS, RUSSELL J. HANNIGAN, NATHAN KUNDTZ, DAVID R. NASH, AND RYAN ALLAN STEVEN as inventors, filed Mar. 15, 2013, is related to the present application.
U.S. Patent Application No. 61/988,023, entitled SURFACE SCATTERING ANTENNAS WITH LUMPED ELEMENTS, naming PAI-YEN CHEN, TOM DRISCOLL, SIAMAK EBADI, JOHN DESMOND HUNT, NATHAN INGLE LANDY, MELROY MACHADO, MILTON PERQUE, DAVID R. SMITH, AND YAROSLAV A. URZHUMOV as inventors, filed May 2, 2014, is related to the present application.
U.S. patent application Ser. No. 14/506,432, entitled SURFACE SCATTERING ANTENNAS WITH LUMPED ELEMENTS, naming PAI-YEN CHEN, TOM DRISCOLL, SIAMAK EBADI, JOHN DESMOND HUNT, NATHAN INGLE LANDY, MELROY MACHADO, JAY MCCANDLESS, MILTON PERQUE, DAVID R. SMITH, AND YAROSLAV A. URZHUMOV as inventors, filed Oct. 3, 2014, is related to the present application.
U.S. Patent Application No. 61/992,699, entitled CURVED SURFACE SCATTERING ANTENNAS, naming PAI-YEN CHEN, TOM DRISCOLL, SIAMAK EBADI, JOHN DESMOND HUNT, NATHAN INGLE LANDY, MELROY MACHADO, MILTON PERQUE, DAVID R. SMITH, AND YAROSLAV A. URZHUMOV as inventors, filed May 13, 2014, is related to the present application.
The present application claims benefit of priority of U.S. Provisional Patent Application No. 62/015,293, entitled MODULATION PATTERNS FOR SURFACE SCATTERING ANTENNAS, naming PAI-YEN CHEN, TOM DRISCOLL, SIAMAK EBADI, JOHN DESMOND HUNT, NATHAN INGLE LANDY, MELROY MACHADO, MILTON PERQUE, DAVID R. SMITH, AND YAROSLAV A. URZHUMOV as inventors, filed Jun. 20, 2014, which was filed within the twelve months preceding the filing date of the present application.
All subject matter of all of the above applications is incorporated herein by reference to the extent such subject matter is not inconsistent herewith.
FIG. 1 is a schematic depiction of a surface scattering antenna.
FIGS. 2A and 2B respectively depict an exemplary adjustment pattern and corresponding beam pattern for a surface scattering antenna.
FIGS. 3A and 3B respectively depict another exemplary adjustment pattern and corresponding beam pattern for a surface scattering antenna.
FIGS. 4A and 4B respectively depict another exemplary adjustment pattern and corresponding field pattern for a surface scattering antenna.
FIGS. 5A-5F depict an example of hologram discretization and aliasing.
FIG. 6 depicts a system block diagram.
A schematic illustration of a surface scattering antenna is depicted in FIG. 1. The surface scattering antenna 100 includes a plurality of scattering elements 102 a, 102 b that are distributed along a wave-propagating structure 104. The wave propagating structure 104 may be a microstrip, a coplanar waveguide, a parallel plate waveguide, a dielectric rod or slab, a closed or tubular waveguide, a substrate-integrated waveguide, or any other structure capable of supporting the propagation of a guided wave or surface wave 105 along or within the structure. The wavy line 105 is a symbolic depiction of the guided wave or surface wave, and this symbolic depiction is not intended to indicate an actual wavelength or amplitude of the guided wave or surface wave; moreover, while the wavy line 105 is depicted as within the wave-propagating structure 104 (e.g. as for a guided wave in a metallic waveguide), for a surface wave the wave may be substantially localized outside the wave-propagating structure (e.g. as for a TM mode on a single wire transmission line or a “spoof plasmon” on an artificial impedance surface). It is also to be noted that while the disclosure herein generally refers to the guided wave or surface wave 105 as a propagating wave, other embodiments are contemplated that make use of a standing wave that is a superposition of an input wave and reflection(s)s thereof. The scattering elements 102 a, 102 b may include scattering elements that are embedded within, positioned on a surface of, or positioned within an evanescent proximity of, the wave-propagation structure 104. For example, the scattering elements can include complementary metamaterial elements such as those presented in D. R. Smith et al, “Metamaterials for surfaces and waveguides,” U.S. Patent Application Publication No. 2010/0156573, and A. Bily et al, “Surface scattering antennas,” U.S. Patent Application Publication No. 2012/0194399, each of which is herein incorporated by reference. As another example, the scattering elements can include patch elements such as those presented in A. Bily et al, “Surface scattering antenna improvements,” U.S. patent application Ser. No. 13/838,934, which is herein incorporated by reference.
The surface scattering antenna also includes at least one feed connector 106 that is configured to couple the wave-propagation structure 104 to a feed structure 108. The feed structure 108 (schematically depicted as a coaxial cable) may be a transmission line, a waveguide, or any other structure capable of providing an electromagnetic signal that may be launched, via the feed connector 106, into a guided wave or surface wave 105 of the wave-propagating structure 104. The feed connector 106 may be, for example, a coaxial-to-microstrip connector (e.g. an SMA-to-PCB adapter), a coaxial-to-waveguide connector, a mode-matched transition section, etc. While FIG. 1 depicts the feed connector in an “end-launch” configuration, whereby the guided wave or surface wave 105 may be launched from a peripheral region of the wave-propagating structure (e.g. from an end of a microstrip or from an edge of a parallel plate waveguide), in other embodiments the feed structure may be attached to a non-peripheral portion of the wave-propagating structure, whereby the guided wave or surface wave 105 may be launched from that non-peripheral portion of the wave-propagating structure (e.g. from a midpoint of a microstrip or through a hole drilled in a top or bottom plate of a parallel plate waveguide); and yet other embodiments may provide a plurality of feed connectors attached to the wave-propagating structure at a plurality of locations (peripheral and/or non-peripheral).
The scattering elements 102 a, 102 b are adjustable scattering elements having electromagnetic properties that are adjustable in response to one or more external inputs. Various embodiments of adjustable scattering elements are described, for example, in D. R. Smith et al, previously cited, and further in this disclosure. Adjustable scattering elements can include elements that are adjustable in response to voltage inputs (e.g. bias voltages for active elements (such as varactors, transistors, diodes) or for elements that incorporate tunable dielectric materials (such as ferroelectrics or liquid crystals)), current inputs (e.g. direct injection of charge carriers into active elements), optical inputs (e.g. illumination of a photoactive material), field inputs (e.g. magnetic fields for elements that include nonlinear magnetic materials), mechanical inputs (e.g. MEMS, actuators, hydraulics), etc. In the schematic example of FIG. 1, scattering elements that have been adjusted to a first state having first electromagnetic properties are depicted as the first elements 102 a, while scattering elements that have been adjusted to a second state having second electromagnetic properties are depicted as the second elements 102 b. The depiction of scattering elements having first and second states corresponding to first and second electromagnetic properties is not intended to be limiting: embodiments may provide scattering elements that are discretely adjustable to select from a discrete plurality of states corresponding to a discrete plurality of different electromagnetic properties, or continuously adjustable to select from a continuum of states corresponding to a continuum of different electromagnetic properties. Moreover, the particular pattern of adjustment that is depicted in FIG. 1 (i.e. the alternating arrangement of elements 102 a and 102 b) is only an exemplary configuration and is not intended to be limiting.
In the example of FIG. 1, the scattering elements 102 a, 102 b have first and second couplings to the guided wave or surface wave 105 that are functions of the first and second electromagnetic properties, respectively. For example, the first and second couplings may be first and second polarizabilities of the scattering elements at the frequency or frequency band of the guided wave or surface wave. In one approach the first coupling is a substantially nonzero coupling whereas the second coupling is a substantially zero coupling. In another approach both couplings are substantially nonzero but the first coupling is substantially greater than (or less than) than the second coupling. On account of the first and second couplings, the first and second scattering elements 102 a, 102 b are responsive to the guided wave or surface wave 105 to produce a plurality of scattered electromagnetic waves having amplitudes that are functions of (e.g. are proportional to) the respective first and second couplings. A superposition of the scattered electromagnetic waves comprises an electromagnetic wave that is depicted, in this example, as a plane wave 110 that radiates from the surface scattering antenna 100.
The emergence of the plane wave may be understood by regarding the particular pattern of adjustment of the scattering elements (e.g. an alternating arrangement of the first and second scattering elements in FIG. 1) as a pattern that defines a grating that scatters the guided wave or surface wave 105 to produce the plane wave 110. Because this pattern is adjustable, some embodiments of the surface scattering antenna may provide adjustable gratings or, more generally, holograms, where the pattern of adjustment of the scattering elements may be selected according to principles of holography. Suppose, for example, that the guided wave or surface wave may be represented by a complex scalar input wave Ψin that is a function of position along the wave-propagating structure 104, and it is desired that the surface scattering antenna produce an output wave that may be represented by another complex scalar wave Ψout. Then a pattern of adjustment of the scattering elements may be selected that corresponds to an interference pattern of the input and output waves along the wave-propagating structure. For example, the scattering elements may be adjusted to provide couplings to the guided wave or surface wave that are functions of (e.g. are proportional to, or step-functions of) an interference term given by Re[ΨoutΨin*]. In this way, embodiments of the surface scattering antenna may be adjusted to provide arbitrary antenna radiation patterns by identifying an output wave Ψout corresponding to a selected beam pattern, and then adjusting the scattering elements accordingly as above. Embodiments of the surface scattering antenna may therefore be adjusted to provide, for example, a selected beam direction (e.g. beam steering), a selected beam width or shape (e.g. a fan or pencil beam having a broad or narrow beamwidth), a selected arrangement of nulls (e.g. null steering), a selected arrangement of multiple beams, a selected polarization state (e.g. linear, circular, or elliptical polarization), a selected overall phase, or any combination thereof. Alternatively or additionally, embodiments of the surface scattering antenna may be adjusted to provide a selected near field radiation profile, e.g. to provide near-field focusing and/or near-field nulls.
Because the spatial resolution of the interference pattern is limited by the spatial resolution of the scattering elements, the scattering elements may be arranged along the wave-propagating structure with inter-element spacings that are much less than a free-space wavelength corresponding to an operating frequency of the device (for example, less than one-third, one-fourth, or one-fifth of this free-space wavelength). In some approaches, the operating frequency is a microwave frequency, selected from frequency bands such as L, S, C, X, Ku, K, Ka, Q, U, V, E, W, F, and D, corresponding to frequencies ranging from about 1 GHz to 170 GHz and free-space wavelengths ranging from millimeters to tens of centimeters. In other approaches, the operating frequency is an RF frequency, for example in the range of about 100 MHz to 1 GHz. In yet other approaches, the operating frequency is a millimeter-wave frequency, for example in the range of about 170 GHz to 300 GHz. These ranges of length scales admit the fabrication of scattering elements using conventional printed circuit board or lithographic technologies.
In some approaches, the surface scattering antenna includes a substantially one-dimensional wave-propagating structure 104 having a substantially one-dimensional arrangement of scattering elements, and the pattern of adjustment of this one-dimensional arrangement may provide, for example, a selected antenna radiation profile as a function of zenith angle (i.e. relative to a zenith direction that is parallel to the one-dimensional wave-propagating structure). In other approaches, the surface scattering antenna includes a substantially two-dimensional wave-propagating structure 104 having a substantially two-dimensional arrangement of scattering elements, and the pattern of adjustment of this two-dimensional arrangement may provide, for example, a selected antenna radiation profile as a function of both zenith and azimuth angles (i.e. relative to a zenith direction that is perpendicular to the two-dimensional wave-propagating structure). Exemplary adjustment patterns and beam patterns for a surface scattering antenna that includes a two-dimensional array of scattering elements distributed on a planar rectangular wave-propagating structure are depicted in FIGS. 2A-4B. In these exemplary embodiments, the planar rectangular wave-propagating structure includes a monopole antenna feed that is positioned at the geometric center of the structure. FIG. 2A presents an adjustment pattern that corresponds to a narrow beam having a selected zenith and azimuth as depicted by the beam pattern diagram of FIG. 2B. FIG. 3A presents an adjustment pattern that corresponds to a dual-beam far field pattern as depicted by the beam pattern diagram of FIG. 3B. FIG. 4A presents an adjustment pattern that provides near-field focusing as depicted by the field intensity map of FIG. 4B (which depicts the field intensity along a plane perpendicular to and bisecting the long dimension of the rectangular wave-propagating structure).
In some approaches, the wave-propagating structure is a modular wave-propagating structure and a plurality of modular wave-propagating structures may be assembled to compose a modular surface scattering antenna. For example, a plurality of substantially one-dimensional wave-propagating structures may be arranged, for example, in an interdigital fashion to produce an effective two-dimensional arrangement of scattering elements. The interdigital arrangement may comprise, for example, a series of adjacent linear structures (i.e. a set of parallel straight lines) or a series of adjacent curved structures (i.e. a set of successively offset curves such as sinusoids) that substantially fills a two-dimensional surface area. These interdigital arrangements may include a feed connector having a tree structure, e.g. a binary tree providing repeated forks that distribute energy from the feed structure 108 to the plurality of linear structures (or the reverse thereof). As another example, a plurality of substantially two-dimensional wave-propagating structures (each of which may itself comprise a series of one-dimensional structures, as above) may be assembled to produce a larger aperture having a larger number of scattering elements; and/or the plurality of substantially two-dimensional wave-propagating structures may be assembled as a three-dimensional structure (e.g. forming an A-frame structure, a pyramidal structure, or other multi-faceted structure). In these modular assemblies, each of the plurality of modular wave-propagating structures may have its own feed connector(s) 106, and/or the modular wave-propagating structures may be configured to couple a guided wave or surface wave of a first modular wave-propagating structure into a guided wave or surface wave of a second modular wave-propagating structure by virtue of a connection between the two structures.
In some applications of the modular approach, the number of modules to be assembled may be selected to achieve an aperture size providing a desired telecommunications data capacity and/or quality of service, and/or a three-dimensional arrangement of the modules may be selected to reduce potential scan loss. Thus, for example, the modular assembly could comprise several modules mounted at various locations/orientations flush to the surface of a vehicle such as an aircraft, spacecraft, watercraft, ground vehicle, etc. (the modules need not be contiguous). In these and other approaches, the wave-propagating structure may have a substantially non-linear or substantially non-planar shape whereby to conform to a particular geometry, therefore providing a conformal surface scattering antenna (conforming, for example, to the curved surface of a vehicle).
More generally, a surface scattering antenna is a reconfigurable antenna that may be reconfigured by selecting a pattern of adjustment of the scattering elements so that a corresponding scattering of the guided wave or surface wave produces a desired output wave. Suppose, for example, that the surface scattering antenna includes a plurality of scattering elements distributed at positions {rj} along a wave-propagating structure 104 as in FIG. 1 (or along multiple wave-propagating structures, for a modular embodiment) and having a respective plurality of adjustable couplings {αj} to the guided wave or surface wave 105. The guided wave or surface wave 105, as it propagates along or within the (one or more) wave-propagating structure(s), presents a wave amplitude Aj and phase φj to the jth scattering element; subsequently, an output wave is generated as a superposition of waves scattered from the plurality of scattering elements:
E ⁡ ( θ , ϕ ) = ∑ j ⁢ R j ⁡ ( θ , ϕ ) ⁢ α j ⁢ A j ⁢ ⅇ ⅈ ⁢ ⁢ φ j ⁢ ⅇ ⅈ ⁡ ( k ⁡ ( θ , ϕ ) · r j ) , ( 1 )
where E(θ, φ) represents the electric field component of the output wave on a far-field radiation sphere, Rj(θ, φ) represents a (normalized) electric field pattern for the scattered wave that is generated by the jth scattering element in response to an excitation caused by the coupling αj, and k(θ, φ) represents a wave vector of magnitude ω/c that is perpendicular to the radiation sphere at (θ, φ). Thus, embodiments of the surface scattering antenna may provide a reconfigurable antenna that is adjustable to produce a desired output wave E(θ, φ) by adjusting the plurality of couplings {αj} in accordance with equation (1).
The wave amplitude Aj and phase φj of the guided wave or surface wave are functions of the propagation characteristics of the wave-propagating structure 104. Thus, for example, the amplitude Aj may decay exponentially with distance along the wave-propagating structure, Aj˜A0 exp(−κxj), and the phase φj may advance linearly with distance along the wave-propagating structure, φj˜φ0+βxj, where κ is a decay constant for the wave-propagating structure, β is a propagation constant (wavenumber) for the wave-propagating structure, and xj is a distance of the jth scattering element along the wave-propagating structure. These propagation characteristics may include, for example, an effective refractive index and/or an effective wave impedance, and these effective electromagnetic properties may be at least partially determined by the arrangement and adjustment of the scattering elements along the wave-propagating structure. In other words, the wave-propagating structure, in combination with the adjustable scattering elements, may provide an adjustable effective medium for propagation of the guided wave or surface wave, e.g. as described in D. R. Smith et al, previously cited. Therefore, although the wave amplitude Aj and phase φj of the guided wave or surface wave may depend upon the adjustable scattering element couplings {αj} (i.e. Ai=Ai({αj}), φi=φi({αj})), in some embodiments these dependencies may be substantially predicted according to an effective medium description of the wave-propagating structure.
In some approaches, the reconfigurable antenna is adjustable to provide a desired polarization state of the output wave E(θ, φ). Suppose, for example, that first and second subsets LP(1) and LP(2) of the scattering elements provide (normalized) electric field patterns R(1)(θ, φ) and R(2)(θ, φ), respectively, that are substantially linearly polarized and substantially orthogonal (for example, the first and second subjects may be scattering elements that are perpendicularly oriented on a surface of the wave-propagating structure 104). Then the antenna output wave E(θ, φ) may be expressed as a sum of two linearly polarized components:
E ⁡ ( θ , ϕ ) = E ( 1 ) ⁡ ( θ , ϕ ) + E ( 2 ) ⁡ ( θ , ϕ ) = Λ ( 1 ) ⁢ R ( 1 ) ⁡ ( θ , ϕ ) + Λ ( 2 ) ⁢ R ( 2 ) ⁡ ( θ , ϕ ) , ⁢ where ( 2 ) Λ ( 1 , 2 ) ⁡ ( θ , ϕ ) = ∑ j ∈ LP ( 1 , 2 ) ⁢ α j ⁢ A j ⁢ ⅇ ⅈ ⁢ ⁢ φ j ⁢ ⅇ ⅈ ⁡ ( k ⁡ ( θ , ϕ ) · r j ) ( 3 )
are the complex amplitudes of the two linearly polarized components. Accordingly, the polarization of the output wave E(θ,φ) may be controlled by adjusting the plurality of couplings {αj} in accordance with equations (2)-(3), e.g. to provide an output wave with any desired polarization (e.g. linear, circular, or elliptical).
Alternatively or additionally, for embodiments in which the wave-propagating structure has a plurality of feeds (e.g. one feed for each “finger” of an interdigital arrangement of one-dimensional wave-propagating structures, as discussed above), a desired output wave E(θ, φ) may be controlled by adjusting gains of individual amplifiers for the plurality of feeds. Adjusting a gain for a particular feed line would correspond to multiplying the Aj's by a gain factor G for those elements j that are fed by the particular feed line. Especially, for approaches in which a first wave-propagating structure having a first feed (or a first set of such structures/feeds) is coupled to elements that are selected from LP(1) and a second wave-propagating structure having a second feed (or a second set of such structures/feeds) is coupled to elements that are selected from LP(2), depolarization loss (e.g., as a beam is scanned off-broadside) may be compensated by adjusting the relative gain(s) between the first feed(s) and the second feed(s).
Turning now to a consideration of modulation patterns for surface scattering antennas: recall, as discussed above, that the guided wave or surface wave may be represented by a complex scalar input wave Ψin that is a function of position along the wave-propagating structure. To produce an output wave that may be represented by another complex scalar wave Ψout, a pattern of adjustments of the scattering elements may be selected that corresponds to an interference pattern of the input and output waves along the wave-propagating structure. For example, the scattering elements may be adjusted to provide couplings to the guided wave or surface wave that are functions of a complex continuous hologram function h=ΨoutΨin*.
In some approaches, the scattering elements can be adjusted only to approximate the ideal complex continuous hologram function h=ΨoutΨin*. For example, because the scattering elements are positioned at discrete locations along the wave-propagating structure, the hologram function must be discretized. Furthermore, in some approaches, the set of possible couplings between a particular scattering elements and the waveguide is a restricted set of couplings; for example, an embodiment may provide only a finite set of possible couplings (e.g. a “binary” or “on-off” scenario in which there are only two available couplings for each scattering element, or a “grayscale” scenario in which there are N available couplings for each scattering element); and/or the relationship between the amplitude and phase of each coupling may be constrained (e.g. by a Lorentzian-type resonance response function). Thus, in some approaches, the ideal complex continuous hologram function is approximated by an actual modulation function defined on a discrete-valued domain (for the discrete positions of the scattering elements) and having a discrete-valued range (for the discrete available tunable settings of the scattering elements).
Consider, for example, a one-dimensional surface scattering antenna on which it is desired to impose an ideal hologram function defined as a simple sinusoid corresponding to a single wavevector (the following disclosure, relating to the one-dimensional sinusoid, is not intended to be limiting and the approaches set forth are applicable to other two-dimensional hologram patterns). Various discrete modulation functions may be used to approximate this ideal hologram function. In a “binary” scenario where only two values of individual scattering element coupling are available, one approach is to apply a Heaviside function to the sinusoid, creating a simple square wave. Regardless of the density of scattering elements, that Heaviside function will have approximately half the cells on and half off, in a steady repeating pattern. Unlike the spectrally pure sinusoid though, a square wave contains an (infinite) series of higher harmonics. In these approaches, the antenna may be designed so that the higher harmonics correspond to evanescent waves, making them non-radiating, but their aliases do still map into non-evanescent waves and radiate as grating lobes.
An illustrative example of the discretization and aliasing effect is shown in FIGS. 5A-5F. FIG. 5A depicts a continuous hologram function that is a simple sinusoid 500; in Fourier space, this is represented as a single Fourier mode 510 as shown in FIG. 5D. When the Heaviside function is applied to the sinusoid, the result is a square wave 502 as shown in FIG. 5B; in Fourier space, the square wave includes the fundamental Fourier mode 510 and an (infinite) series of higher harmonics 511, 512, 513, etc. as shown in FIG. 5E. Finally, when the square wave is sampled at a discrete set of locations corresponding to the discrete locations of the scattering elements, the result is a discrete-valued function 504 on a discrete domain, as shown in FIG. 5C (here assuming a lattice constant a).
The sampling of the square wave at a discrete set of locations leads to an aliasing effect in Fourier space, as shown in FIG. 5F. In this illustration, the sampling with a lattice constant a leads to a “folding” of the Fourier spectrum around the Nyquist spatial frequency π/a, creating aliases 522 and 523 for the original harmonics 512 and 513, respectively. Supposing that the aperture has an evanescent cutoff given by 2πf/c as shown (where f is an operating frequency of the antenna and c is the speed of light in an ambient medium surrounding the antenna, which can be vacuum, air, a dielectric material, etc.), one of the harmonics (513) is aliased into the non-evanescent spatial frequency range (523) and can radiate as a grating lobe. Note that in this example, the first harmonic 511 is unaliased but also within the non-evanescent spatial frequency range, so it can generate another undesirable side lobe
The Heaviside function is not the only choice for a binary hologram, and other choices may eliminate, average, or otherwise mitigate the higher harmonics and the resulting side/grating lobes. A useful way to view these approaches is as attempting to “smooth” or “blur” the sharp corners in the Heaviside without resorting to values other than 0 and 1. For example, the single step of the Heaviside function may be replaced by a function that resembles a pulse-width-modulated (PWM) square wave with a duty cycle that gradually increases from 0 to 1 over the range of the sinusoid. Alternatively, a probabilistic or dithering approach may be used to determine the settings of the individual scattering elements, for example by randomly adjusting each scattering element to the “on” or “off” state according to a probability that gradually increases from 0 to 1 over the range of the sinusoid.
In some approaches, the binary approximation of the hologram may be improved by increasing the density of scattering elements. An increased density results in a larger number of adjustable parameters that can be optimized, and a denser array results in better homogenization of electromagnetic parameters.
Alternatively or additionally, in some approaches the binary approximation of the hologram may be improved by arranging the elements in a non-uniform spatial pattern. If the scattering elements are placed on non-uniform grid, the rigid periodicity of the Heaviside modulation is broken, which spreads out the higher harmonics. The non-uniform spatial pattern can be a random distribution, e.g. with a selected standard deviation and mean, and/or it can be a gradient distribution, with a density of scattering elements that varies with position along the wave-propagating structure. For example, the density may be larger near the center of the aperture to realize an amplitude envelope.
Alternatively or additionally, in some approaches the binary approximation of the hologram may be improved by arranging the scattering elements to have non-uniform nearest neighbor couplings. Jittering these nearest-neighbor couplings can blur the k-harmonics, yielding reduced side/grating lobes. For example, in approaches that use a via fence to reduce coupling or crosstalk between adjacent unit cells, the geometry of the via fence (e.g. the spacing between vias, the sizes of the via holes, or the overall length of the fence) can be varied cell-by-cell. In other approaches that use a via fence to separate the cavities for a series of scattering elements that are cavity-fed slots, again the geometry of the via fence can be varied cell-by-cell. This variation can correspond to a random distribution, e.g. with a selected standard deviation and mean, and/or it can be a gradient distribution, with a nearest-neighbor coupling that varies with position along the wave-propagating structure. For example, the nearest-neighbor coupling may be largest (or smallest) near the center of the aperture.
Alternatively or additionally, in some approaches the binary approximation of the hologram may be improved by increasing the nearest-neighbor couplings between the scattering elements. For example, small parasitic elements can be introduced to act as “blurring pads” between the unit cells. The pad can be designed to have a smaller effect between two cells that are both “on” or both “off,” and a larger effect between an “on” cell and an “off” cell, e.g. by radiating with an average of the two adjacent cells to realize a mid-point modulation amplitude.
Alternatively or additionally, in some approaches the binary approximation of the hologram may be improved using error propagation or error diffusion techniques to determine the modulation pattern. An error propagation technique may involve considering the desired value of a pure sinusoid modulation and tracking a cumulative difference between that and the Heaviside (or other discretization function). The error accumulates, and when it reaches a threshold it carries over to the current cell. For a two-dimensional scattering antenna composed of a set of rows, the error propagation may be performed independently on each row; or the error propagation may be performed row-by-row by carrying over an error tally from the end of row to the beginning of the next row; or the error propagation may be performed multiple times along different directions (e.g. first along the rows and then perpendicular to the rows); or the error propagation may use a two-dimensional error propagation kernel as with Floyd-Steinberg or Jarvis-Judice-Ninke error diffusion. For an embodiment using a plurality of one-dimensional waveguides to compose a two-dimensional aperture, the rows for error diffusion can correspond to individual one-dimensional waveguides, or the rows for error diffusion can be oriented perpendicularly to the one-dimensional waveguides. In other approaches, the rows can be defined with respect to the waveguide mode, e.g. by defining the rows as a series of successive phase fronts of the waveguide mode (thus, a center-fed parallel plate waveguide would have “rows” that are concentric circles around the feed point). In yet other approaches, the rows can be selected depending on the hologram function that is being discretized—for example, the rows can be selected as a series of contours of the hologram function, so that the error diffusion proceeds along directions of small variation of the hologram function.
Alternatively or additionally, in some approaches grating lobes can be reduced by using scattering elements with increased directivity. Often the grating lobes appear far from the main beam; if the individual scattering elements are designed to have increased broadside directivity, large-angle aliased grating lobes may be significantly reduced in amplitude.
Alternatively or additionally, in some approaches grating lobes can be reduced by changing the input wave Ψin along the wave-propagating structure. By changing the input wave throughout a device, the spectral harmonics are varied, and large grating lobes may be avoided. For example, for a two-dimensional scattering antenna composed of a set of parallel one-dimensional rows, the input wave can be changed by alternating feeding directions for successive rows, or by alternating feeding directions for the top and bottom halves of the antenna. As another example, the effective index of propagation along the wave-propagating structure can be varied with position along the wave-propagating structure, by varying some aspect of the wave-propagating structure geometry (e.g. the positions of the vias in a substrate-integrated waveguide), by varying dielectric value (e.g. the filling fraction of a dielectric in a closed waveguide), by actively loading the wave-propagating structure, etc.
Alternatively or additionally, in some approaches the grating lobes can be reduced by introducing structure on top of the surface scattering antenna. For example, a fast-wave structure (such as a dispersive plasmonic or surface wave structure or an air-core-based waveguide structure) placed on top of the the surface-scattering antenna can be designed to propagate the evanescent grating lobe and carry it out to a load dump before it aliases into the non-evanescent region. As another example, a directivity-enhancing structure (such as an array of collimating GRIN lenses) can be placed on top of the surface scattering antenna to enhance the individual directivities of the scattering elements.
While some approaches, as discussed above, arrange the scattering elements in a non-uniform spatial pattern, other approaches maintain a uniform arrangement of the scattering elements but vary their “virtual” locations to be used in calculating the modulation pattern. Thus the scattering elements can physically still exist on a uniform grid (or any other fixed physical pattern), but their virtual location is shifted in the computation algorithm. For example, the virtual locations can be determined by applying a random displacement to the physical locations, the random displacement having a zero mean and controllable distribution, analogous to classical dithering. Alternatively, the virtual locations can be calculated by adding a non-random displacement from the physical locations, the displacement varying with position along the wave-propagating structure (e.g. with intentional gradients over various length scales).
In some approaches, undesirable grating lobes can be reduced by flipping individual bits corresponding to individual scattering elements. In these approaches, each element can be described as a single bit which contributes spectrally to both the desired fundamental modulation and to the higher harmonics that give rise to grating lobes. Thus, single bits that contribute to harmonics more than the fundamental can be flipped, reducing the total harmonics level while leaving the fundamental relatively unaffected.
Alternatively or additionally, undesirable grating lobes can be reduced by applying a spectrum (in k-space) of modulation fundamentals rather than a single fundamental, i.e. range of modulation wavevectors, to disperse energy put into higher harmonics. This is a form of modulation dithering. Because higher harmonics pick up an additional 2π wave-vector phase when they alias back into the visible, grating lobes resulting from different modulation wavevectors can be spread in radiative angle even while the main beams overlap. This spectrum of modulation wavevectors can be flat, Gaussian, or any other distribution across a modulation wavevector bandwidth.
Alternatively or additionally, undesirable grating lobes can be reduced by “chopping” the range-discretized hologram (e.g. after applying the Heaviside function but before sampling at the discrete set of scattering element locations) to selectively reduce or eliminate higher harmonics. Selective elimination of square wave harmonics is described, for example, in H. S. Patel and R. G. Hoft, “Generalized Techniques of Harmonic Elimination and Voltage Control in Thyristor Inverters: Part I—Harmonic Elimination,” IEEE Trans. Ind. App. Vol. IA-9, 310 (1973), herein incorporated by reference. For example, the square wave 502 of FIG. 5B can be modified with “chops” that eliminate the harmonics 511 and 513 (as shown in FIG. 5E) so that neither the harmonic 511 nor the aliased harmonic 531 (as shown in FIG. 5F) will generate grating lobes.
Alternatively or additionally, undesirable grating lobes may be reduced by adjusting the wavevector of the modulation pattern. Adjusting the wavevector of the modulation pattern shifts the primary beam, but shifts grating lobes coming from aliased beams to a greater degree (due to the additional 2π phase shift on every alias). Adjustment of the phase and wavevector of the applied modulation pattern can be used to intentionally form constructive and destructive interference of the grating lobes, side lobes, and main beam. Thus, allowing very minor changes in the angle and phase of the main radiated beam can grant a large parameter space in which to optimize/minimize grating lobes.
Alternatively or additionally, the antenna modulation pattern can be selected according to an optimization algorithm that optimizes a particular cost function. For example, the modulation pattern may be calculated to optimize: realized gain (maximum total intensity in the main beam); relative minimization of the highest side lobe or grating lobe relative to main beam; minimization of main-beam FWHM (beam width); or maximization of main-beam directivity (height above all integrated side lobes and grating lobes); or any combination thereof (e.g. by using a collective cost function that is a weighted sum of individual cost functions, or by selecting a Pareto optimum of individual cost functions). The optimization can be either global (searching the entire space of antenna configurations to optimize the cost function) or local (starting from an initial guess and applying an optimization algorithm to find a local extremum of the cost function).
Various optimization algorithms may be utilized to perform the optimization of the desired cost function. For example, the optimization may proceed using discrete optimization variables corresponding to the discrete adjustment states of the scattering elements, or the optimization may proceed using continuous optimization variables that can be mapped to the discrete adjustment states by a smoothed step function (e.g. a smoothed Heaviside function for a binary antenna or a smoothed sequential stair-step function for a grayscale antenna). Other optimization approaches can include optimization with a genetic optimization algorithm or a simulated annealing optimization algorithm.
The optimization algorithm can involve an iterative process that includes identifying a trial antenna configuration, calculating a gradient of the cost function for the antenna configuration, and then selecting a subsequent trial configuration, repeating the process until some termination condition is met. The gradient can be calculated by, for example, calculating finite-difference estimates of the partial derivatives of the cost function with respect to the individual optimization variables. For N scattering elements, this might involve performing N full-wave simulations, or performing N measurements of a test antenna in a test environment (e.g. an anechoic chamber). Alternatively, the gradient may be calculable by an adjoint sensitivity method that entails solving a single adjoint problem instead of N finite-difference problems; adjoint sensitivity models are available in conventional numerical software packages such as HFSS or CST Microwave Studio. Once the gradient is obtained, a subsequent trial configuration can be calculated using various optimization iteration approaches such as quasi-Newton methods or conjugate gradient methods. The iterative process may terminate, for example, when the norm of the cost function gradient becomes sufficiently small, or when the cost function reaches a satisfactory minimum (or maximum).
In some approaches, the optimization can be performed on a reduced set of modulation patterns. For example, for a binary (grayscale) antenna with N scattering elements, there are 2N (or gN, for g grayscale levels) possible modulation patterns, but the optimization may be constrained to consider only those modulation patterns that yield a desired primary spectral content in the output wave Ψout, and/or the optimization may be constrained to consider only those modulation patterns which have a spatial on-off fraction within a known range relevant for the design.
While the above discussion of modulation patterns has focused on binary embodiments of the surface scattering antenna, it will be appreciated that all of the various approaches described above are directly applicable to grayscale approaches where the individual scattering elements are adjustable between more than two configurations.
With reference now to FIG. 6, an illustrative embodiment is depicted as a system block diagram. The system includes a surface scattering antenna 600 coupled to control circuitry 610 operable to adjust the surface scattering to any particular antenna configuration. The system optionally includes a storage medium 620 on which is written a set of pre-calculated antenna configurations. For example, the storage medium may include a look-up table of antenna configurations indexed by some relevant operational parameter of the antenna, such as beam direction, each stored antenna configuration being previously calculated according to one or more of the approaches described above. Then, the control circuitry 610 would be operable to read an antenna configuration from the storage medium and adjust the antenna to the selected, previously-calculated antenna configuration. Alternatively, the control circuitry 610 may include circuitry operable to calculate an antenna configuration according to one or more of the approaches described above, and then to adjust the antenna for the presently-calculated antenna configuration.
discretizing a hologram function for a surface scattering antenna; and
identifying an antenna configuration that reduces artifacts attributable to the discretizing.
adjusting the surface scattering antenna to the identified antenna configuration.
operating the surface scattering antenna in the identified antenna configuration.
storing the identified antenna configuration in a storage medium.
5. The method of claim 1, wherein the surface scattering antenna defines an aperture and the discretizing includes identifying a discrete plurality of locations on the aperture for a discrete plurality of scattering elements of the surface scattering antenna.
6. The method of claim 5, wherein the discretizing includes identifying a discrete set of states for each of the scattering elements corresponding to a discrete set of function values at each of the locations of the scattering elements.
7. The method of claim 6, wherein the identifying of the antenna configuration includes:
selecting, for the plurality of locations, a plurality of function values from the discrete set of function values, where the selected plurality optimizes a desired cost function for an antenna pattern of the antenna.
8. The method of claim 7, wherein the selecting that optimizes the desired cost function is a selecting with a discrete optimization algorithm.
9. The method of claim 8, wherein the discrete set of function values is a binary set of function values.
10. The method of claim 8, wherein the discrete set of function values is a grayscale set of function values.
11. The method of claim 7, wherein the selecting that optimizes the desired cost function is a selecting with a continuous optimization algorithm.
12. The method of claim 7, wherein the selecting that optimizes the desired cost function includes evaluating the desired cost function for a sequence of trials, each trial consisting of a plurality of trial function values for the plurality of locations, where each of the trial function values selected from the discrete set of function values.
13. The method of claim 12, wherein the evaluating of the desired cost function for the sequence of trials includes, for each trial in the sequence of trials:
identifying a trial antenna configuration corresponding to the plurality of trial function values;
performing a full-wave simulation of the trial antenna configuration; and
evaluating the desired cost function with results of the full-wave simulation.
14. The method of claim 12, wherein the evaluating of the desired cost function for the sequence of trials includes, for each trial in the sequence of trials:
measuring a test antenna in the trial antenna configuration; and
evaluating the desired cost function with data from the measuring.
15. The method of claim 7, wherein the cost function maximizes a gain of the antenna in a selected direction, maximizes a directivity of the antenna in a selected direction, minimizes a half-power beamwidth of a main beam of the antenna pattern, minimizes a height of a highest side lobe relative to a main beam of the antenna pattern, or minimizes a height of a highest grating lobe relative to a main beam of the antenna pattern.
16. A method of controlling an surface scattering antenna with a plurality of adjustable scattering elements, comprising:
reading an antenna configuration from a storage medium, the antenna configuration being selected to reduce artifacts attributable to a discretization of a hologram function; and
adjusting the plurality of adjustable scattering elements to provide the antenna configuration.
operating the antenna in the antenna configuration.
18. The method of claim 16, wherein the adjustable scattering elements are adjustable between a discrete set of states corresponding to a discrete set of function values at each location in a plurality of locations for the plurality of adjustable scattering elements.
19. The method of claim 18, wherein the antenna configuration is selected to optimize, in a space of antenna configurations, a desired cost function for the antenna configuration.
20. The method of claim 19, wherein the antenna configuration is selected with a discrete optimization algorithm.
21. The method of claim 20, wherein the discrete set of function values is a binary set of function values.
22. The method of claim 20, wherein the discrete set of function values is a grayscale set of function values.
23. The method of claim 19, wherein the antenna configuration is selected with a continuous optimization algorithm.
24. The method of claim 19, wherein the antenna configuration is selected with an optimization algorithm that includes:
evaluating the desired cost function for a sequence of trial antenna configurations.
25. The method of claim 24, wherein the evaluating of the desired cost function for the sequence of trials includes, for each trial antenna configuration in the sequence of trial antenna configurations:
26. The method of claim 24, wherein the evaluating of the desired cost function for the sequence of trials includes, for each trial antenna configuration in the sequence of trial antenna configurations:
27. The method of claim 19, wherein the cost function maximizes a gain of the antenna in a selected direction, maximizes a directivity of the antenna in a selected direction, minimizes a half-power beamwidth of a main beam of the antenna pattern, minimizes a height of a highest side lobe relative to a main beam of the antenna pattern, or minimizes a height of a highest grating lobe relative to a main beam of the antenna pattern.
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