Source: http://www.google.com/patents/US20020093388?dq=6760745
Timestamp: 2016-05-25 01:59:59
Document Index: 160060006

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Patent US20020093388 - Voltage-controlled oscillator and communication device - Google PatentsSearch Images Maps Play YouTube News Gmail Drive More »Sign inPatentsAn impedance transformer is so designed that the impedance and the reactance of a tuning part reach zero at a central control voltage. An open stub is employed for setting the characteristic impedance of a tuning stub to the minimum value when a variable reactance circuit is inductive while setting the...http://www.google.com/patents/US20020093388?utm_source=gb-gplus-sharePatent US20020093388 - Voltage-controlled oscillator and communication deviceAdvanced Patent SearchPublication numberUS20020093388 A1Publication typeApplicationApplication numberUS 09/985,940Publication dateJul 18, 2002Filing dateNov 6, 2001Priority dateNov 6, 2000Also published asUS6628174Publication number09985940, 985940, US 2002/0093388 A1, US 2002/093388 A1, US 20020093388 A1, US 20020093388A1, US 2002093388 A1, US 2002093388A1, US-A1-20020093388, US-A1-2002093388, US2002/0093388A1, US2002/093388A1, US20020093388 A1, US20020093388A1, US2002093388 A1, US2002093388A1InventorsToshikazu Imaoka, Katsuaki OnodaOriginal AssigneeToshikazu Imaoka, Katsuaki OnodaExport CitationBiBTeX, EndNote, RefManReferenced by (7), Classifications (4), Legal Events (4) External Links: USPTO, USPTO Assignment, EspacenetVoltage-controlled oscillator and communication device
DESCRIPTION OF THE PREFERRED EMBODIMENTS [0058] [0058]FIG. 1 is a block diagram showing the structure of a voltage-controlled oscillator according to an embodiment of the present invention. [0059] The voltage-controlled oscillator shown in FIG. 1 is formed by an oscillation part 100, a tuning part 200, an output circuit 300 and a terminal circuit 400. [0060] The oscillation part 100, including an amplification circuit 110 and a feedback circuit 120, performs oscillation. The amplification circuit 110 is formed by an active element such as a transistor and a bias application circuit for the active element. The bias application circuit is band-limited to prevent passage of a prescribed frequency (oscillation frequency). The feedback circuit 120, forming a feedback loop along with the amplification circuit 110, returns an output signal from the amplification circuit 110 to an input side in the same phase and grows oscillation. [0061] The tuning part 200 includes an impedance transformer 210 and a variable reactance circuit 220. The impedance transformer 210 includes a tuning stub formed by a transmission line (hereinafter referred to as an open stub) having an opened forward end or a transmission line (hereinafter referred to as a short stub) having a shorted forward end. The variable reactance circuit 220 is formed by a variable capacitance element such as a varactor diode and an inductor defined by a bonding wire or a wire. The oscillation frequency can be tuned by changing the reactance of the tuning part 200 and changing the phase quantity of the feedback loop. [0062] The output circuit 300 is formed by a dc component removing circuit defined by a capacitor for removing a dc component, an attenuator for reducing load fluctuation and the like. The terminal circuit 400 consumes power of frequencies other than the oscillation frequency as heat, for stabilizing oscillation. [0063] This voltage-controlled oscillator varies the oscillation frequency with a control voltage applied to the variable capacitance element of the variable reactance circuit 220. [0064] In the voltage-controlled oscillator according to this embodiment, tuning sensitivity can be maximized by optimizing the impedance transformer 210 by a method described later. The term “tuning sensitivity” stands for the rate of change (width of change) of the oscillation frequency with respect to change of the control voltage. [0065] [0065]FIG. 2 is a circuit diagram showing an exemplary structure of the voltage-controlled oscillator shown in FIG. 1 in detail. FIG. 3 is a plan view of the voltage-controlled oscillator shown in FIG. 2. The voltage-controlled oscillator shown in FIGS. 2 and 3 is an internal feedback and self-biased voltage-controlled oscillator. [0066] Referring to FIG. 3, a metal-semiconductor field-effect transistor (MESFET: hereinafter referred to as a transistor) 4 of GaAs is formed on a dielectric substrate 1. A grounding conductor is formed on the rear surface of the dielectric substrate 1. A gate-side feedback microstrip line 5, a drain-side feedback microstrip line 6 and an output microstrip line 7 are formed on the dielectric substrate 1. A gate electrode G, a drain electrode D and a source electrode S of the transistor 4 are connected to first ends of the gate-side feedback microstrip line 5, the drain-side feedback microstrip line 6 and the output microstrip line 7 respectively. [0067] A sectorial tuning stub (radial stub) 2 is formed on the dielectric substrate 1. Low characteristic impedance can be readily implemented by employing the sectorial tuning stub 2. A second end of the microstrip line 5 is connected to an anode A of a varactor diode 3. A cathode C of the varactor diode 3 is connected to a first end of the tuning stub 2. In the example shown in FIG. 3, a second end of the tuning stub 2 is opened. [0068] The cathode C of the varactor diode 3 is connected to a pad electrode 12 receiving a positive control voltage VC(+) through a control bias resistor 11. The control bias resistor 11 has a large resistance value of 10 kΩ, for example. Therefore, the control voltage VC(+) can be applied to the cathode C of the varactor diode 3 without leaking any signal. [0069] The gate electrode G of the transistor 4 is connected to a pad electrode 20 receiving a ground potential GND through a band rejection filter 13 rejecting passage of a high frequency of the oscillation band and a terminal resistor 14. [0070] A second end of the microstrip line 6 is opened. The drain electrode D of the transistor 4 is connected to a pad electrode 15 through a band rejection filter 16 rejecting passage of the high frequency of the oscillation band. The pad electrode 15 is grounded through a capacitor 17 (not shown in FIG. 3). A drain bias Vdd is applied to the pad electrode 15. [0071] The source electrode S of the transistor 4 is connected to a pad electrode 21 receiving the ground potential GND through a band rejection filter 18 rejecting passage of the high frequency of the oscillation band and a self-bias resistor 19. [0072] As shown in FIG. 2, a second end of the microstrip line 7 is connected to an output node 10 through an attenuator 8 and a dc component removing circuit 9. The dc component removing circuit 9 is formed by a capacitor. FIG. 3 illustrates neither the attenuator 8 nor the dc component removing circuit 9. [0073] In the voltage-controlled oscillator shown in FIGS. 2 and 3, the transistor 4 forms the amplification circuit 110 shown in FIG. 1, and the microstrip lines 5, 6 and 7 form the feedback circuit 120. The tuning stub 2 forms the impedance transformer 210, and the varactor diode 3 forms the variable reactance circuit 220. The attenuator 8 and the dc component removing circuit 9 form the output circuit 300. The band rejection filter 16 and the capacitor 17 form a drain bias application circuit, while the band rejection filter 18 and the self-bias resistor 19 form a source bias application circuit. The band rejection filter 13 and the terminal resistor 14 form the terminal circuit 400. [0074] Operations of the voltage-controlled oscillator shown in FIGS. 2 and 3 are now described. The transistor 4 amplifies a small microwave signal generated from the gate electrode G and outputs the same to the drain electrode D. The microwave signal having a prescribed frequency is totally reflected by the open end of the microstrip line 6, and fed back to the gate electrode G through capacities between the drain electrode D, the source electrode S and the gate electrode G. The microwave signal fed back to the gate electrode G is reflected by the node between the microstrip line 5 and the varactor diode 3 with phase change responsive to the impedance of the node. Thus, a feedback loop is formed at the prescribed frequency, and an output signal OUT oscillating at the prescribed frequency is obtained from the source electrode S of the transistor 4. When the control voltage VC(+) is varied, the capacitance value of the varactor diode 3 changes. Thus, the oscillation frequency changes. [0075] [0075]FIG. 4 is a circuit diagram noting the oscillation frequency of the voltage-controlled oscillator shown in FIG. 2. In the following description, it is assumed that fn denotes the oscillation frequency. [0076] Referring to FIG. 4, oscillation is performed under such a condition that the electric length (around phase) of the overall circuit of the voltage-controlled oscillator at the oscillation frequency fn, i.e., the electric length between the open ends of the tuning stub 2 and the microstrip line 6 is integral times 2π and the gain exceeds 1. [0077] The characteristic impedance Zst and the length Lst of the tubing stub 2 are so set that the impedance Zmod of the tuning part 200 as viewed from the node P between the oscillation part 100 and the tuning part 200 reaches zero at the oscillation frequency fn, i.e., the node P is shorted at the oscillation frequency fn. Thus, the electric length of the tuning part 200 at the oscillation frequency fn is (π/2) rad. The electric length of the oscillation part 100 at the oscillation frequency fn is (2πn−π/2) rad, where n denotes a positive integer. [0078] [0078]FIG. 5 is a Smith chart showing control voltage dependency of the impedance of the tuning part 200 shown in FIG. 4 at the oscillation frequency fn. [0079] Referring to FIG. 5, it is assumed that Xmod denotes the reactance of the tuning part 200 at the oscillation frequency fn as viewed from the node P between the oscillation part 100 and the tuning part 200. Assuming that the resistance component of the tuning part 200 is zero, the impedance Zmod of the tuning part 200 at the oscillation frequency fn is equal to the reactance Xmod at the oscillation frequency fn. [0080] It is assumed that VC denotes the central value (hereinafter referred to as a central control voltage) of the control voltage VC(+) applied to the cathode C of the varactor diode 3 with an upper limit VC+ and a lower limit VC−. In this example, the central control voltage VC is +3 V, the upper limit VC+ of the control voltage VC(+) is +5 V, and the lower limit VC− of the control voltage VC(+) is +1 V. [0081] Referring to FIG. 5, the right point R denotes the reactance at the open end of the tuning stub 2. The right points P0, P1 and P2 denote the reactance values at the central control voltage VC, the upper limit VC+ and the lower limit VC− of the control voltage VC(+) respectively. [0082] When the control voltage VC(+) changes from the lower limit VC− to the upper limit VC+ and the locus of the impedance Zmod includes the point P0 of reactance zero (short) and the reactance substantially reaches zero at the central control voltage VC, change (θ) of the phase with respect to change of the reactance Xmod is increased, as shown in FIG. 5. Thus, the rate of change of the oscillation frequency fn with respect to change of the control voltage VC(+) can be increased. Therefore, the parameters of the tuning stub 2 are so set that the node P between the oscillation part 100 and the tuning part 200 is shorted when the control voltage VC(+) is at the central control voltage VC. [0083] Methods of optimizing the impedance transformer 210 are now described. [0084] (1) First Optimization Method [0085] The impedance transformer 210 of the tuning part 200 can be formed by employing an open stub (line having an open forward end) or a short stub (line having a shorted forward end) as the tuning stub 2. [0086] [0086]FIG. 6 illustrates a tuning part 200 employing an open stub, and FIG. 7 illustrates a tuning part 200 employing a short stub. [0087] Referring to each of FIGS. 6 and 7, the varactor diode 3 is equivalently simplified to a variable capacitor and a fixed inductor. Symbol L denotes an inductive (inductor) component such as a bonding wire or a pad, and symbol C denotes a capacitive (capacitor) component such as a capacitor or a parasitic capacitor of the varactor diode 3. [0088] The impedance transformer 210 is so designed that the impedance Zmod and the reactance Xmod of the tuning part 200 are equal to j and 0 respectively when the control voltage VC(+) is at the central control voltage VC. [0089] At this time, four cases are assumed in response to the structure (open or short stub) of the impedance transformer 210 and the characteristic (capacitive or inductive) of the variable reactance circuit 220, for obtaining optimization conditions for maximizing tuning sensitivity in the respective cases. [0090] It is assumed that A denotes the inductive component of the variable reactance circuit 220, and B denotes the capacitive component of the variable reactance circuit 220. The value of the inductive component A is regularly constant while the value of the capacitive component B, which is variable, is assumed to be that at the central control voltage VC. It is also assumed that α (=A/B) denotes the ratio of the inductive component A to the capacitive component B of the variable reactance circuit 220. The reactance XVD of only the variable reactance circuit 220 at the central control voltage VC is (A−B). [0091] It is further assumed that Zstub denotes the characteristic impedance of the tuning stub 2 as viewed from the node Q between the varactor diode 3 and the tuning stub 2. [0092] In the following description, ωx denotes an oscillation angular frequency at an arbitrary control voltage Vx, ωC denotes an oscillation angular frequency (center oscillation angular frequency) at the central control voltage VC, and ωn denotes an oscillation angular frequency normalized with the oscillation angular frequency ωC, where ωn=ωx/ωC. [0093] Cx denotes the capacity of the varactor diode 3 at the control voltage Vx, and Cn denotes the capacity of the varactor diode 3 normalized with the capacity CC, where Cn=Cx/CC. At the central control voltage VC, ωn=1 and Cn=1. [0094] Zst denotes the characteristic impedance [Ω] of the tuning stub 2, and θst denotes the electric length [rad] of the tuning stub 2 at the center oscillation angular frequency ωC. [0095] (Open Stub) [0096] In the case of the open stub, the characteristic impedance Zst is expressed as follows: Z st =B(α−1)tan θst=(A−B)tan θst (A1) [0097] It is assumed that Cn(ωn) denotes the capacity of the varactor diode 3 at the oscillation angular frequency ωn. At the central control voltage VC (ωn=1), the rate Cn′(ωn) of change of the capacity Cn(ωn) of the varactor diode 3 is expressed as follows: C n′(1)=(1−α)�{2θst/sin(2θst)}−(1+α) (A2) [0098] The capacity Cn(ωn) indicates that the capacity Cn is a function of the oscillation angular frequency ωn. The rate Cn′(ωn) of change, obtained by differentiating the capacity Cn(ωn) by the oscillation angular frequency ωn, expresses the rate of change of the capacity of the varactor diode 3 with respect to change of the oscillation angular frequency ωn. A method of deriving the above equation (A2) is described later. [0099] [0099]FIG. 8 shows the relation between the electric length θst of the tuning stub 2, the characteristic impedance Zst of the tuning stub 2 and the rate Cn′(1) of change of the capacity of the varactor diode 3 with reference to the tuning stub 2 formed by an open stub at the center oscillation angular frequency ωC. The rate Cn′(1) of change is negative since the capacity Cn is reduced as the oscillation angular frequency is increased. [0100] It is assumed that the ratio α is equal to 1.2 when the variable reactance circuit 220 is inductive (α>1), and the ratio α is equal to 0.8 when the variable reactance circuit 220 is capacitive (α<1). [0101] It is understood from FIG. 8 that the rate Cn′(1) of change of the capacity of the varactor diode 3 is reduced as the characteristic impedance Zst of the tuning stub 2 is reduced when the variable reactance circuit 220 is inductive (α>1). Therefore, the characteristic impedance Zst of the tuning stub 2 is preferably set to the minimum value. [0102] If the implementable characteristic impedance Zst of the tuning stub 2 is limited due to a reason in fabrication or the like, however, the characteristic impedance Zst of the tuning stub 2 is minimized within the limit. [0103] If the variable reactance circuit 220 is capacitive (α<1), the rate Cn′(1) of change of the capacity of the varactor diode 3 has the minimum value when the electric length θst is substantially at (23π/32) [rad]. Therefore, the characteristic impedance Zst of the tuning stub 2 is preferably substantially −1.22 times the reactance XVD (=A−B), where −1.22 is the value of tan θst when the electric length θst is equal to (23π/32) [rad]. [0104] If the implementable characteristic impedance Zst of the tuning stub 2 is limited, however, the characteristic impedance Zst of the tuning stub 2 is set as close as possible to −1.22 times the reactance XVD (=A−B) within the limit. [0105] The electric length θst of the tuning stub 2 may be slightly displaced from (23π/32) [rad] when the rate Cn′(1) of change of the capacity of the varactor diode 3 has the minimum value. In this case, the characteristic impedance Zst of the tuning stub 2 may be set to a value slightly displaced from −1.22 times the reactance XVD. [0106] The characteristic impedance Zst of the tuning stub 2 may be set to tan(23π/32) times the reactance XVD of the variable reactance circuit 220 at the central control voltage VC, or within the range between tan{(23π/32)−(π/10)} times and tan{(23π/32)+(π/10)} times the reactance XVD of the variable reactance circuit 220. The electric length θst of the tuning stub 2 may be set to (23π/32) [rad], or within the range between tan{(23π/32)−(π/10)} [rad] and tan{(23π/32)+(π/10)} [rad]. [0107] (Short Stub) [0108] In the case of a short stub, the characteristic impedance Zst is expressed as follows: Z st =B(1−α)cot θst=−(A−B)cot θst (B1) [0109] It is assumed that Cn(ωn) denotes the capacity of the varactor diode 3 at the oscillation angular frequency ωn. At the central control voltage VC (ωn=1), the rate Cn′(ωn) of change of the capacity Cn(ωn)) of the varactor diode 3 is expressed as follows: C n′(1)=(α−1)�{2θst/sin(2θst)}−(1+α) (B2) [0110] The capacity Cn(ωn) indicates that the capacity Cn is a function of the oscillation angular frequency ωn. The rate Cn′(ωn) of change, obtained by differentiating the capacity Cn(ωn) by the oscillation angular frequency ωn, expresses the rate of change of the capacity of the varactor diode 3 with respect to change of the oscillation angular frequency ωn. A method of deriving the above equation (B2) is described later. [0111] [0111]FIG. 9 shows the relation between the electric length θst of the tuning stub 2, the characteristic impedance Zst of the tuning stub 2 and the rate Cn′(1) of change of the varactor 4 diode 3 with reference to the tuning stub 2 formed by a short stub. The rate Cn′(1) of change of the capacity is negative since the capacity Cn is reduced as the oscillation angular frequency ωn is increased. [0112] It is assumed that the ratio α is equal to 1.2 when the variable reactance circuit 220 is inductive (α>1), and the ratio α is equal to 0.8 when the variable reactance circuit 220 is capacitive (α<1). [0113] It is understood from FIG. 9 that the rate Cn′(1) of change of the capacity of the varactor diode 3 has the minimum value when the electric length θst is substantially (23π/32) [rad] if the variable reactance circuit 220 is inductive (α>1). Therefore, the characteristic impedance Zst of the tuning stub 2 is preferably substantially 0.82 times the reactance XVD (=A−B), where 0.82 is the value of cot θst when the electric length θst is equal to (23π/32) [rad]. [0114] If the implementable characteristic impedance Zst of the tuning stub 2 is limited due to a reason in fabrication or the like, however, the characteristic impedance Zst of the tuning stub 2 is set as close as possible to 0.82 times the reactance XVD (=A−B) within the limit. [0115] The electric length θst of the tuning stub 2 maybe slightly displaced from (23π/32) [rad] when the rate Cn′(1) of change of the capacity of the varactor diode 3 has the minimum value. In this case, the characteristic impedance Zst of the tuning stub 2 may be set to a value slightly displaced from 0.82 times the reactance XVD. [0116] The characteristic impedance Zst of the tuning stub 2 may be set to cot (23π/32) times the reactance XVD of the variable reactance circuit 220 at the central control voltage VC, or within the range between cot{(23π/32)−(π/10)} times and cot{(23π/32)+(π/10)} times the reactance XVD of the variable reactance circuit 220. The electric length θst of the tuning stub 2 may be set to (23π/32) [rad], or within the range between cot{(23π/32)−(π/10)} [rad] and cot{(23π/32)+(π/10)} [rad]. [0117] If the variable reactance circuit 220 is capacitive (α<1), the rate Cn′(1) of change of the capacity of the varactor diode 3 is reduced as the characteristic impedance Zst of the tuning stub 2 is increased. Therefore, the characteristic impedance Zst of the tuning stub 2 is preferably maximized. [0118] If the implementable characteristic impedance Zst of the tuning stub 2 is limited, however, the characteristic impedance Zst of the tuning stub 2 is set to the maximum within the limit. [0119] [0119]FIG. 10 shows optimum values of the characteristic impedance Zst of the tuning stub 2 and the electric length θst of the tuning stub 2 as to six cases (a) to (f). [0120] The central oscillation frequency fn is 10 GHz. The inductive component A is equal to ω�L=2πfnL, and the capacitive component B is equal to 1/(ω�C)=1/2πfnC. [0121] In the cases (a) to (d), the characteristic impedance Zst is limited to 20 to 60 Ω. In the cases (e) and (f), the characteristic impedance Zst is not limited. The tuning stub 2 is formed by an open stub in the cases (a), (b) and (e), and formed by a short stub in the cases (c), (d) and (f). [0122] It is assumed that the capacitive component B is 50 [Ω] in all cases (a) to (f). The inductive component A is 60 [Ω] and the variable reactance circuit 220 is inductive (α>1) in the cases (a), (d) and (f). The inductive component A is 40 [Ω] and the variable reactance circuit 220 is capacitive (α<1) in the cases (b), (c) and (e). [0123] In the case (a), the optimum value of the characteristic impedance Zst reaches the lower limit of 20 Ω when the electric length θst is 1.107 rad. In the case (b), the optimum value of the characteristic impedance Zst reaches the lower limit of 20 Ω when the electric length θst is 2.034 rad. In the case (c), the optimum value of the characteristic impedance Zst reaches the upper limit of 60 Ω when the electric length θst is 0.165 rad. In the case (d), the optimum value of the characteristic impedance Zst reaches the lower limit of 20 Ω when the electric length θst is 2.678 rad. [0124] In the case (e), the optimum value of the characteristic impedance Zst reaches 12.2 Ω when the electric length θst is (23π/32) rad. In the case (f), the optimum value of the characteristic impedance Zst reaches 8.2 Ω when the electric length θst is (23π/32) rad. [0125] (2) Second Optimization Method [0126] (Open Stub) [0127] The reactance XVD (=A−B) of only the variable reactance circuit 220 at the central control voltage VC is set substantially to zero. Thus, the rate Cn′(1) of change of the capacity of the varactor diode 3 is minimized to −2 regardless of the characteristic impedance Zst of the tuning stub 2 when the tuning stub 2 is formed by an open stub. In this case, the tuning part 200 serially resonates at the center oscillation angular frequency ωC, and tuning sensitivity is maximized. The characteristic impedance Zst of the tuning stub 2 is arbitrary, and the electric length θst of the tuning stub 2 is substantially (π/2) rad. [0128] In order to set the reactance XVD (=A−B) of only the variable reactance circuit 220 at the central control voltage VC substantially to zero, a fixed capacitor is added to the varactor diode 3 in a serial or parallel manner, or an inductor formed by a bonding wire or a wire is adjusted. [0129] The electric length θst of the tuning stub 2 may be set to (π/2) rad, or set within the range between {(π/2)−(π/10)} rad and {(π/2)+(π/10)} rad. [0130] (Short Stub) [0131] The reactance XVD (=A−B) of only the variable reactance circuit 220 at the central control voltage VC is set substantially to zero. Thus, the rate Cn′(1) of change of the capacity of the varactor diode 3 is minimized to −2 regardless of the characteristic impedance Zst of the tuning stub 2 when the tuning stub 2 is formed by a short stub. In this case, the tuning part 200 serially resonates at the center oscillation angular frequency ωC, and tuning sensitivity is maximized. The characteristic impedance Zst of the tuning stub 2 is arbitrary, and the electric length θst of the tuning stub 2 is substantially 0 rad. [0132] In order to set the reactance XVD (=A−B) of only the variable reactance circuit 220 at the central control voltage VC substantially to zero, a fixed capacitor is added to the varactor diode 3 in a serial or parallel manner, or an inductor formed by a bonding wire or a wire is adjusted. [0133] The electric length θst of the tuning stub 2 may be set to 0 rad, or set within the range between 0 rad and (π/10) rad. [0134] [0134]FIG. 11 is a plan view of a voltage-controlled oscillator employing the second optimization method. In the voltage-controlled oscillator shown in FIG. 11, a fixed capacitor 31 is connected between a cathode C of a varactor diode 3 and a tuning stub 2, and another fixed capacitor 32 is connected in parallel with the varactor diode 3. [0135] Thus, the reactance XVD (=A−B) of only a variable reactance circuit 220 at the central control voltage VC is set substantially to zero. The electric length θst of the tuning stub 2 is set substantially to (π/2) rad. [0136] The remaining parts of the voltage-controlled oscillator shown in FIG. 11 are similar in structure to those of the voltage-controlled oscillator shown in FIG. 3. [0137] The relation between the oscillation angular frequency ωn and the capacity Cn of the varactor diode 3 was investigated in voltage-controlled oscillators according to Inventive Examples 1 and 2 and comparative example. [0138] The voltage-controlled oscillators according to Inventive Examples 1 and 2 have the structures shown in FIGS. 3 and 11 respectively. [0139] [0139]FIG. 12 is a plan view showing the voltage-controlled oscillator according to comparative example. The voltage-controlled oscillator shown in FIG. 12 is provided with a linear tuning stub 2 a in place of the sectorial tuning stub 2. The remaining parts of the voltage-controlled oscillator shown in FIG. 12 are similar in structure to those of the voltage-controlled oscillator shown in FIG. 3. [0140] The characteristic impedance Zst of the tuning stub 2 of the variable reactance circuit 220 was set to 20 Ω in Inventive Example 1, while the characteristic impedance Zst of the tuning stub 2 a was set to 50 Ω in comparative example. In comparative example, the electric length θst of the tuning stub 2 a was 1.373 rad (78.7 deg). [0141] In Inventive Example 2, the ratio α (A/B) was set to 1 by either one of the following two methods, and the electric length θst of the tuning stub 2 was set to (π/2) rad. The characteristic impedance Zst of the tuning stub 2 was arbitrary. In this case, the capacity Cn was equal to 1/ωn 2, and the rate of change of the oscillation angular frequency ωn was maximized. [0142] For example, the inductive component A is set to 60 [Ω] and the fixed capacitor 31 of 1.59 pH is serially added to the varactor diode 3 so that the total of the capacitance values of the varactor diode 3 and the fixed capacitor 31 is 0.265 pH, thereby setting the capacitive component B to 60 [Ω]. Thus, the inductive and capacitive components A and B are equally set to 60 [Ω]. Alternatively, the capacitive component B is set to 50 [Ω] and the inductive component L is set to 0.796 nH by reducing the length of the bonding wire of the variable reactance circuit 220, thereby setting the inductive component A to 50 [Ω]. Thus, the inductive and capacitive components A and B are equally set to 50 [Ω]. [0143] [0143]FIGS. 13 and 14 illustrate the relation between the oscillation angular frequency ωn and the capacity Cn of the varactor diode 3 in Inventive examples 1 and 2 and comparative example. The variable reactance circuit 220 according to Inventive example is inductive in FIG. 13 and capacitive in FIG. 14. [0144] Referring to FIG. 13, Inventive Example 1 having the inductive variable reactance circuit 220 corresponds to the case (a) shown in FIG. 10 with the inductive and capacitive components A and B of 60 [Ω] and 50 [Ω] respectively and the reactance XVD (ωn=1 and Cn=1) of the variable reactance circuit 220 equal to A−B>0 at the central control voltage VC. [0145] Referring to FIG. 14, Inventive Example 1 having the inductive variable reactance circuit 220 corresponds to the case (b) shown in FIG. 10 with the inductive and capacitive components A and B of 40 [Ω] and 50 [Ω] respectively and the reactance XVD (ωn=1 and Cn=1) of the variable reactance circuit 220 equal to A−B<0 at the central control voltage VC. [0146] It is understood from FIGS. 13 and 14 that the rate of change of the capacity Cn necessary for tuning the oscillation frequency in the range of 9.0 to 11.0 GHz with reference to 10 GHz is 0.78 to 1.35 in Inventive Example 1 and 0.74 to 1.52 in comparative example. In other words, Inventive Example 1 attains the same tuning range as comparative example with small change of the capacity Cn of the varactor diode 3. [0147] In Inventive Example 2, the rate of change of the capacity Cn necessary for tuning the oscillation frequency in the range of 9.0 to 11.0 GHz with reference to 10 GHz is 0.84 to 1.24. In other words, Inventive Example 2 attains the same tuning range as Inventive Example 1 and comparative example with smaller change of the capacity Cn of the varactor diode 3. [0148] In the voltage-controlled oscillator according to this embodiment, as hereinabove described, the rate of change (width of change) of the oscillation frequency is increased with respect to change of the capacity Cn of the varactor diode 3. In other words, the rate of change (width of change) of the oscillation frequency is increased with respect to change of the control voltage. Therefore, a broadband tuning range can be attained. [0149] Further, a desired range can be selected from the broadband tuning range, whereby the tuning range can be optimized. [0150] In addition, a range having excellent linearity of the capacity-control voltage characteristic of the varactor diode 3 can be selectively used due to the broadband tuning range. Thus, linearity of change of the oscillation frequency is improved with respect to change of the control voltage for the voltage-controlled oscillator. [0151] The necessary tuning range can be ensured with small change of the capacity Cn of the varactor diode 3, whereby capacity change in the tuning range as well as a series resistance component of the varactor diode 3 are reduced. Therefore, fluctuation of the phase noise characteristic resulting from change of the oscillation frequency is reduced. [0152] Further, the Q-value of the tuning part 200 as well as the phase noise characteristic are improved by performing tuning in a region (deep bias region) of the control voltage providing a small series resistance component of the varactor diode 3. [0153] [0153]FIG. 15 is a plan view more specifically showing an exemplary structure of the voltage-controlled oscillator shown in FIG. 1. FIG. 16 is a circuit diagram of the voltage-controlled oscillator shown in FIG. 15. [0154] The voltage-controlled oscillator shown in FIGS. 15 and 16 is a voltage-controlled oscillator module obtained by mounting a packaged transistor, a semiconductor element such as a varactor diode and chip components such as a resistor and a capacitor on a dielectric substrate 1 of alumina ceramic having a circuit pattern formed by a thin-film conductor such as gold. Referring to FIG. 15, rectangles “C” denote chip capacitors, and rectangles “R” denote chip resistors. [0155] A transistor 4 of GaAs is formed on the dielectric substrate 1. A grounding conductor is formed on the rear surface of the dielectric substrate 1. A gate-side feedback microstrip line 5, a drain-side feedback microstrip line 6 and an output microstrip line 7 are formed on the dielectric substrate 1. A gate electrode G, a drain electrode D and a source electrode S of the transistor 4 are connected to first ends of the gate-side feedback microstrip line 5, the drain-side feedback microstrip line 6 and the output microstrip line 7 respectively. [0156] A sectorial tuning stub (radial stub) 2 is formed on the dielectric substrate 1. Low characteristic impedance can be readily implemented by employing the sectorial tuning stub 2. A second end of the microstrip line 5 is connected to an anode A of a varactor diode 3. A cathode C of the varactor diode 3 is connected to a first end of the tuning stub 2. In the example shown in FIG. 15, a second end of the tuning stub 2 is opened. [0157] The cathode C of the varactor diode 3 is connected to a pad electrode 12 receiving a positive control voltage VC(+) through a control bias resistor 11 and connected to a pad electrode 21 receiving a ground potential GND through a resistor #1. The control bias resistor 11 and the resistor #1 form a control voltage application circuit. The control bias resistor 11 has a large resistance value of 10 kΩ, for example. Therefore, the control voltage VC(+) can be applied to the cathode C of the varactor diode 3 without leaking any signal. [0158] The gate electrode G of the transistor 4 is connected to a pad electrode 20 receiving the ground potential GND through a band rejection filter 13 rejecting passage of a high frequency of the oscillation band and a terminal resistor 14. [0159] A second end of the microstrip line 6 is opened. The drain electrode D of the transistor 4 is connected to a pad electrode 15 through a band rejection filter 16 rejecting passage of the high frequency of the oscillation band. The pad electrode 15 is grounded through a bypass capacitor #2. A drain bias Vdd is applied to the pad electrode 15. [0160] The source electrode S of the transistor 4 is connected to a pad electrode 21 receiving the ground potential GND through a band rejection filter 18 rejecting passage of the high frequency of the oscillation band and a self-bias resistor 19. [0161] A second end of the microstrip line 7 is connected to an output node 10 through an attenuator 8 and a dc component removing circuit 9. The dc component removing circuit 9 is formed by a capacitor. The second end of the microstrip line 7 is provided with a radial stub #3 having an opened forward end forming a feedback circuit. [0162] In the voltage-controlled oscillator of this example, an impedance transformer 210 is formed with a radial tuning stub 2, provided with an opened forward end, having low characteristic impedance corresponding to 28 Ω thereby attaining a broadband tuning range. [0163] The control voltage VC(+) is divided by the resistors thereby reducing a voltage applied to the varactor diode 3. [0164] [0164]FIG. 17 is a circuit diagram showing the structure of the control voltage application circuit. [0165] Referring to FIG. 17, it is assumed that Rx, Ry and Rz denote the resistance values of the resistors 11 and #1 and a resistor #4 respectively. When the resistance values Rx, Ry and Rz of the resistors 11, #1 and #4 are sufficiently large, currents Ivc and Ivd flowing through the resistors 11 and #4 respectively are expressed as follows: I vc=(V Cc�α)/(R x +R y) [0166] where VC, equal to VCc�α, denotes a variable voltage. I vd =Vdd/(Rz+Ry) [0167] A voltage VVD applied across the varactor diode 3 is expressed as follows: V VD = ( I vd + I vc )  R y = { Vdd / ( R z + R y ) + ( V Cc � α ) / ( R x + R y ) }  R y ( 1 ) [0168] It is understood from the above equation that the voltage VVD applied across the varactor diode 3 can be controlled by the control voltage VC(+) and the drain bias Vdd. [0169] In this example, the resistance values Rx, Ry and Rz of the resistors 11, #1 and #4 are set as follows: [0170] Rx=10[KΩ]
[0173] In this example, the resistor #4 is not connected, and the drain bias Vdd is not used. [0174] In this case, ⅓ of the control voltage VC(+) is applied across the varactor diode 3. In other words, the voltage VVD is equal to VC(+)/3. [0175] [0175]FIG. 18 illustrates control voltage dependency of the oscillation frequency fo. Referring to FIG. 18, a line {circle over (1)} shows a characteristic obtained when attaining a broadband tuning range, and a line {circle over (2)} shows a characteristic obtained by extending control voltage change necessary for obtaining the same tuning range by improving the control voltage application circuit from a range A of 0 to 3 V to a range B of 0 to 9 V. Thus, precision of a voltage value required to the control voltage in tuning is relaxed as follows: [0176] In the case of the characteristic {circle over (1)}, the tuning range is about 1 GHz at the oscillation frequency fo of 7.08 to 8.04 [GHz] when change of the control voltage VC is 0 to 9 V. When change of the control voltage VC is 0 to 3 V, the tuning range is about 0.4 GHz at the oscillation frequency fo of 7.08 to 7.48 [GHz]. In the case of the characteristic {circle over (2)}, on the other hand, the tuning range is about 0.4 GHz at the oscillation frequency fo of 7.08 to 7.48 [GHz] when change of the control voltage VC is 0 to 9 V. [0177] Thus, precision of the voltage value required to the control voltage in tuning is so relaxed that the oscillation frequency can be correctly controlled. [0178] [0178]FIG. 19 illustrates control voltage dependency of tuning sensitivity Kv. The tuning sensitivity Kv indicates the rate of change of the oscillation frequency with respect to change of the control voltage VC. [0179] As shown in FIG. 19, linearity of the tuning sensitivity Kv (fluctuation of the tuning sensitivity with respect to the control voltage) can be apparently improved by improving the characteristic {circle over (1)} to the characteristic {circle over (2)}. [0180] As hereinabove described, the tuning range can be optimized bys electing a desired range in the broadband tuning range. [0181] The voltage-controlled oscillator, provided in the form of a module in this example, may alternatively be formed by a monolithic oscillator integrated on a semiconductor substrate. [0182] [0182]FIG. 20 is a model diagram showing the structure of a transmitter/receiver for a radio communication system employing the voltage-controlled oscillator shown in FIG. 1. [0183] The transmitter/receiver shown in FIG. 20 is formed by a transmission system 500, a receiving system 510, a local oscillator 520, a signal path switch 530 and an antenna 550. [0184] The transmission system 500 includes a frequency converter 501, an amplifier 502 and a band-pass filter 503. The receiving system 510 includes a frequency converter 511, an amplifier 512 and a band-pass filter 513. The local oscillator 520, formed by the voltage-controlled oscillator shown in FIG. 1, generates a reference signal of a prescribed oscillation frequency. [0185] The frequency converter 501 of the transmission system 500 mixes a transmitted signal TS of a prescribed frequency with the reference signal generated from the local oscillator 520, thereby converting the same to a transmitted signal of a high frequency. The amplifier 502 amplifies the transmitted signal obtained by the frequency converter 501. The band-pass filter 503 passes a signal component of a prescribed band in the transmitted signal amplified by the amplifier 502, and supplies the same to the antenna 550 through the signal path switch 530. Thus, the antenna 550 transmits a radio wave of a microwave or a quasi-millimeter wave. [0186] The radio wave of a microwave or a quasi-millimeter wave received by the antenna 550 is supplied to the band-pass filter 513 of the receiving system 510 through the signal path switch 530. The band-pass filter 513 passes a received signal of a prescribed band included in the radio wave. The amplifier 512 amplifies the received signal passed through the band-pass filter 513. The frequency converter 501 mixes the received signal amplified by the amplifier 512 with the reference signal generated from the local oscillator 520, thereby converting the same to a received signal RS of a low frequency. [0187] The transmitter/receiver shown in FIG. 20 employs the voltage-controlled oscillator shown in FIG. 1 as the local oscillator 520, whereby a broadband tuning range can be attained. [0188] The oscillation frequency of the local oscillator 520 linearly changes with respect to the control voltage, while fluctuation of the phase noise characteristic resulting from change of the oscillation frequency is reduced. Further, the phase noise characteristic of the local oscillator 520 is improved. [0189] Consequently, fluctuation of the communication quality resulting from switching of the frequency band or the channel allocated to the communication system as well as deterioration of the communication quality resulting from phase noise are reduced. [0190] (Method of Deriving Equation) [0191] Methods of deriving the above equations (A2) and (B2) are now described with reference to FIGS. 6 and 7. [0192] It is assumed that the resistance component of the variable reactance circuit 220 is zero, and only the capacity of the varactor diode 3 is taken into consideration as the capacitive component. [0193] It is further assumed that Zstub denotes the impedance of the tuning stub 2 as viewed from the node Q between the variable reactance circuit 220 and the impedance transformer 210. [0194] As described above, ωx denotes the oscillation angular frequency at the arbitrary control voltage Vx, ωC denotes the oscillation angular frequency (center oscillation angular frequency) at the central control voltage VC, and ωn denotes the oscillation angular frequency normalized with the oscillation angular frequency ωC, where ωn=ωx/ωC. [0195] Cx denotes the capacity of the varactor diode 3 at the arbitrary control voltage Vx, CC denotes the capacity (central capacity) of the varactor diode 3 at the central control voltage VC, and Cn denotes the capacity of the varactor diode 3 normalized with the capacity CC, where Cn=Cx/CC. At the central control voltage VC, ωn=1 and Cn=1. [0196] Zst denotes the characteristic impedance [Ω]of the tuning stub 2, and θst denotes the electric length [rad] of the tuning stub 2 at the center oscillation angular frequency ωC. [0197] (When Employing Open Stub) [0198] The impedance Zstub of the tuning stub 2 as viewed from the node Q between the variable reactance circuit 220 and the impedance transformer 210 is expressed as follows: Z stub =−jZ st cot(ωn�θst) (a1) [0199] The impedance Zmod of the tuning part 200 as viewed from the node P between the oscillation part 100 and the tuning part 200 is expressed as follows: Z mod =j{−Z st cot(ωn�θst)+X VD }=jX mod (a2) [0200] where XVD denotes the reactance of the variable reactance circuit 220. From the above equation (a2), the reactance Xmod of the tuning part 200 as viewed from the node P between the oscillation part 100 and the tuning part 200 is expressed as follows: X mod =−Z st cot(ωn�θst)+X VD (a3) [0201] Assuming that Lp denotes the inductance of the variable reactance circuit 220, the inductive component of the variable reactance circuit 220 is expressed as follows: ωn �L p=ωx�ωC �L p=ωn �A (a4) [0202] where A=ωC�Lp=constant [0203] The capacitive component of the variable reactance circuit 220 is expressed as follows: 1 / ( ω x � C x ) = 1 / ( ω n � ω c � C n � C C ) = B / ( ω n � C n ) ( a5 ) [0204] where B=1/(ωC/CC)=constant [0205] From the above equations (a4) and (a5), the reactance XVD of the variable reactance circuit 220 is expressed as follows: X VD = ω x � L p - 1 / ( ω x � C x ) = ω n � A - B / ( ω n � C n ) = X VD  ( ω n , C n ) ( a6 ) [0206] Thus, the reactance XVD of the variable reactance circuit 220 is expressed in the functions of the oscillation angular frequency ωn and the capacity Cn. [0207] Setting A/B=ω2 C�Lp�CC=α, the variable reactance circuit 220 is inductive when α>1, capacitive when α<1, and series-resonant when α=1. [0208] From the above equation (a6), the reactance XVD(ωn, Cn) of the variable reactance circuit 220 is expressed as follows: X VD(ωn , C n)=B{ω n�α−1/(ωn �C n)} (a7) [0209] From the above equations (a3) and (a7), therefore, the reactance Xmod(ωn, Cn) is expressed as follows: X mod  ( ω n , C n ) = - Z st  cot  ( ω n � θ st ) + X VD  ( ω n , C n ) = - Z st  cot  ( ω n � θ st ) + B ( ω n � α - 1 / ( ω n � C n ) } ( a8 ) [0210] The voltage-controlled oscillator is so designed that the reactance Xmod is shorted at the center oscillation angular frequency (ωn=1) and the central capacity (Cn=1). Thus, the reactance Xmod(1, 1) is expressed as follows when ωn=1 and Cn=1 from the above equation (a8): X mod(1, 1)=−Z st cot θst +B(α−1)=0 (a9) [0211] The following equation holds from the above equation (a9): Z st =B(α−1)/cot θst =B(α−1)tan θst (a10) [0212] From the above equations (a8) and (a10), therefore, the reactance Xmod is expressed as follows: X mod(ωn , C n)=B{(1−α)tan θst cot(ωn�θst)+ωn�α−1/(ωn �C n)} (a11) [0213] The oscillation angular frequency (ωx=ωn�ωC) is given by an oscillation angular frequency ωn satisfying Xmod(ωn, Cn)=0 when the capacity Cn of the varactor diode 3 changes. [0214] Considering a capacity Cn satisfying Xmod(ωn, Cn)=0 when the oscillation angular frequency ωn changes, the following equation holds from the above equation (a11): (1−α)tan θst cot(ωn�θst)+ωn�α−1/(ωn �C n)=0 (a12) [0215] The above equation (a12) is rearranged with the capacity Cn as follows: 1/(ωn �C n)=ωn�α+(1−α)tan θst cot(ωn�θst) (a13) [0216] Further, the following equation is obtained from the above equation (a13): C n(ωn)=1/[ωn{ωn�α+(1−α)tan θst cot(ωn�θst)}] (a14) [0217] Differentiating the capacity Cn(ωn) of the above equation (a14) by the oscillation angular frequency ωn, the rate Cn′(1) of change of the capacity at the center oscillation angular frequency (ωC=1) is obtained as follows: C n′(1)=(1−α)�2θst/sin(2θst)−(1+α) (A2) [0218] (When Employing Short Stub) The impedance Zstub of the tuning stub 2 as viewed from the node Q between the variable reactance circuit 220 and the impedance transformer 210 is expressed as follows: Z stub =jZ st tan(ωn�θst) (b1) [0219] The impedance Zmod of the tuning part 200 as viewed from the node P between the oscillation part 100 and the tuning part 200 is expressed as follows: Z mod =j{Z st tan(ωn�θst)+X VD }=jX mod (b2) [0220] where XVD denotes the reactance of the variable reactance circuit 220. From the above equation (b2), the reactance Xmod of the tuning part 200 as viewed from the node P between the oscillation part 100 and the tuning part 200 is expressed as follows: X mod =Z st tan(ωn�θst)+X VD (b3) [0221] Assuming that Lp denotes the inductance of the variable reactance circuit 220, the inductive component of the variable reactance circuit 220 is expressed as follows: ωn �L p=ωx�ωC �L p=ωn �A (b4) [0222] where A=ωC�Lp=constant [0223] The capacitive component of the variable reactance circuit 220 is expressed as follows: 1 / ( ω x � C x ) = 1 / ( ω n � ω C � C n � C C ) = B / ( ω n � C n ) ( b5 ) [0224] where B=1/(ωC/CC)=constant. [0225] From the above equations (b4) and (b5), the reactance XVD of the variable reactance circuit 220 is expressed as follows: X VD = ω x � L p - 1 / ( ω x � C x ) = ω n � A - B / ( ω n � C n ) = X VD  ( ω n , C n ) ( b6 ) [0226] Thus, the reactance XVD of the variable reactance circuit 220 is expressed in the functions of the oscillation angular frequency ωn and the capacity Cn. [0227] Setting A/B=ω2 C�Lp�CC=α, the variable reactance circuit 220 is inductive when α>1, capacitive when α<1, and series-resonant when α=1. [0228] From the above equation (b6), the reactance XVD(ωn, Cn) of the variable reactance circuit 220 is expressed as follows: X VD(ωn , C n)=B{ω n�α−1/(ωn �C n)} (b7) [0229] From the above equations (b3) and (b7), therefore, the reactance Xmod(ωn, Cn) is expressed as follows: X mod  ( ω n , C n ) = Z st  tan  ( ω n � θ st ) + X VD  ( ω n , C n ) = Z st  tan  ( ω n � θ st ) + B  { ω n � α - 1 / ( ω n � C n ) } ( b8 ) [0230] The voltage-controlled oscillator is so designed that the reactance Xmod is shorted at the center oscillation angular frequency (ωn=1) and the central capacity (Cn=1). Thus, the reactance Xmod(1, 1) is expressed as follows when ωn=1 and Cn=1 from the above equation (b8): X mod(1, 1)=Z st tan θst +B(α−1)=0 (b9) [0231] The following equation holds from the above equation (b9): Z st =B(1−α)/tan θst=B(1−α)cot θst (b10) [0232] From the above equations (b8) and (b10), therefore, the reactance Xmod is expressed as follows: X mod(ωn , C n)=B{(1−α)cot θst tan(ωn�θst)+ωn�α=1/(ωn �C n)} (b11) [0233] The oscillation angular frequency (ωx=ωn�ωC) is given by an oscillation angular frequency ωn satisfying Xmod(ωn, Cn)=0 when the capacity Cn of the varactor diode 3 changes. [0234] Considering a capacity Cn satisfying Xmod(ωn, Cn)=0 when the oscillation angular frequency ωn changes, the following equation holds from the above equation (b11): (1−α)cot θst tan(ωn�θst)+ωn�α−1/(ωn �C n)=0 (b12) [0235] The above equation (b12) is rearranged with the capacity Cn as follows: 1/(ωn �C n)=ωn�α+(1−α)cot θst tan(ωn�θst) (b13) [0236] Further, the following equation is obtained from the above equation (b13): C n(ωn)=1/[ωn{ωn�α+(1−α)cot θst tan(ωn�θst)}] (b14) [0237] Differentiating the capacity Cn(ωn) of the above equation (b14) by the oscillation angular frequency ωn, the rate Cn′(1) of change of the capacity at the center oscillation angular frequency (ωC=1) is obtained as follows: C n′(1)=(α−1)�2θst/sin(2θst)−(1+α) (B2) [0238] Thus, the characteristic impedance Zst of the tuning stub 2 and the rate Cn′(1) of change of the capacity of the varactor diode 3 at the center oscillation angular frequency (ωC=1) are obtained as follows, where α=ω2 C�Lp�CC and series resonance is attained when α is equal to 1: [0239] (when employing an open stub) Z st =B(α−1)tan θst (A1) C n′(1)=(1−α)�2θst/sin(2θst)−(1+α) (A2) [0240] (when employing a short stub) Z st =B(1−α)cot θst (B1) C n′(1)=(α−1)�2θst/sin(2θst)−(1+α) (B2) [0241] Although the present invention has been described and illustrated in detail, it is clearly understood that the same is by way of illustration and example only and is not to be taken by way of limitation, the spirit and scope of the present invention being limited only by the terms of the appended claims. Referenced byCiting PatentFiling datePublication dateApplicantTitleUS6949982Mar 5, 2004Sep 27, 2005Paratek Microwave, Inc.Voltage controlled oscillators incorporating parascan R varactorsUS8108900 *Dec 31, 2008Jan 31, 2012Echostar Technologies L.L.C.Drift compensator for a tuning deviceUS20040233005 *Mar 5, 2004Nov 25, 2004Du Toit Nicolaas D.Voltage controlled oscillators incorporating parascan R varactorsUS20090172748 *Dec 31, 2008Jul 2, 2009Echostar Technologies L.L.C.Drift Compensator for a Tuning DeviceEP1852969A1 *Feb 2, 2006Nov 7, 2007Kabushiki Kaisha Kobe Seiko ShoNegative resistance input amplifier circuit and oscillation circuitWO2004082127A2 *Mar 4, 2004Sep 23, 2004Paratek Microwave, Inc.Voltage controlled oscillators and synthesizers incorporating parascan� varactorsWO2004082127A3 *Mar 4, 2004Apr 14, 2005Paratek Microwave IncVoltage controlled oscillators and synthesizers incorporating parascan� varactors* Cited by examinerClassifications U.S. Classification331/100International ClassificationH03B5/18Cooperative ClassificationH03B5/1852European ClassificationH03B5/18F1BLegal EventsDateCodeEventDescriptionFeb 5, 2002ASAssignmentOwner name: SANYO ELECTRIC CO., LTD., JAPANFree format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:IMAOKA, TOSHIKAZU;ONODA, KATSUAKI;REEL/FRAME:012545/0813Effective date: 20020115Apr 18, 2007REMIMaintenance fee reminder mailedSep 30, 2007LAPSLapse for failure to pay maintenance feesNov 20, 2007FPExpired due to failure to pay maintenance feeEffective date: 20070930RotateOriginal ImageGoogle Home - Sitemap - USPTO Bulk Downloads - Privacy Policy - Terms of Service - About Google Patents - Send FeedbackData provided by IFI CLAIMS Patent Services