Source: https://patents.google.com/patent/JP2009142116A/en
Timestamp: 2020-08-15 09:17:20
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JP2009142116A - Position sensorless controller of permanent magnetic motor - Google Patents
Position sensorless controller of permanent magnetic motor Download PDF
JP2009142116A
JP2009142116A JP2007317827A JP2007317827A JP2009142116A JP 2009142116 A JP2009142116 A JP 2009142116A JP 2007317827 A JP2007317827 A JP 2007317827A JP 2007317827 A JP2007317827 A JP 2007317827A JP 2009142116 A JP2009142116 A JP 2009142116A
JP5130031B2 (en
Shigehisa Aoyanagi
2007-12-10 Application filed by Hitachi Car Eng Co Ltd, Hitachi Industrial Equipment Systems Co Ltd, Hitachi Ltd, 株式会社日立カーエンジニアリング, 株式会社日立産機システム, 株式会社日立製作所 filed Critical Hitachi Car Eng Co Ltd
2009-06-25 Publication of JP2009142116A publication Critical patent/JP2009142116A/en
2013-01-30 Publication of JP5130031B2 publication Critical patent/JP5130031B2/en
238000004364 calculation methods Methods 0.000 claims description 59
<P>PROBLEM TO BE SOLVED: To achieve stable operation without stepping out even in a low-speed rotation region even when a setting error (R-R<SP>*</SP>) occurs between a resistance value R obtained by adding a winding resistance value of a motor and a wiring resistance value between an inverter (power converter) and the motor, and R<SP>*</SP>set in a control system (vector operation, axis error estimation). <P>SOLUTION: In the estimation operation of an axial error, without using the setting value of a resistor, a d-axis voltage command value is added to a multiplication value of three signal values such as a q-axis current detection value or current command value, an inductance value and a speed estimation value, and the resulting value is divided by a multiplication value of a speed demand value ω<SB>r</SB><SP>*</SP>and an induced voltage constant Ke<SP>*</SP>, or an arctangent operation is performed on that value. <P>COPYRIGHT: (C)2009,JPO&INPIT
In the position sensorless vector control system of the permanent magnet motor, the present invention is based on the atmospheric temperature at which the motor is installed, the load state of the motor, and the extension / reduction of the wiring between the motor and the power converter (inverter). The present invention relates to a control technique that suppresses a “step-out phenomenon in a low-speed rotation region” due to “change in resistance component” and realizes vector control with low sensitivity to the resistance component.
As a technique of the position sensorless vector control method, as described in Japanese Patent Application Laid-Open No. 2001-251889, an error value (hereinafter referred to as a phase value θ of a permanent magnet motor) and a “phase estimation value θc * based on a control axis” is described. There is known a method for estimating the axial error Δθ) by calculation.
This method uses the voltage command value (Vdc * , Vqc * ), the current detection value (Idc, Iqc), and the speed command value (ω 1 c), which are the vector control outputs, in accordance with (Equation 1 ) to determine the axis error. This is a method for performing the estimation calculation.
Vdc * : d-axis voltage command value, Vqc * : q-axis voltage command value
Idc: d-axis current detection value, Iqc: q-axis current detection value
R: Addition value of the winding resistance value of the motor and the wiring resistance value of the motor and the power converter
Ke: Induced voltage constant *: Set value Further, the estimated speed value ω 1 c is controlled so that the estimated value Δθc of the axis error is “zero”, and the phase estimated value θc * is generated by performing an integration process. Yes.
JP 2001-251889 A
In the vector control calculation and the axis error estimation calculation, it is necessary to set a resistance value R obtained by adding the winding resistance value of the motor and the wiring resistance value between the power converter (inverter) and the motor.
If a setting error (R−R * ) has occurred in the set value R * , if an impact load disturbance or the like enters in the low speed rotation range, the actual phase error value Δθ and the estimated phase error value Δθc will be shifted. As a result, the optimum phase could not be controlled, and a step-out state (operation impossible) occurred.
One feature of the present invention is that the d-axis voltage command value Vdc ** is replaced with the q-axis current detection value Iqc or the current command value Iq without using the resistance set value R * in the axis error estimation calculation. * , Adding the multiplication value of three signals such as the inductance value Lq * and the estimated speed value ω 1 c, and dividing the calculated value by the multiplied value of the speed command value ω r * and the induced voltage constant Ke *. It is characterized by.
It is possible to provide a position sensorless control device for a permanent magnet motor that can realize a stable operation without stepping out even if there is a setting error of the motor constant.
FIG. 1 shows a configuration example of a “permanent magnet motor position sensorless control device” according to an embodiment of the present invention.
The permanent magnet motor 1 outputs a motor torque obtained by combining a torque component due to the magnetic flux of the permanent magnet and a torque component due to the inductance of the armature winding.
The power converter 2 outputs a voltage proportional to the three-phase AC voltage command values Vu * , Vv * , Vw * , and varies the output voltage and the rotational speed of the permanent magnet motor 1.
The current detector 3 detects the three-phase AC currents Iu, Iv, Iw of the permanent magnet motor 1.
The coordinate conversion unit 4 outputs the detected current values Idc and Iqc of the d-axis and the q-axis from the detected values Iuc, Ivc and Iwc of the three-phase alternating currents Iu, Iv and Iw and the estimated phase value θc * .
The axis error estimator 5 outputs the low-pass filter output values Id * td and Iq * td of the voltage command values Vdc * and Vqc * , the speed estimate value ω 1 c, the speed command value ω r * , the current command values Id * and Iq *. Based on the motor constant (Lq, Ke), an estimation calculation of an axis error which is a deviation between the phase estimation value θc * and the motor phase value θ is performed, and an estimation value Δθc is output.
The speed estimation unit 6 outputs a speed estimation value ω 1 c from the deviation between the axis error command value Δθc * which is “zero” and the axis error estimation value Δθc.
The phase estimation unit 7 integrates the speed estimation value ω 1 c and outputs the phase estimation value θc * to the coordinate conversion units 4 and 13.
The speed control unit 8 outputs the q-axis current command value Iq * from the deviation between the speed command value ω r * and the estimated speed value ω 1 c.
The d-axis current command setting unit 9 outputs a d-axis current command value Id * which is “zero” in the low-speed rotation range.
The q-axis current control unit 10 outputs a second q-axis current command value Iq ** from the deviation between the first q-axis current command value Iq * and the detected current value Iqc.
The d-axis current control unit 11 outputs a second d-axis current command value Id ** from the deviation between the first d-axis current command value Id * and the detected current value Idc.
Based on the electric constants (R, Ld, Lq, Ke) of the permanent magnet motor 1, the second current command values Id ** , Iq **, and the estimated speed value ω 1 c, the vector control unit 12 Outputs q-axis voltage command values Vdc * and Vqc * .
The coordinate conversion unit 13 outputs three-phase AC voltage command values Vu * , Vv * , Vw * from the voltage command values Vdc * , Vqc * and the phase estimation value θc * .
The low-pass filter 14 receives the q-axis current command value Iq * and outputs the current command value Iq * td used for the axis error estimation unit 5.
First, the basic voltage and phase control method will be described.
The basic operation of the voltage control is performed in vector control using the first current command values Id * and Iq * and the current detection values Idc and Iqc given from the host in the d-axis and q-axis current control units 10 and 11. The intermediate second current command values Id ** and Iq ** to be used are calculated.
The vector control unit 12 uses the second current command values Id ** , Iq ** , the estimated speed value ω 1 c, and the set value of the motor constant, and the voltage command values Vdc * , Vqc * shown in (Equation 2) . To control the three-phase voltage command values Vu * , Vv * , Vv * of the inverter.
On the other hand, with respect to the basic operation of phase control, the axis error estimation unit 5 uses the low-pass filter output value Iq * td of the voltage command value Vdc * and current command value Iq * , the estimated speed value ω 1 c, and the estimated speed command value ω r *. And the motor constant set values (Lq * , Ke * ), the estimation calculation of the axis error value Δθ (= θc * −θ) that is the deviation between the phase estimation value θc * and the motor phase value θ is ).
Further, the speed estimation unit 6 controls the speed estimation value ω 1 c by the calculation shown in (Equation 4) so that the estimated value Δθc of the axis error is “zero”.
Here, Kp: proportional gain, Ki: integral gain The proportional gain Kp and the integral gain Ki are set as shown in (Equation 4).
Here, N: Ratio of breakage point of proportional / integral gain of speed estimation unit 6 [times]
ω PLL : Control response angular frequency [rad / s] of the speed estimation unit 6
The phase estimation unit 7 controls the phase estimation value θc * by the calculation shown in (Equation 6) using the speed estimation value ω 1 c.
Next, before the “axis error estimator 5” which is a feature of the present invention, the operation characteristics of “when the conventional axis error estimation method (Formula 1) is used” will be described.
FIG. 2 shows “operating characteristics when R * set in the calculation of (Equation 1) and the actual resistance value R coincide with each other”.
Set the speed command value ω r * to the low speed (2.5% of the rated speed) and apply the load torque τ L in a ramp from point a (zero) to point b (100% load) in the figure. To do.
During the period in which the load torque τ L is applied (between a and b), the motor speed ω r decreases by a predetermined value, but after the point b, the speed command value ω r * is followed and stable operation is performed. Has been realized.
In the vicinity of points a and b in the figure, it can be seen that the axis error is generated in an absolute value of about 3 to 4 degrees.
On the other hand, FIG. 3 shows “operating characteristics in the case where there is a setting deviation between R * set for the calculation of the number (1) and the actual resistance value R (R / R * = 0.5)”.
The axial error Δθ occurring in the vicinity of the point c in the figure occurs about 5 degrees, which is about twice that in the case of FIG. From the load torque value of about 40%, it can be seen that the shaft error Δθ is oscillating and that the step out is performed at the point d.
That is, if there is a setting deviation between R * set in the speed estimation unit 6 and the actual resistance value R, a step-out phenomenon may occur.
Next, the “cause of the step-out phenomenon” will be described.
Substituting the voltage command values (Vd ** , Vq ** ) of (Equation 2) into the calculation formula of the conventional axis error estimation shown in (Equation 1),
Here, using the motor constants (R, Ld, Lq, Ke) and the control constants (R * , Ld * , Lq * , Ke * ) set in the vector control unit 12 and the axis error estimation unit 5, the q axis When the output value Iq ** of the current control unit 10 and the output Id ** of the d-axis current control calculation unit 11 are expressed, (Equation 8) is obtained.
Substituting (Equation 8) into (Equation 7),
If it is assumed in (Equation 9) that the change in Δθ is small, it can be approximated as cos Δθ≈1, sin Δθ≈Δθ.
Here, when d-axis current command value Id * = 0 is set, current control is performed, and q-axis inductance Lq is known (Lq * = Lq), (Equation 10) can be obtained.
In (Equation 10), focusing on the resistance value R of the denominator term,
In the case of R = R * , Δθc = Δθ,
In the case of R ≠ R * , it can be seen that the estimated value Δθc includes a resistance setting error component [(R−R * ) · Iqc].
When the motor speed ω r is a predetermined value, the shaft error Δθ is constantly zero, but Δθ is generated during a transient such as a change in ω r .
When Δθ occurs when the q-axis current detection value Iqc is “positive”,
The denominator value of (Equation 10) becomes large, and the estimated value Δθc is calculated to be smaller than Δθ.
In other words, even if an axial error Δθ occurs, the control system recognizes that the change width of Δθc is small (the change in θc * is small and the error with θ is large), so that it is easy to step out. End up.
The denominator value of (Equation 10) becomes smaller, and the estimated value Δθc is calculated to be larger than Δθ.
The resistance setting error component [(R−R * ) · Iqc] has a “negative” polarity and cancels the induced voltage component [ω 1 c · Ke], so that the denominator value decreases. When the denominator value becomes “zero”, the step-out state occurs.
That is, in the low-speed rotation range, if there is a resistance setting error (R−R * ), there is a problem that the stepping out easily occurs.
From here, it becomes the characteristic of this invention.
The configuration of the axis error estimation unit 5 will be described. The axis error estimation unit 5 will be described with reference to FIG.
In the axis error estimator 5, the numerator 51 calculates the d-axis voltage command value Vdc * from the low-pass filter output of the frequency estimate value ω 1 c and the q-axis current command value and the q-axis inductance value Lq that is a motor constant. The multiplication results of the three signals of the set value Lq * are added.
The denominator calculating unit 52 multiplies two signals of the speed command value ω r * and an induced voltage constant Ke * that is a motor constant.
A division calculation is performed between the numerator calculation value and the denominator calculation value, and an axis error estimated value Δθc is output. If the change in the axis error Δθ is large, Δθ can be estimated with high accuracy by performing the arctangent calculation shown in (Equation 11) instead of the division calculation.
In the present invention, focusing on the denominator value of (Equation 10) and reducing the sensitivity to the resistance setting error (R−R * ), the calculation of the denominator value is ω r * · Ke * . A multiplication value is used.
That is, the above-described (Equation 3) or (Equation 11) is used in the calculation of the estimated value Δθc in the axis error estimation unit 5.
Here, the effect of axial error estimation using (Equation 3), which is a feature of the present invention, will be described.
FIG. 5 shows the characteristics when R> R * (R / R * = 2), and FIG. 6 shows the characteristics when R> R * (R / R * = 0.5).
5 and 6, the sensitivity is reduced with respect to the resistance setting error (R−R * ), and it can be seen that the operation is stably performed even when R ≠ R * .
At this time, the axial error Δθ occurs in an absolute value of about 5 deg in both FIGS.
It can be seen that the magnitude of the axis error Δθ does not change even when (R−R * ) exists.
The round transfer function G ω — est (S) from the motor speed ω r to the estimated speed value ω 1 c can be expressed by the equation (12).
Furthermore, since the integral value of the error between the estimated speed value ω 1 c and the motor speed ω r becomes a shaft error, the transfer function G PLL (S) from the motor speed ω r to the estimated shaft error value Δθ c is If 5) is also used, (Equation 13) is obtained.
Here, N is a ratio / folding ratio [multiple] of the proportional / integral gain of the speed estimation unit 6 and is usually about 5 times. When “5” is substituted for “N” in (Equation 13), 14) is obtained.
Here, the control response of the speed estimation calculation is about 10 rad / s to about 1000 rad / s, and if this numerical value is substituted into the ω PLL of (Equation 14) and the axis error Δθ and the waveform substantially match, It can be proved that the calculation method of (Equation 3) or (Equation 11) is used.
In this embodiment, the output signal of the low-pass filter 14 is used for the axis error estimation unit 5, but the control gain of the speed control unit 8 is low, and the movement of Iq * that is the output signal is slow. For example, instead of Iq * td , Iq * may be directly used to perform an estimation calculation of Δθc. Further, the current detection value Iq may be used instead of Iq * td .
Further, in this embodiment, the speed command value ω r * is used for the calculation of the denominator term of the axis error estimation unit 5. However, when the speed command value ω r * changes suddenly, the speed command value ω r * is changed to ω r * . A low-pass filter corresponding to the control gain of the speed control unit 8 may be passed and this signal may be used to perform an estimation calculation of Δθc. Then, instead of the speed command value ω r * , the speed estimated value ω 1 c may be used.
In the first embodiment, paying attention to the denominator term of the axis error estimator 5 and reducing the sensitivity to the resistance setting error (R−R * ), the calculation of the denominator requires ω r * .Ke. Although the multiplication value of * is used, this embodiment is a method of controlling only by the numerator term of the axis error estimation calculation.
FIG. 7 shows this embodiment. In the figure, 1-4, 6-14, and 21 are the same as those in FIG. The configuration of the axis error information estimation unit 5a will be described.
The axis error information estimation unit 5a will be described with reference to FIG.
In the axis error information estimation unit 5a, the d-axis voltage command value Vdc * is set to the frequency estimate value ω 1 c and the low-pass filter output Iq * td of the q-axis current command value and the set value of the q-axis inductance value Lq that is a motor constant. The result of multiplication of the three signals Lq * is added.
This calculation is performed to output an axis error estimated value Δθc. The axis error information estimation unit 5a calculates and outputs the voltage value Δed * including the axis error Δθ by using (Equation 15).
In (Equation 15), when cos Δθ≈1, sin Δθ≈Δθ is approximated and the q-axis inductance Lq is known (Lq * = Lq), (Equation 16) can be obtained.
Also in this case, the sensitivity can be reduced with respect to the resistance setting error (R−R * ) with a simple configuration, and even when R ≠ R * , the operation can be stably performed.
So far, in the first and second embodiments, the second current command values Id ** and Iq ** are created from the first current command values Id * and Iq * and the detected current values Idc and Iqc. The vector control calculation was performed using this current command value.
1) Voltage correction values ΔVd * and ΔVq * are generated from the current detection values Idc and Iqc in the first current command values Id * and Iq * , and the voltage correction values and the first current command values Id * , Voltage command values Vdc * and Vqc * are calculated according to (Equation 17) using Iq * , estimated speed value ω 1 c, and electric constants (R * , Ld * , Lq * , Ke * ) of permanent magnet motor 1. Vector control calculation method,
2) The first d-axis current command Id * (= 0), the first-order lag signal Iqc td and the speed command value ω r * of the q-axis current detection value Iqc, the electric constants (R * , Ld ) of the permanent magnet motor 1 (* , Lq * , Ke * ) can also be applied to a control calculation method for calculating voltage command values Vdc * , Vqc * according to (Equation 18).
In this embodiment as well, the output signal of the low-pass filter 14 is used for the axis error estimator 5, but the control gain of the speed controller 8 is low, and the movement of the output signal Iq * is slow. For example, instead of Iq * td , Iq * may be directly used to perform an estimation calculation of Δθc. Further, the current detection value Iq may be used instead of Iq * td . Then, instead of the speed command value ω r * , the speed estimated value ω 1 c may be used.
In the first and second embodiments, the three-phase AC currents Iu to Iw detected by the expensive current detector 3 are detected. However, the first and second embodiments are mounted for detecting the overcurrent of the power converter 2. The three-phase motor currents Iu ^, Iv ^, Iw ^ can be reproduced from the direct current flowing through the one-shunt resistor, and the "low-cost system" using this reproduced current value can be dealt with.
As described above, the present invention provides the resistance value R obtained by adding the winding resistance value of the motor, the wiring resistance value of the inverter (power converter) and the motor, and the control system (vector calculation, axis error estimation). Even when a setting error (R−R * ) occurs between R * to be set, even in a low-speed rotation region, stable operation can be realized without step-out.
In addition, a “permanent magnet motor position sensorless control device” that can be commonly applied to a system that performs inexpensive current detection can be provided.
The block diagram of the position sensorless control apparatus of the permanent magnet motor which shows one Example of this invention. Operating characteristics when there is a setting deviation between the actual resistance value R and the control system setting value R * when using the conventional axis error estimation calculation (R / R * = 1). Operating characteristics when there is a setting deviation between the actual resistance value R and the control system setting value R * (R / R * = 0.5). The block diagram of the axis | shaft error estimation calculation which is the characteristics of this invention. Operating characteristics (R / R * = 2) when there is a setting deviation between the actual resistance value R and the control system setting value R * when the axis error estimation calculation of the present invention is used. Operating characteristics in the case where there is a setting deviation between the actual resistance value R and the control system setting value R * when the axis error estimation calculation of the present invention is used (R / R * = 0.5). The block diagram of the position sensorless control apparatus of the permanent magnet motor which shows the other Example of this invention. The block diagram of the axial error information estimation calculation which is the characteristics of this invention.
DESCRIPTION OF SYMBOLS 1 Permanent magnet motor 2 Power converter 3 Current detectors 4 and 13 Coordinate conversion part 5 Axis error estimation part 6 Speed estimation part 7 Phase estimation part 8 Speed control part 9 d-axis current command setting part 10 q-axis current control part 11 d Axis current control unit 12 Vector control unit 14 Low-pass filter Id * first d-axis current command value Id ** second d-axis current command value Iq * first q-axis current command value Iq ** second q-axis Current command value Vdc * d-axis voltage command value Vqc * q-axis voltage command value Idc d-axis current detection value Iqc q-axis current detection value θc Motor phase value θc * phase estimation value Δθ axis error Δθc Estimated value Δed * Voltage value including shaft error Δθ ω r Motor speed ω 1 c Speed estimated value
Deviation between the vector control calculation that controls the output frequency and output voltage of the power converter that drives the permanent magnet motor, and the phase estimated value obtained by integrating the speed estimated value of the permanent magnet motor and the phase value of the permanent magnet motor Axis error information estimation calculation for estimating axis error information including information on a certain axis error, and a speed estimation calculation for controlling the estimated value of the axis error information so as to match the command value of the estimated value of the axis error information In the permanent magnet motor position sensorless control device,
In the axis error information estimation calculation,
Axis error information is estimated using a vector control voltage command value, a q-axis current detection value or current command value, a motor constant inductance value, and a speed estimation value or speed command value. Position sensorless control device for permanent magnet motor.
Axis error information estimation calculation is
The d-axis voltage command value, which is the output of the vector control calculation, is
A permanent magnet for estimating the axis error information by adding a current detection value or current command value of a q-axis and a product of three signals such as an inductance value and a speed estimation value or a speed command value. Motor position sensorless control device.
Deviation between the phase control value obtained by integrating the speed estimation value of the permanent magnet motor and the phase value of the permanent magnet motor by vector control calculation that controls the output frequency and output voltage of the power converter that drives the permanent magnet motor In a position sensorless control device for a permanent magnet motor having an axis error estimation calculation for estimating a certain axis error, and a speed estimation calculation for controlling the axis error estimated value so as to match the command value of the axis error,
In the axis error estimation calculation,
Axis error is estimated using vector control voltage command value, current detection value or current command value, motor constant inductance value, induced voltage constant, speed estimation value or speed command value. A position sensorless control device for a permanent magnet motor.
The axis error estimation calculation is
The d-axis voltage command value, which is the output of the vector control calculation, is added to the q-axis current detection value or current command value and the multiplication value of three signals such as the inductance value and the speed estimation value or speed command value. A position sensorless control device for a permanent magnet motor, wherein a value is divided by a product of a speed command value and an induced voltage coefficient.
The d-axis voltage command value, which is the output of the vector control calculation, is added to the q-axis current detection value or current command value and the multiplication value of three signals such as the inductance value and the speed estimation value or speed command value. A position sensorless control device for a permanent magnet motor, wherein the value is an arctangent calculation using a product of a speed command value and an induced voltage constant.
The q-axis current command value used for the axis error information estimation calculation is
A position sensorless control device for a permanent magnet motor, characterized by being a low-pass filter output value of a q-axis current command value given from a host.
The q-axis current command value used for the axis error estimation calculation is
To create a low-pass filter output value for the current command value,
A position sensorless control device for a permanent magnet motor, which is a first order lag signal based on a control response angular frequency or control gain set for current control.
A position sensorless control device for a permanent magnet motor, wherein the speed command value used for the axis error information estimation calculation is a low-pass filter output value of a speed command value given from a host.
A position sensorless control device for a permanent magnet motor, wherein the speed command value used for the axis error estimation calculation is a low-pass filter output value of a speed command value given from a host.
To create the low-pass filter output value of the speed command value,
A position sensorless control device for a permanent magnet motor, which is a first order lag signal based on a control response angular frequency or control gain set for speed control.
A position sensorless control device for a permanent magnet motor, wherein a d-axis current command value as a magnetic flux axis is zero in a low speed range of 5% or less of a rated speed.
Even if the wiring between the motor and the inverter is extended or shortened,
The axis error value Δθ should be generated by the following equation when the motor speed ω r changes, regardless of the resistance value set in the control system.
A position sensorless control device for a permanent magnet motor.
Here, ω p is a proportional gain or a control response angular frequency in the speed estimation calculation.
A position sensorless control device for a permanent magnet motor, wherein the proportional gain or the control response angular frequency is 10 rad / s to 1000 rad / s.
JP2007317827A 2007-12-10 2007-12-10 Position sensorless control device for permanent magnet motor Active JP5130031B2 (en)
US12/330,629 US8106619B2 (en) 2007-12-10 2008-12-09 Position sensorless controller for permanent magnet motor
JP5130031B2 JP5130031B2 (en) 2013-01-30
JP2007317827A Active JP5130031B2 (en) 2007-12-10 2007-12-10 Position sensorless control device for permanent magnet motor
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RU2507658C1 (en) * 2009-12-28 2014-02-20 Мицубиси Электрик Корпорейшн Device to control excitation of ac motor
JP2006087152A (en) * 2004-09-14 2006-03-30 Hitachi Ltd Controller and module of permanent magnet synchronous motor
JP2008167566A (en) * 2006-12-28 2008-07-17 Hitachi Industrial Equipment Systems Co Ltd High-response control device of permanent magnet motor
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