Source: http://www.google.com/patents/US6608527?dq=7493558
Timestamp: 2016-05-07 00:10:01
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Matched Legal Cases: ['Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60']

Patent US6608527 - Adaptive radio transceiver with low noise amplification - Google PatentsSearch Images Maps Play YouTube News Gmail Drive More »Sign inPatentsAn exemplary embodiment of the present invention described and shown in the specification and drawings is a transceiver with a receiver, a transmitter, a local oscillator (LO) generator, a controller, and a self-testing unit. All of these components can be packaged for integration into a single IC including...http://www.google.com/patents/US6608527?utm_source=gb-gplus-sharePatent US6608527 - Adaptive radio transceiver with low noise amplificationAdvanced Patent SearchPublication numberUS6608527 B2Publication typeGrantApplication numberUS 10/192,515Publication dateAug 19, 2003Filing dateJul 8, 2002Priority dateOct 21, 1999Fee statusPaidAlso published asUS6404293, US6417737, US7031668, US7555263, US7720444, US7970358, US8041294, US20030042984, US20030067359, US20060205374, US20090286487, US20100295598Publication number10192515, 192515, US 6608527 B2, US 6608527B2, US-B2-6608527, US6608527 B2, US6608527B2InventorsShervin Moloudi, Maryam RofougaranOriginal AssigneeBroadcom CorporationExport CitationBiBTeX, EndNote, RefManPatent Citations (25), Non-Patent Citations (9), Referenced by (52), Classifications (23), Legal Events (6) External Links: USPTO, USPTO Assignment, EspacenetAdaptive radio transceiver with low noise amplification
US 6608527 B2Abstract
The present application is a continuation of patent application Ser. No. 09/692,421, filed Oct. 18, 2000 now U.S. Pat. No. 6,417,737, which is continuation of co-pending patent application Ser. No. 09/634,552, filed Aug. 8, 2000, priority of which is hereby claimed under 35 U.S.C. �120. The present application also claims priority under 35 U.S.C. �119(e) to provisional Application Nos. 60/160,806, filed Oct. 21, 1999; Application No. 60/163,487, filed Nov. 4, 1999; Application No. 60/163,398, filed Nov. 4, 1999; Application No. 60/164,442, filed Nov. 9, 1999; Application No. 60/164,194, filed Nov. 9, 1999; Application No. 60/164,314, filed Nov. 9, 1999; Application No. 60/165,234, filed Nov. 11, 1999; Application No. 60/165,239, filed Nov. 11, 1999; Application No. 60/165,356; filed Nov. 12, 1999; Application No. 60/165,355, filed Nov. 12, 1999; Application No. 60/172,348, filed Dec. 16, 1999; Application No. 60/201,335, filed May 2, 2000; Application No. 60/201,157, filed May 2, 2000; Application No. 60/201,179, filed May 2, 2000; Application No. 60/202,997, filed May 10, 2000; Application No. 60/201,330, filed May 2, 2000. All these applications are expressly incorporated herein by referenced as though fully set forth in full.
P i,BP =BW�P i,LP +jω 0=αi �BW+j(ω0+βi �BW (6)
The complex filter is realized by cascading n biquad stages. Therefore, similar to real-domain bandpass filters, an nth order complex filter uses 2�n integrators. Based on equation (3), each biquad stage has a pole equal to −1/RC+j2Q/RC. Thus: α i � BW = - 1 RC   and ( 7 ) ω 0 + β i � BW = 2  Q RC ( 8 ) Since the LP equivalent poles are located in the left-half plane, ai is always negative. The above equations set the value of Q and RC in each stage. The gain of each biquad stage can be adjusted based on the desired gain in the complex filter, and noise-linearity trade-off: increasing the gain of one biquad stage lowers the noise contributed by the following biquad stages, but it also degrades the linearity of the complex filter.
Therefore, the RSSI maximum input level is S, and the ideal RSSI minimum input level is S/An, where A is the gain of each differential amplifier and n is the number of the differential amplifiers. Thus, the ideal dynamic range is calculated as follows: Ideal   Dynamic   Range = 20  log  S S A n = 20  log   A n = 20  ( n )  log   A ( 22 ) However, in the case of a large amount of gain, the input level will be limited with the input noise and the dynamic range will also be limited to: Dynamic   Range = 20  log  S σ n   σ n = total   noise   rms   σ n = ( BW ) � Noise   Factor ( 23 ) If each differential amplifier has the same input dynamic range VL and each full-wave rectifier has similar input dynamic range Vi, then the dynamic range of the logarithmic differential amplifier and the total RSSI circuitry are the same.
This is further simplified to: RSSI = ( A   β ) 2 ( A   β ) 2 - 1  V in 2  [ ( A   β ) 2  ( n - m - 1 )  _ ] + m   β 2  S 2 ( 26 ) RSSI ≈ 1 ( A   β ) 2 - 1  V in 2  ( A   β ) 2  ( n - m ) + m   β 2  S 2 ( 27 ) The above equation is a first order approximation to the logarithmic function shown in equation (28) according to the first two terms of the Taylor expansion at a given operating point.
ΔRSSI=C log A 2n (30) C
To find the relation between the gain of a differential amplifier, the gain of a rectifier, and the maximum input range of the combined differential amplifier and the rectifier, the RSSI will be calculated for the two consecutive differential amplifier and rectifier combinations (see equations (33) and (34)) for both ideal RSSI equations (32) and approximated RSSI equation (27): V in1 = S ( A ) n - m ( 33 ) V in2 = S ( A ) n - m - 1 ( 34 ) (Ideal)RSSI 2 −RSSI 1=log(A)2 (35)
(Approximated)RSSI 2 −RSSI 1=β2 S 2 (36)
Using equations (18) and (12), the following expression is achieved: Δ   RSSI n = β 2  S 2 ( 38 ) Plugging equation (19) into (8) results in the following: RSSI = 1 ( A   β ) 2 - 1   ( A   β ) 2  ( n - m )  V in 2 + m   Δ   RSSI n ;   S A n - m < V in < S A n - m - 1 ( 39 ) FIG. 16(a) shows a schematic diagram for an exemplary embodiment of the differential amplifier used in the type II core amplifier. The differential input signal is fed to the gates of transistor amplifiers 955, 957. The amplified differential output signal is provided at the drains of the transistor amplifiers 955, 957. The gain of the transistor amplifiers is set by load transistors 958, 860, each connected between the drain of one of the transistor amplifiers and a power source. More particularly, the gain of the differential amplifier is determined by the ratio of the square root of transistor amplifiers-to-load transistors. Gain  ( A ) = w in w in = 200 6 ≈ 5.8 ( 40 ) The sources of the transistor amplifiers 955, 957 are connected in common and coupled to a constant current source transistor 952. In the described exemplary embodiment, the controller provides the bias to the gate of the transistor 952 to set the current.
Capacitances associated with bias resistors may also be addressed. Consider a typical distributed model for a polysilicon (“poly” for short) resistor. Around 4fF to substrate can be associated with every kilo-ohm of resistance in a poly resistor. This means that, for example in a 20K ohm resistor, around 80fF of distributed capacitance to the substrate exists. This can contribute to power loss because part of the power will be drained into the substrate. One way of biasing the input stage and the output stage is through a resistive voltage divider as shown in FIG. 26(a). The biasing of the input stage is shown for the transistor 616 in FIG. 25, however, those skilled in the art will readily appreciate that the same biasing circuit can be used for the transistor 614 (FIG. 25). One drawback from this approach, however, is that the gate of the transistor will see the capacitance from the two resistors 658, 660 of the voltage divider. Capacitor 662 is a coupling capacitor, which couples the previous stage to the voltage divider. Switch 664 is for powering down the stage of the power amplifier that is connected to the voltage divider. The switch 664 is on in normal operation and is off in power down mode.
where ω1 is the VCO radian frequency, and ω2 is the divider radian frequency, equal to 0.5ω1. By simplifying equation (25) and equation (26), the signals at the output of mixers will be: V out_I =  - Sin   ( θ 2 ) � Sin   ( ( ω 1 - ω 2 )  t + θ 2 ) +  Cos   ( θ 2 ) � Cos   ( ( ω 1 + ω 2 )  t + θ 2 )   and ( 52 ) V out_Q =  - Sin   ( θ 2 ) � Cos   ( ( ω 1 - ω 2 )  t + θ 2 ) +  Cos   ( θ 2 ) � Sin   ( ( ω 1 + ω 2 )  t + θ 2 ) ( 53 ) The above equations show that regardless of the value of θ, the outputs are always in quadrature. However, other effects should be evaluated. First, a spur at ω1−ω2=0.5ω1 is produced at the output. This spur can be attenuated by 2�22=44 dB by the LC filters at the mixer and its buffer outputs. Thus, for 60 dB rejection, the single sideband mixers need to provide an additional 16 dB of rejection (about 0.158). Based on equation (53), tan(θ/2)=0.158, or θ≈18�, phase accuracy of better than 18� can generally be achieved. Second, phase error at the VCO output lowers the mixer gain (term Cos(θ/2) in equation (52) or (53)). For a phase error of 18�, the gain reduction is, however, only 0.1 dB, which is negligible. For θ=90� (a single-phase VCO), both sidebands are equally upconverted at the mixer output. However, the LC filters reject the lower sideband by about 44 dB. The mixer gain will also be 3 dB lower. This will slightly increase the power consumption of the LO generator. If θ=180� (the VCO I and Q outputs are switched), the lower sideband is selected, and the desired sideband is completely rejected.
Patent CitationsCited PatentFiling datePublication dateApplicantTitleUS3909527Sep 18, 1973Sep 30, 1975Mitsubishi Electric CorpFrequency shift keying system and methodUS4055807Mar 25, 1976Oct 25, 1977Motorola, Inc.Antenna switchUS4283739Aug 30, 1979Aug 11, 1981Texas Instruments IncorporatedColor television receiversUS4315227Dec 5, 1979Feb 9, 1982Bell Telephone Laboratories, IncorporatedGeneralized switched-capacitor active filterUS4916411May 19, 1989Apr 10, 1990Hewlett-Packard CompanyVariable frequency jitter generatorUS5283484Oct 13, 1992Feb 1, 1994Motorola, Inc.Voltage limiter and single-ended to differential converter using sameUS5375257Dec 6, 1993Dec 20, 1994Raytheon CompanyMicrowave switchUS5404050 *Dec 9, 1993Apr 4, 1995U.S. Philips CorporationSingle-to-differential converterUS5412351Oct 7, 1993May 2, 1995Nystrom; ChristianQuadrature local oscillator networkUS5559475Mar 14, 1995Sep 24, 1996Mitsubishi Denki Kabushiki KaishaFrequency synthesizer for synthesizing signals of a variety of frequencies by cross modulationUS5574986Nov 16, 1994Nov 12, 1996U.S. Philips CorporationTelecommunication system, and a first station, a second station, and a transceiver for use in such a systemUS5590412Nov 18, 1994Dec 31, 1996Sanyo Electric Co., Ltd.Communication apparatus using common amplifier for transmission and receptionUS5736903Apr 24, 1996Apr 7, 1998Harris CorporationCarrier buffer having current-controlled tracking filter for spurious signal suppressionUS5778306Nov 8, 1996Jul 7, 1998Motorola Inc.Low loss high frequency transmitting/receiving switching moduleUS5878089Feb 21, 1997Mar 2, 1999Usa Digital Radio Partners, L.P.Coherent signal detector for AM-compatible digital audio broadcast waveform recoveryUS5878331Jul 30, 1996Mar 2, 1999Mitsubishi Denki Kabushiki KaishaIntegrated circuitUS5945878 *Feb 17, 1998Aug 31, 1999Motorola, Inc.Single-ended to differential converterUS6006112Nov 26, 1997Dec 21, 1999Lucent Technologies, Inc.Transceiver with RF loopback and downlink frequency scanningUS6016422Oct 31, 1997Jan 18, 2000Motorola, Inc.Method of and apparatus for generating radio frequency quadrature LO signals for direct conversion transceiversUS6094108 *Sep 15, 1998Jul 25, 2000Mitsubishi Denki Kabushiki KaishaUnbalance-to-balance converter utilizing a transistorUS6134430Dec 9, 1997Oct 17, 2000Younis; Saed G.Programmable dynamic range receiver with adjustable dynamic range analog to digital converterUS6150901Nov 20, 1998Nov 21, 2000Rockwell Collins, Inc.Programmable RF/IF bandpass filter utilizing MEM devicesUS6160449Jul 22, 1999Dec 12, 2000Motorola, Inc.Power amplifying circuit with load adjust for control of adjacent and alternate channel powerUS6308047Feb 3, 1999Oct 23, 2001Mitsubishi Denki Kabushiki KaishaRadio-frequency integrated circuit for a radio-frequency wireless transmitter-receiver with reduced influence by radio-frequency power leakageUS6441688 *Aug 18, 2000Aug 27, 2002Motorola, Inc.Single-to-differential buffer amplifier* Cited by examinerNon-Patent CitationsReference1Durham, A.M., "Circuit Architectures for High Linearity Monolithic Continuous-Time Filtering," IEEE Transactions on Circuits and Systems-II: Analog and Digital Signal-Processing, Sep. 1992, 7 pages, vol. 39, No. 9, IEEE.2Rofougaran, A. et al., "A 1 GHZ CMOS RF Front-End IC for a Direct-Conversion Wireless Receiver," IEEE Journal of Solid-State Circuits, Jul. 1996, 10 pages, vol. 31, No. 7, IEEE.3Rofougaran, A. et al., "A 1 GHz CMOS RF Front-End IC With Wide Dynamic Range," Integrated Circuits & Systems Laboratory Electrical Engineering Department University of California, Los Angeles, CA; European Solid-State Conf. Lille, France, Sep. 1995, pp. 250-253 of the conference proceedings.4Rofougaran, Ahmadreza, "A 900 MHz CMOS RF Power Amplifier With Programmable Output,"1994 Symposium on VLSI Circuits Digest of Technical Papers, 1994, pp. 133-134, Integrated Circuits & Systems Laboratory, Electrical Engineering Department, University of California, Los Angeles, CA.5Rofougaran, Ahmadreza, "A Single-Chip 900 MHz Spread-Spectrum Wireless Transceiver in 1-mum CMOS-Part I: Architecture and Transmitter Design," IEEE Journal of Solid-State Circuits, Apr. 1998, pp. 515-534, vol. 33, No. 4, IEEE.6Rofougaran, Ahmadreza, "A Single-Chip 900-MHz Spread-Spectrum Wireless Transceiver in 1-mum CMOS-Part II: Receiver Design," IEEE Journal of Solid-State Circuits, Apr. 1998, pp. 535-547, vol. 33, No. 4, IEEE.7Rofougaran, Ahmadreza, "A Single-Chip Spread-Spectrum Wireless Transceiver in CMOS," Final Report, Integrated Circuits & Systems Laboratory Electrical Engineering Department, University of California, Los Angeles, CA 1999.8Rofougaran, Ahmadreza, "A Single-Chip 900 MHz Spread-Spectrum Wireless Transceiver in 1-μm CMOS-Part I: Architecture and Transmitter Design," IEEE Journal of Solid-State Circuits, Apr. 1998, pp. 515-534, vol. 33, No. 4, IEEE.9Rofougaran, Ahmadreza, "A Single-Chip 900-MHz Spread-Spectrum Wireless Transceiver in 1-μm CMOS-Part II: Receiver Design," IEEE Journal of Solid-State Circuits, Apr. 1998, pp. 535-547, vol. 33, No. 4, IEEE.Referenced byCiting PatentFiling datePublication dateApplicantTitleUS6870432 *Jun 2, 2003Mar 22, 2005Intel CorporationUnilateral coupling for a quadrature voltage controlled oscillatorUS6972610Nov 17, 2004Dec 6, 2005Broadcom CorporationHigh linearity passive mixer and associated LO bufferUS6989705Oct 26, 2004Jan 24, 2006Broadcom CorporationHigh linearity passive mixer and associated LO bufferUS7050769 *Jun 14, 2002May 23, 2006Hitachi Ulsi Systems Co., Ltd.Electronic apparatus and design methodUS7061279 *Dec 30, 2004Jun 13, 2006Broadcom CorporationSystem and method for high frequency, high output swing buffersUS7102411 *Mar 6, 2003Sep 5, 2006Broadcom CorporationHigh linearity passive mixer and associated LO bufferUS7202740 *Jan 5, 2005Apr 10, 2007Broadcom CorporationGain boosting for tuned differential LC circuitsUS7323945Feb 10, 2005Jan 29, 2008Bitwave Semiconductor, Inc.Programmable radio transceiverUS7324615Dec 15, 2003Jan 29, 2008Microchip Technology IncorporatedTime signal receiver and decoderUS7375590 *Mar 15, 2006May 20, 2008Richwave Technology Corp.Single-ended input to differential-ended output low noise amplifier implemented with cascode and cascade topologyUS7378881 *Apr 11, 2003May 27, 2008Opris Ion EVariable gain amplifier circuitUS7421052 *May 27, 2005Sep 2, 2008Intel CorporationOscillator frequency selectionUS7432763Mar 26, 2007Oct 7, 2008Broadcom CorporationGain boosting for tuned differential LC circuitsUS7437345 *Sep 22, 2004Oct 14, 2008Sony CorporationImage rejection mixer and multiband generatorUS7482887Feb 10, 2005Jan 27, 2009Bitwave Semiconductor, Inc.Multi-band tunable resonant circuitUS7508898Aug 11, 2005Mar 24, 2009Bitwave Semiconductor, Inc.Programmable radio transceiverUS7580684Feb 10, 2005Aug 25, 2009Bitwave Semiconductor, Inc.Programmable radio transceiverUS7623886 *Nov 24, 2009NDSSI Holdings, LLCMethod and apparatus for transmitter calibrationUS7646254Jan 12, 2010Honeywell International Inc.Radiation hard oscillator and differential circuit designUS7672645Jun 15, 2006Mar 2, 2010Bitwave Semiconductor, Inc.Programmable transmitter architecture for non-constant and constant envelope modulationUS7751513Nov 3, 2006Jul 6, 2010Infineon Technologies AgSignal processing method, particularly in a radio-frequency receiver, and signal conditioning circuitUS7839219 *Aug 8, 2008Nov 23, 2010Industrial Technology Research InstituteLow-noise amplifier circuit including band-stop filterUS7962174 *Jun 14, 2011Andrew LlcTransceiver architecture and method for wireless base-stationsUS8324968 *Mar 30, 2011Dec 4, 2012Denso CorporationAmplifier circuit, signal processor circuit, and semiconductor integrated circuit deviceUS8410856Dec 17, 2010Apr 2, 2013Industrial Technology Research InstituteMethod and apparatus for canceling balun amplifier noiseUS8576012 *Mar 30, 2012Nov 5, 2013Realtek Semiconductor Corp.Single-to-differential conversion circuitUS20030022638 *Jun 14, 2002Jan 30, 2003Shun ImaiElectronic apparatus and design methodUS20040224728 *Sep 19, 2003Nov 11, 2004Sony CorporationMethod and system for power save mode in wireless communication systemUS20040251975 *Jun 2, 2003Dec 16, 2004Shenggao LiUnilateral coupling for a quadrature voltage controlled oscillatorUS20050073352 *Nov 17, 2004Apr 7, 2005Broadcom CorporationHigh linearity passive mixer and associated LO bufferUS20050088204 *Oct 26, 2004Apr 28, 2005Broadcom CorporationHigh linearity passive mixer and associated LO bufferUS20050159129 *Sep 22, 2004Jul 21, 2005Sony CorporationImage rejection mixer and multiband generatorUS20060145728 *Dec 30, 2004Jul 6, 2006John LeeteSystem and method for high frequency, high output swing buffersUS20060145762 *Jan 5, 2005Jul 6, 2006Broadcom CorporationGain boosting for tuned differential LC circuitsUS20060269014 *May 27, 2005Nov 30, 2006Intel CorporationOscillator frequency selectionUS20070111691 *Nov 6, 2006May 17, 2007Andre HankeSignal conditioning circuit, especially for a receiver arrangement for mobile radioUS20070116160 *Nov 3, 2006May 24, 2007Carsten EisenhutSignal processing method, particularly in a radio-frequency receiver, and signal conditioning circuitUS20070135058 *Dec 14, 2005Jun 14, 2007Tzero Technologies, Inc.Method and apparatus for transmitter calibrationUS20070188238 *Mar 15, 2006Aug 16, 2007Jiong-Guang SuSingle-Ended Input to Differential-Ended Output Low Noise Amplifier Implemented with Cascode and Cascade TopologyUS20070200628 *Mar 26, 2007Aug 30, 2007Broadcom CorporationGain boosting for tuned differential LC circuitsUS20080014866 *Jul 12, 2006Jan 17, 2008Lipowski Joseph TTransceiver architecture and method for wireless base-stationsUS20090002079 *Sep 8, 2008Jan 1, 2009Bitwave Semiconductor, Inc.Continuous gain compensation and fast band selection in a multi-standard, multi-frequency synthesizerUS20090058538 *Aug 30, 2007Mar 5, 2009Honeywell International Inc.Radiation Hard Oscillator and Differential Circuit DesignUS20090108944 *Aug 8, 2008Apr 30, 2009Industrial Technology Research InstituteLow-noise amplifier circuit including band-stop filterUS20090133083 *Nov 21, 2007May 21, 2009Texas Instruments IncorporatedPassive circuit for improved differential amplifier common mode rejectionUS20100003942 *Sep 19, 2007Jan 7, 2010Takeshi IkedaLoop antenna input circuit for am and am radio receiver using the sameUS20110241780 *Oct 6, 2011Denso CorporationAmplifier circuit, signal processor circuit, and semiconductor integrated circuit deviceUS20120249186 *Mar 30, 2012Oct 4, 2012Realtek Semiconductor CorporationSingle-to-differential conversion circuitCN100468955CFeb 21, 2006Mar 11, 2009立积电子股份有限公司Cascode and serial low-noise amplifier implemented by single-end input and differential outputDE102004021867A1 *May 4, 2004Dec 1, 2005Infineon Technologies AgVerfahren zur Signalverarbeitung, insbesondere in einem Hochfrequenzempf�nger und SignalaufbereitungsschaltungDE102004021867B4 *May 4, 2004Feb 16, 2012Infineon Technologies AgVerfahren zur Signalverarbeitung, insbesondere in einem Hochfrequenzempf�nger und SignalaufbereitungsschaltungDE102004022324A1 *May 6, 2004Dec 1, 2005Infineon Technologies AgSignalaufbereitungsschaltung, insbesondere f�r eine Empf�ngeranordnung f�r den Mobilfunk* Cited by examinerClassifications U.S. Classification330/301International ClassificationH03H11/12, H03H21/00Cooperative ClassificationH04B1/30, H03B21/01, H03H11/1291, H04B17/14, H03H21/0001, H04B1/40, H03J2200/10, H04B17/104, H03H2011/0494, H03H21/0012, H04B17/19European ClassificationH04B1/40, H04B1/30, H03B21/01, H04B17/00A3S, H04B17/00A1T, H04B17/00A2S, H03H21/00A, H03H21/00B, H03H11/12FLegal EventsDateCodeEventDescriptionMar 17, 2003ASAssignmentOwner name: CLEVELAND CLINIC FOUNDATION, THE, OHIOFree format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:MCCARTHY, PATRICK;REEL/FRAME:013844/0490Effective date: 20020925Feb 7, 2007FPAYFee paymentYear of fee payment: 4Jul 20, 2007ASAssignmentOwner name: BROADCOM CORPORATION, CALIFORNIAFree format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:MOLOUDI, SHERVIN;ROFOUGARAN, MARYAM;REEL/FRAME:019588/0106Effective date: 20001120Feb 1, 2011FPAYFee paymentYear of fee payment: 8Feb 19, 2015FPAYFee paymentYear of fee payment: 12Feb 11, 2016ASAssignmentOwner name: BANK OF AMERICA, N.A., AS COLLATERAL AGENT, NORTHFree format text: PATENT SECURITY AGREEMENT;ASSIGNOR:BROADCOM CORPORATION;REEL/FRAME:037806/0001Effective date: 20160201RotateOriginal ImageGoogle Home - Sitemap - USPTO Bulk Downloads - Privacy Policy - Terms of Service - About Google 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