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Timestamp: 2014-09-01 23:44:35
Document Index: 584935480

Matched Legal Cases: ['Application No.\n60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60']

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07/19/07 | Class 324 Monitor | RSS | Industry | Agents | Inventors
Linear capacitance measurement and touchless switch * PDF is temporarily not available for this patent. Please check back later. Thank you for your patience. Title: Linear capacitance measurement and touchless switch.Abstract: Capacitance measurement apparatus that enhances the sensitivity and accuracy of capacitive transducers, proximity sensors, and touchless switches. Each of two capacitors (C1, C2) under measurement has one end connected to ground and is kept at substantially the same voltage potential by operational amplifier (A1) or amplifiers (A0, A1) using negative feedback. The apparatus is driven by a periodic e.g. sinusoidal signal source (G1) or sources (G1, G2) and includes a difference amplifier (A2) operative to produce an electrical signal having a linear relationship with a specified arithmetic function of the capacitances of the two capacitors (C1, C2). A touchless switch is implemented using the capacitance measurement apparatus. The touchless switch includes two sensor electrodes (E1, E2) that correspond to the two capacitors (C1, C2) under measurement and in one embodiment has a front surface in the form of a container. ...
- Boston, MA, USInventor: Ying Lau LeeUSPTO Applicaton #: #20070164756 - Class: 324662000 (USPTO) - 07/19/07 - Class 324 The Patent Description & Claims data below is from USPTO Patent Application 20070164756, Linear capacitance measurement and touchless switch.Negative Feedback CROSS REFERENCE TO RELATED APPLICATIONS
[0001] This application is a continuation-in-part (CIP) application of
prior U.S. patent application Ser. No. 11/202,486 filed Aug. 12, 2005
entitled LINEAR CAPACITANCE MEASUREMENT AND TOUCHLESS SWITCH. This
application claims benefit of U.S. Provisional Patent Application No.
60/690,486 filed Jun. 15, 2005 entitled LINEAR CAPACITANCE MEASUREMENT
AND TOUCHLESS SWITCH, U.S. Provisional Patent Application No. 60/662,378
filed Mar. 17, 2005 entitled CAPACITANCE MEASUREMENT AND TOUCHLESS
SWITCH, U.S. Provisional Patent Application No. 60/619,697 filed Oct. 19,
2004 entitled DIFFERENTIAL CAPACITANCE MEASUREMENT AND TOUCHLESS SWITCH,
and U.S. Provisional Patent Application No. 60/601,610 filed Aug. 16,
2004 entitled DIFFERENTIAL CAPACITANCE MEASUREMENT AND TOUCHLESS SWITCH.
[0003] The present invention relates generally to capacitance measurement
apparatus and techniques, and more specifically to proximity detectors
such as touchless switches that employ capacitance measurement
[0004] In recent years, there has been an increasing need for improved
techniques of operating publicly accessible facilities and equipment
without requiring a user to make physical contact with a surface of a
manual activation device such as a touch switch. Such facilities and
equipment include elevators, vending machines, security access panels,
information terminals, etc. By not requiring a user to physically touch a
switch that may have been touched and contaminated by others who had
previously used the facilities or equipment, the spread of germs and
diseases may be significantly reduced.
[0005] For example, a user typically operates a public facility such as an
elevator by physically touching one or more switches, which may have been
previously touched by a substantial number of individuals. Some of these
individuals may have come from environments where they may have been
exposed to contaminants such as potentially harmful or contagious toxins
or pathogenic disease organisms. When such individuals make physical
contact with one or more of the switches required to operate an elevator,
there is a risk that the individuals may deposit contaminants onto the
surface of the switches, where they may remain viable for an extended
period of time. These contaminants may be later transferred from the
switches to subsequent elevator users who physically touch the switches,
thereby potentially causing the subsequent users to become afflicted with
diseases or other serious medical conditions.
[0006] During outbreaks of the severe acute respiratory syndrome (SARS) in
Asia, many members of the public were afraid to use any public facilities
that required them to touch a manual activation device such as a touch
switch. To mitigate the fears of the public, programs were instituted for
periodically cleaning and disinfecting the surfaces of these devices.
Such programs are typically ineffective, because no matter how well these
activation devices are cleaned and disinfected, they may become
contaminated once again by a subsequent user. As a result, the risk of
transferring potentially harmful contaminants from manual activation
devices such as touch switches to subsequent users of publicly accessible
facilities and equipment continues unabated. Capacitance-based proximity
detectors have been employed to implement activation devices that do not
require a user to physically touch a surface of the device. Such
proximity detectors operate according to the principle that an electric
field and a capacitance are generated between two conductive objects that
have different voltage potentials and are physically separated from one
another. The capacitance between the two conductive objects generally
increases as the surface areas of the objects increase, or as the
[0007] Conventional capacitance-based proximity detectors have drawbacks,
however, when they are used to implement a touchless switch. For example,
it is generally difficult to adjust the sensitivity of a
capacitance-based proximity detector to assure that a touchless switch
employing such a proximity detector can be reliably activated by a broad
range of users, and that the switch is not susceptible to noise and/or
environmental changes. This is due to the relatively small equivalent
capacitance that the capacitance-based proximity detector is required to
measure when implementing a touchless switch.
[0008] Specifically, when a human body is very near or proximate to a
sensor electrode of a capacitance-based proximity detector, the proximity
detector effectively measures the equivalent capacitance of two series
capacitors, assuming that the stray capacitance between the capacitance
sensing circuitry and circuit ground is ignored. One of the series
capacitors is formed between the sensor electrode and the human body, and
the other capacitor is formed between the human body and earth ground.
The amount of capacitance between the sensor electrode and the human body
depends primarily on the distance between them, and to a lesser extent on
the size and characteristics of the human body. For example, when the
human body is not very near the sensor electrode, the amount of
capacitance between the sensor electrode and the human body is
significantly smaller than the amount of capacitance between the human
body and ground. Accordingly, a touchless switch implemented using a
capacitance-based proximity detector must measure an equivalent
capacitance that is significantly smaller than the capacitance typically
measured by a conventional touch switch.
[0009] FIG. 1 depicts a touchless switch implemented using a
capacitance-based proximity detector 100 including a sensor electrode
112, capacitance sensing circuitry 114, and the equivalent capacitances
of the capacitors formed between a human finger and the sensor electrode
112 (C.sub.A), the rest of the human body and the sensor electrode 112
(C.sub.B), the human body and ground (C.sub.C), and the capacitance
sensing circuitry 114 and ground (C.sub.D), which in this analysis can be
ignored. When the human finger is proximate to the sensor electrode 112,
the capacitance between the human body and the sensor electrode 112 can
be taken as the sum of the capacitance C.sub.A between the finger and the
sensor electrode 112, and the capacitance C.sub.B between the rest of the
human body and the sensor electrode 112. If the human finger is not very
near the sensor electrode 112, then any changes in the capacitance
C.sub.A between the finger and the sensor electrode 112 are typically
very small. As a result, any extraneous common-mode disturbances
resulting from electrical noise or interference, changes in the
characteristics of the environment, changes in the capacitance C.sub.C
between the human body and ground, and/or changes in the capacitance
C.sub.B between the rest of the human body and the sensor electrode 112
due to changes in the distance between the rest of the human body and the
sensor electrode 112, changes in the size or characteristics of the human
body, etc., may be equal to or greater than the corresponding changes in
the capacitance C.sub.A between the human finger and the sensor electrode
112. Accordingly, if the sensitivity of the capacitance-based proximity
detector 100 is adjusted to be highly sensitive, then the proximity
detector 100 may be actuated unintentionally, due to the various
extraneous common-mode disturbances listed above. However, if the
capacitance-based proximity detector 100 has reduced sensitivity, then
the proximity detector 100 may be inoperable due to the inability to
detect the small amount of capacitance between the finger of a user and
the sensor electrode 112 at a reasonable distance.
[0010] A touch switch implemented using the capacitance-based proximity
detector 100 generally fares much better than a touchless switch because
when a human finger touches the surface of a touch switch, the area of
contact is typically much larger than just the area of a fingertip.
Further, the distance between a finger and a sensor electrode of the
touch switch is typically much smaller than the corresponding distance
between a finger and the sensor electrode 112 of the touchless switch,
even if the sensor electrode of the touch switch is disposed behind an
insulating surface. The changes in capacitance between a human finger and
the sensor electrode of a touch switch are therefore much larger than the
corresponding changes in capacitance between a human finger and the
sensor electrode 112 of a touchless switch. Accordingly, the problems
described above relating to the detection of changes in the capacitance
C.sub.A between a human finger and the sensor electrode 112 of the
touchless switch, e.g., the changes in the capacitance C.sub.B or C.sub.C
due to different users, are relatively insignificant in a touch switch.
[0011] One way of avoiding the problems described above relating to
extraneous common-mode disturbances in a touchless switch is to employ
known differential signal measurement techniques. Such differential
signal measurement techniques can be used in touchless switches that
include two sensor electrodes arranged so that the switch is actuated
when the capacitance between a human finger and one of the sensor
electrodes exceeds a preset threshold level relative to a second
capacitance between the finger and the other sensor electrode. By
directly comparing these first and second capacitances in a differential
measurement to determine whether to actuate the touchless switch,
extraneous common-mode disturbances that can adversely affect the
measurement can be effectively canceled out.
[0012] U.S. Pat. No. 6,310,611 filed Oct. 30, 2001 entitled DIFFERENTIAL
TOUCH SENSOR AND CONTROL CIRCUIT THEREFORE (the '611 patent) discloses a
touch sensor that employs a differential signal measurement technique. As
disclosed in the '611 patent, the touch sensor includes a first sensor
electrode, a second sensor electrode positioned proximate to the first
electrode, a differential circuit connected to the first and second
electrodes, and a pulse or other signal source configured to generate an
electric field between the first and second electrodes. Although the
touch sensor of the '611 patent is configured to perform a differential
measurement, the touch sensor does not operate by measuring capacitance.
Instead, the touch sensor measures changes in the voltage difference
between the two sensor electrodes caused by the introduction of an object
affecting the electric field around the two electrodes. The touch sensor
employs a differential circuit to provide an output signal that is
responsive to this difference in voltage between the two electrodes.
[0013] The touch sensor disclosed in the '611 patent has drawbacks,
however, when used to implement a touchless switch. For example, the
above-described changes in the voltage difference between the two sensor
electrodes of the touch sensor resulting from the introduction of an
object are caused by the interaction of the electric fields associated
with the sensor electrodes and the object. This interaction of electric
fields is relatively complex because the two sensor electrodes and the
object are at different voltage potentials, and there is no precise
relationship governing the voltage difference between the sensor
electrodes and the proximity of the object to the sensor electrodes.
Furthermore, the methods disclosed in the '611 patent to measure the
voltage difference between the sensor electrodes are only effective if
the voltage difference is significant enough as in the case of a touch
switch. Therefore, the approach disclosed in the '611 patent is not
precise or sensitive enough to be used in a touchless switch. U.S. Pat.
No. 6,456,477 filed Sep. 24, 2002 entitled LINEAR CAPACITANCE DETECTION
CIRCUIT (the '477 patent) discloses capacitance detection circuitry that
employs a differential signal measurement technique. As disclosed in the
'477 patent, the linear capacitance detection circuitry includes a
circuit that measures a difference in capacitance between a first
capacitor and a second capacitor by driving the two capacitors with
pulses. The capacitance detection circuitry further includes an
operational amplifier with negative feedback configured to maintain the
two capacitors at substantially equal voltage potentials. As a result,
there is a linear relationship between an electrical signal produced by
the operational amplifier and the ratio of the capacitances of the two
capacitors. The approach disclosed in the '477 patent also has drawbacks,
however, in that it requires pulse signals, which can introduce transient
noises and instability to the operational amplifier and can adversely
affect the accuracy of the operational amplifier output. Although low
pass filters and a feedback capacitor may be employed at the inputs of
the operational amplifier to mitigate the effects of transient noises and
instability, the addition of such components adversely affects the
accuracy and sensitivity of the capacitance detection circuitry.
[0014] It would therefore be desirable to have a capacitance measurement
apparatus and technique, and a proximity detector such as a touchless
switch employing a capacitance measurement technique, that avoid the
drawbacks of the above-described approaches.
[0015] In accordance with the present invention, a capacitance measurement
apparatus and technique are provided that can be employed to enhance the
sensitivity and accuracy of many different types of capacitive
transducers, proximity sensors, and touchless switches. The presently
disclosed capacitance measurement apparatus directly and accurately
produces a linear response to changes in each of the ratios of the
capacitance of a capacitor/capacitive transducer to the capacitance of
one or more other different capacitors/capacitive transducers with
adjustable offset, while maintaining all of the capacitors/capacitive
transducers at substantially identical voltage potentials at all times.
The presently disclosed capacitance measurement apparatus also produces a
linear response to changes in each of the differences between the
capacitance of a capacitor/capacitive transducer multiplied by a first
constant factor and the capacitance of one or more other different
capacitors/capacitive transducers each being multiplied by a respective
second constant factor, while maintaining all of the
capacitors/capacitive transducers at substantially identical voltage
potentials at all times.
[0016] Additionally, the presently disclosed capacitance measurement
apparatus directly and accurately produces a linear response to changes
in the capacitance, or changes in the reciprocal of the capacitance, of a
capacitor/capacitive transducer with adjustable offset, without requiring
special calibration or adjustment over a wide range of capacitance
values. The presently disclosed capacitance measurement apparatus also
provides a simple way of measuring the capacitance, or the reciprocal of
the capacitance, of a large number of capacitors/capacitive transducers,
or comparing the capacitance of a large number of capacitors/capacitive
transducers with the capacitance of a large number of sets of
capacitors/capacitive transducers.
[0017] The presently disclosed capacitance measurement apparatus employs a
plurality of operational amplifiers for maintaining the voltage
potentials of multiple capacitors/capacitive transducers undergoing
comparison or measurement at substantially the same voltage potential at
all times. Because the multiple capacitors/capacitive transducers are
maintained at substantially the same voltage potential, there is
essentially no capacitance between them. For this reason, the capacitance
measurement apparatus can be employed to measure small changes in the
capacitances of the multiple capacitors/capacitive transducers without
having adjacent capacitors/capacitive transducers affect the capacitance
measurement, even when the capacitors/capacitive transducers are
positioned relatively close to one another. In one embodiment, the
capacitance measurement apparatus includes a first operational amplifier
Al, and a second operational amplifier A2 configured as a difference
amplifier. A difference amplifier, as the term is used herein, refers to
a circuit or device that amplifies a difference between two input signals
and includes different types of differential DC amplifiers such as
instrumentation amplifiers, etc. Each of two capacitors C1 and C2
undergoing comparison or measurement has one end connected to circuit
ground, and another end connected to one of the differential inputs of
operational amplifier A1. Capacitor C1 is connected to the inverting
input of operational amplifier A1, and capacitor C2 is connected to the
non-inverting input of operational amplifier A1. Both capacitors C1 and
C2 are driven by the output of a periodic varying voltage source such as
a sinusoidal voltage source through respective resistors connected to
corresponding inputs of operational amplifier A1. A feedback resistor is
connected between the output of operational amplifier Al and its
inverting input. Due to the high open loop gain of operational amplifier
A1, capacitors C1 and C2 are maintained at substantially the same voltage
potential at all times. There is a linear relationship between the
magnitude of the current flowing through the feedback resistor and the
ratio of the capacitance of capacitor C1 to the capacitance of capacitor
C2. Further, the current flowing through the feedback resistor is
in-phase or out-of-phase with the currents flowing through the resistors
connected to the periodic varying voltage source, depending on whether
the ratio is less than or greater than a specified value. The phase and
magnitude of the current flowing through the feedback resistor can be
measured by difference amplifier A2, having one of its differential
inputs connected to the output of operational amplifier Al and another
differential input connected to one of the differential inputs of
operational amplifier A1.
[0018] In a second embodiment, the capacitance measurement apparatus
includes a first operational amplifier A1, and a second operational
amplifier A2 configured as a difference amplifier. Each of two capacitors
C1 and C2 undergoing comparison or measurement has one end connected to
circuit ground, and another end connected to one of the differential
inputs of operational amplifier A1. Capacitor C1 is connected to the
inverting input of operational amplifier A1, and capacitor C2 is
connected to the non-inverting input of operational amplifier A1. The
non-inverting input of operational amplifier Al is driven directly by the
output of a first periodic varying current source such as a sinusoidal
current source, while the inverting input of operational amplifier Al is
driven directly by a second periodic varying current source whose output
is K (a constant) times that of the first periodic varying current
source. A feedback resistor is connected between the output of
operational amplifier A1 and its inverting input. Due to the high open
loop gain of operational amplifier A1, capacitors C1 and C2 are
maintained at substantially the same voltage potential at all times.
There is a linear relationship between the magnitude of the current
flowing through the feedback resistor and the ratio of the capacitance of
capacitor C1 to the capacitance of capacitor C2. Further, the current
flowing through the feedback resistor is in-phase or out-of-phase with
the outputs of the periodic varying current sources, depending on whether
the ratio is less than or greater than the value of K. The phase and
inputs connected to the output of operational amplifier A1 and another
[0019] In a third embodiment, the capacitance measurement apparatus
includes first and second operational amplifiers A0 and A1, and a third
operational amplifier A2 configured as a difference amplifier. Each of
two capacitors C1 and C2 undergoing comparison or measurement, having
capacitances of c1 and c2, respectively, has one end connected to circuit
ground, and another end connected to the inverting input of operational
amplifier A0 or A1. Capacitor C1 is connected to the inverting input of
operational amplifier A1, and capacitor C2 is connected to the inverting
input of operational amplifier A0. The non-inverting inputs of
operational amplifiers A0 and A1 are both driven directly by a periodic
varying voltage source such as a sinusoidal voltage source. A first
feedback resistor R1 having a resistance of r1 is connected between the
output of operational amplifier A1 and its inverting input. A second
feedback resistor R2 having a resistance of r2 is connected between the
output of operational amplifier A0 and its inverting input. Due to the
high open loop gain of operational amplifiers A0 and A1, the two
capacitors C1 and C2 are maintained at substantially the same voltage
potential as the periodic varying voltage source at all times. The output
of operational amplifier A1 is connected to the non-inverting input of
difference amplifier A2, and the output of operational amplifier A0 is
connected to the inverting input of difference amplifier A2. The output
of difference amplifier A2 is proportional to (r1*c1-r2*c2), and is
in-phase or out-of-phase with the currents flowing through resistors R1
and R2, depending on whether (r1*c1-r2*c2) is greater than or less than
[0020] Each embodiment of the presently disclosed capacitance measurement
apparatus can be configured for comparing the capacitance of a
capacitor/capacitive transducer to the capacitance of a plurality of
other different capacitors/capacitive transducers, while maintaining all
of the capacitors/capacitive transducers at substantially the same
voltage potential. Additionally, by switching the respective
capacitors/capacitive transducers in and out for subsequent comparison or
measurement, each embodiment of the capacitance measurement apparatus can
sequentially measure the capacitance, or the reciprocal of the
capacitance, of multiple capacitors/capacitive transducers, or compare
the capacitance of multiple capacitors/capacitive transducers to the
capacitance of multiple sets of capacitors/capacitive transducers.
[0021] Touchless switches and proximity sensors are also provided that
employ embodiments of the presently disclosed capacitance measurement
apparatus. The touchless switches are configured to be actuated by a
human finger or a finger-like object, requiring the finger or finger-like
object to reach a specified boundary before actuating the switch. The
touchless switches have reduced susceptibility to unintended actuations,
and reduced sensitivity to changes in environmental factors such as
temperature, humidity, etc., and to electrical noise, while having a
simple and rugged construction. The touchless switches can be used in
hygiene-sensitive applications, industrial control panels, and a wide
variety of facilities and equipment accessible to the general public,
including but not limited to elevators, vending machines, security access
panels, information terminals, etc.
[0022] In one embodiment, the touchless switch includes a front surface,
and two adjacent sensor electrodes maintained at substantially the same
voltage potential disposed on or behind the front surface of the switch.
As a result, there is substantially no capacitance between the two sensor
electrodes, and therefore the sensor electrodes operate essentially
independent of one another. One of the sensor electrodes is a center
electrode, and the other sensor electrode is an outer electrode. The
center electrode is spaced from and at least partly surrounded by the
outer electrode. When the tip of a human finger or finger-like object is
near or proximate to the center electrode, the presence of the finger or
finger-like object can be detected using an embodiment of the capacitance
measurement apparatus disclosed herein. The capacitance measurement
apparatus can be employed to measure the ratio of the capacitance of the
two sensor electrodes with respect to the finger or finger-like object,
or the difference between the capacitance of one sensor electrode with
respect to the finger or finger-like object multiplied by a first
constant factor, and the capacitance of the other sensor electrode with
respect to the finger or finger-like object multiplied by a second
constant factor, thereby substantially canceling out extraneous
common-mode disturbances such as the capacitance between the rest of the
human body and the sensor electrodes, the capacitance between the human
body and earth ground, environmental changes, electrical noise, etc.,
which tend to affect both sensor electrodes equally due to their close
proximity to one another. The capacitance ratio and difference
measurements performed by the capacitance measurement apparatus are
facilitated by the fixed geometrical shape, size, and relative positions
of the two sensor electrodes. The outer electrode may be placed in front
of the center electrode so that initially, when the finger or finger-like
object moves toward the center electrode, the capacitance ratio or
difference measurement is less than a preset threshold. As the finger or
finger-like object moves closer to the center electrode, the capacitance
ratio or difference measurement eventually exceeds the preset threshold,
thereby actuating the switch. The touchless switch may also include a
guard electrode surrounding the back and sides of the two sensor
electrodes. The guard electrode and the sensor electrodes are maintained
at substantially the same voltage potential so that each sensor electrode
forms a capacitor only with objects disposed in front of it. Leads
extending from the two sensor electrodes to the capacitance measurement
apparatus may also be guarded using a twin-axial cable or two coaxial
cables, in which the outer conductors of the cables are employed as guard
shields and maintained at substantially the same voltage potential as the
inside cable conductors connected to the sensor electrodes. The front
surface of the touchless switch may take the form of the surface of a
container, in which the brim of the container surface defines an
imaginary boundary plane that the finger or finger-like object must reach
to actuate the switch.
[0023] The presently disclosed capacitance measurement apparatus may be
employed to detect the proximity of conductive objects larger than a
human finger such as the palm of a human hand. The presently disclosed
capacitance measurement apparatus may also be employed to detect the
location, position, and/or movement of a conductive object such as a
human appendage within a specified area.
[0024] Other features, functions, and aspects of the invention will be
evident from the Detailed Description of the Invention that follows.
[0025] The invention will be more fully understood with reference to the
following Detailed Description of the Invention in conjunction with the
[0026] FIG. 1 illustrates various equivalent capacitances formed between a
human body, earth ground, and a sensor electrode coupled to capacitance
[0027] FIG. 2a is a schematic diagram of first capacitance measurement
circuitry according to the present invention;
[0028] FIG. 2b is a schematic diagram of circuitry employing the first
capacitance measurement circuitry of FIG. 2a, for producing a linear
response to changes in each of the ratios of the capacitance of a
capacitor to the capacitance of one or more other different capacitors;
[0029] FIG. 3a is a schematic diagram of second capacitance measurement
[0030] FIG. 3b is a schematic diagram of circuitry employing the second
capacitance measurement circuitry of FIG. 3a, for producing a linear
[0031] FIG. 4a is a schematic diagram of third capacitance measurement
[0032] FIG. 4b is a schematic diagram of circuitry employing the third
capacitance measurement circuitry of FIG. 4a, for producing a linear
response to changes in each of the differences between the capacitance of
a capacitor multiplied by a first constant factor and the capacitance of
one or more other different capacitors, after each is multiplied by a
respective second constant factor;
[0033] FIGS. 5a-5d are perspective views of illustrative shapes of a front
surface of a touchless switch according to the present invention;
[0034] FIGS. 6a-6d are cross-sectional views of illustrative arrangements
and relative positions of two sensor electrodes and the front surfaces of
the touchless switches of FIGS. 5a-5d, respectively;
[0035] FIG. 7 is a cross-sectional view of an illustrative arrangement and
relative positions of two sensor electrodes, a front surface, and a guard
electrode of a touchless switch according to the present invention;
[0036] FIG. 8a is a diagram of a touchless switch employing the first
capacitance measurement circuitry of FIG. 2a;
[0037] FIG. 8b is a diagram of a set of touchless switches employing the
first capacitance measurement circuitry of FIG. 2a;
[0038] FIG. 9a is a diagram of a touchless switch employing the second
capacitance measurement circuitry of FIG. 3a;
[0039] FIG. 9b is a diagram of a set of touchless switches employing the
second capacitance measurement circuitry of FIG. 3a;
[0040] FIG. 10a is a diagram of a touchless switch employing the third
capacitance measurement circuitry of FIG. 4a; and
[0041] FIG. 10b is a diagram of a set of touchless switches employing the
third capacitance measurement circuitry of FIG. 4a.
[0042] The entire disclosures of U.S. patent application Ser. No.
11/202,486 filed Aug. 12, 2005 entitled LINEAR CAPACITANCE MEASUREMENT
AND TOUCHLESS SWITCH, U.S. Provisional Patent Application No. 60/690,486
filed Jun. 15, 2005 entitled LINEAR CAPACITANCE MEASUREMENT AND TOUCHLESS
SWITCH, U.S. Provisional Patent Application No. 60/662,378 filed Mar. 17,
2005 entitled CAPACITANCE MEASUREMENT AND TOUCHLESS SWITCH, U.S.
Provisional Patent Application No. 60/619,697 filed Oct. 19, 2004
entitled DIFFERENTIAL CAPACITANCE MEASUREMENT AND TOUCHLESS SWITCH, and
U.S. Provisional Patent Application No. 60/601,610 filed Aug. 16, 2004
entitled DIFFERENTIAL CAPACITANCE MEASUREMENT AND TOUCHLESS SWITCH, are
[0043] A capacitance measurement apparatus and technique are disclosed
that can be employed to enhance the sensitivity and accuracy of many
different types of capacitive transducers, proximity sensors, and
touchless switches. FIG. 2a depicts a first illustrative embodiment of
capacitance measurement circuitry 200a, in accordance with the present
invention. In the illustrated embodiment, the capacitance measurement
circuitry 200a includes a periodic varying voltage source G1, a first
operational amplifier A1, and a second operational amplifier A2
configured as a difference amplifier. Each of two capacitors/capacitive
transducers C1 and C2 undergoing comparison, having capacitances of c1
and c2, respectively, has one end connected to circuit ground, and
another end connected to one of the differential inputs of operational
amplifier A1. Capacitor C1 is connected to the inverting input of
operational amplifier A1 at node 101, and capacitor C2 is connected to
the non-inverting input of operational amplifier A1 at node 102. The
nodes 101 and 102 are driven by an output Vs of the periodic varying
voltage source G1, which may be a sinusoidal voltage source, through
resistors R1 and R2, respectively. Resistor R1 has a resistance r1, and
resistor R2 has a resistance r2. Output V1 of operational amplifier A1 is
fed back to the inverting input of operational amplifier A1 via feedback
resistor R3 having a resistance of r3. Because operational amplifier A1
has a very high open loop gain, the two inputs of operational amplifier
A1 are maintained at substantially the same voltage potential, thereby
causing the effective RC time constants for capacitors C1 and C2 at nodes
101 and 102 to be substantially the same. The magnitude i3 of current I3
flowing through resistor R3 is substantially equal to the magnitude i2 of
current I2 flowing into capacitor C2 multiplied by the factor
(r2/r1-c1/c2), i.e., i3=i2*(r2/r1-c1/c2). Current I3 flowing through
resistor R3 will be in-phase or out-of-phase with current I1 flowing
through resistor R1 and current I2 flowing through resistor R2, depending
on whether the ratio of the capacitances c1/c2 is less than or greater
than the value r2/r1. More specifically, if c1/c2 is less than r2/r1,
then (r2/r1-c1/c2) is positive and the currents I2 and I3 will be
in-phase but if c1/c2 is greater than r2/r1, then (r2/r1-c1/c2) is
negative and the currents I2 and I3 will be out of phase. At steady
state, the magnitude i2 of current I2 flowing into capacitor C2 is a
function of time, and therefore the magnitude i3 of current I3 flowing
through resistor R3 at a fixed time of a cycle of current I3 is an
accurate measure of the value (r2/r1-c1/c2). The voltage across resistor
R3 is equal to i3*r3, and is equivalent to the difference of the voltage
potential between node 101 (or node 102) and output V1 of operational
amplifier A1. This voltage can be measured by connecting node 101 (or
node 102) to one of the two inputs of difference amplifier A2, and by
connecting output V1 of operational amplifier A1 to the other input of
difference amplifier A2. It should be noted that the configuration of
difference amplifier A2, as shown in FIG. 2a, is described herein for
purposes of illustration, and that other suitable circuit configurations
may be employed. For example, alternative configurations of difference
amplifier A2 may include more than one operational amplifier. Output Vd
of difference amplifier A2 is proportional to the magnitude i3 of current
I3 flowing through resistor R3, and will be in-phase with currents I1 and
I2 when c1/c2 is greater than r2/r1. It is noted that the phase of output
Vd reverses if the inputs to difference amplifier A2 are interchanged.
[0044] Output Vd of difference amplifier A2 is therefore proportional to a
signal representing current I2 modulated by the value (c1/c2-r2/r1). If
current I2 is sinusoidal, then the change in the ratio of the
capacitances c1/c2 can be measured using a synchronous demodulator.
Further, there is a linear relationship between output Vd at a fixed time
of a cycle of the output (e.g., at the peak of the cycle), the average
absolute value of its positive and/or negative cycles, or the signal
extracted from output Vd using synchronous demodulation (if output Vs of
the voltage source G1 is sinusoidal), and the ratio of the capacitances
c1/c2. Accordingly, there is a linear relationship between output Vd at a
fixed time of a cycle of the output, the average absolute value of its
positive and/or negative cycles, or the signal extracted from the output
using synchronous demodulation, and the capacitance c1 if capacitor C2
has a fixed capacitance, or the reciprocal of the capacitance c2 if
capacitor C1 has a fixed capacitance, which is particularly useful when
measuring distances because the capacitance between two conductive
objects, e.g., two plates, is inversely proportional to the distance
[0045] FIG. 2b depicts circuitry 200b employing capacitance measurement
circuitry 200a1-200an to produce a linear response to changes in each of
the ratios of the capacitance of a capacitor/capacitive transducer to the
capacitance of one or more other different capacitors/capacitive
transducers, while keeping all of the capacitors/capacitive transducers
at substantially identical voltage potentials at all times. Each of a
plurality of capacitors/capacitive transducers C11-C1n having
capacitances of c11-c1n, respectively, is compared to the capacitance c2
of capacitor/capacitive transducer C2 coupled between the non-inverting
input of an operational amplifier A0 and ground (see FIG. 2b). Each of
the capacitance measurement circuitry 200a1-200an operates like the
capacitance measurement circuitry 200a (see FIG. 2a), with the exception
that the voltage potentials at capacitors C11-C1n are compared to the
level of the output of operational amplifier A0, which is configured as a
voltage follower to produce substantially the same voltage potential as
that across capacitor C2. It is noted that capacitor C2 is driven by
output Vs of the voltage source G1 through resistor R2. Thus, there is a
linear relationship between each of outputs Vd1-Vdn at a fixed time of a
cycle of the output (e.g., at the peak of the cycle), the average
extracted from the output using synchronous demodulation (if output Vs of
the voltage source G1 is sinusoidal),. and the ratio of the capacitances
c11/c2 through c1n/c2, respectively. As a result, there is a linear
relationship between each of outputs Vd1-Vdn at a fixed time of a cycle
of the output, the average absolute value of its positive and/or negative
cycles, or the signal extracted from the output using synchronous
demodulation, and the capacitances c11-c1n, respectively, if C2 has a
[0046] FIG. 3a depicts a second illustrative embodiment of capacitance
measurement circuitry 300a, in accordance with the present invention. In
the illustrated embodiment, the capacitance measurement circuitry 300a
includes periodic varying current sources G1 and G2, a first operational
amplifier A1, and a second operational amplifier A2 configured as a
difference amplifier. Each of two capacitors/capacitive transducers C1
and C2 undergoing comparison, having capacitances c1 and c2,
respectively, has one end connected to circuit ground, and another end
connected to one of the differential inputs of operational amplifier A1.
Capacitor C1 is connected to the inverting input of operational amplifier
Al at node 101, and capacitor C2 is connected to the non-inverting input
of operational amplifier A1 at node 102. The node 102 is driven by output
current I2 of the periodic varying current source G2. The node 101 is
driven by output current I1 of the periodic varying current source G1, in
which I1 is equal to K times I2, K being a constant greater than or equal
to zero. Output V1 of operational amplifier A1 is fed back to its
inverting input via feedback resistor R1 having a resistance r1. Because
operational amplifier A1 has a very high open loop gain, the two inputs
of operational amplifier A1 are maintained at substantially the same
voltage potential. Therefore, the magnitude i3 of current I3 flowing
through resistor R1 is substantially equal to the magnitude i2 of current
I2 flowing into capacitor C2 multiplied by the factor (K-c1/c2), i.e.,
i3=i2*(K-c1/c2). Current I3 flowing through resistor R1 will be in-phase
or out-of-phase with currents I1 and I2, depending on whether the ratio
of the capacitances c1/c2 is less than or greater than K. More
specifically, if c1/c2 is less than K, then (K-c1/c2) is positive and
currents I2 and I3 will be in-phase but if c1/c2 is greater than K, then
(K-c1/c2) is negative and currents I2 and I3 will be out of phase. At
steady state, the magnitude i2 of current I2 flowing into capacitor C2 is
only a function of time, and therefore the magnitude i3 of current I3
flowing through resistor R1 at a fixed time of a cycle of the current I3
is an accurate measure of the value (K-c1/c2). The voltage across
resistor R1 is substantially equal to i3*r1, and is equivalent to the
difference of the voltage potential between node 101 (or node 102) and
output V1 of operational amplifier A1. The voltage across resistor R1 can
be measured by connecting node 101 (or node 102) to one of the two inputs
of difference amplifier A2, and by connecting output V1 of operational
amplifier A1 to the other input of difference amplifier A2. It should be
noted that the configuration of difference amplifier A2, as shown in FIG.
3a, is described herein for purposes of illustration, and that other
suitable circuit configurations may be employed. For example, alternative
configurations of difference amplifier A2 may include more than one
operational amplifier. Output Vd of difference amplifier A2 is
proportional to the magnitude i3 of current I3 flowing through resistor
R1, and will be in-phase with currents I1 and I2 when the ratio of the
capacitances c1/c2 is greater than K. It is noted that the phase of
output Vd reverses if the inputs to difference amplifier A2 are
interchanged. Output Vd of difference amplifier A2 is therefore
proportional to a signal representing current I2 modulated by the value
(c1/c2-K). In the event the constant K equals 0, i.e., there is no
current source G1, output Vd of difference amplifier A2 is proportional
to a signal representing current I2 modulated by the value c1/c2. If
capacitances c1/c2 can be measured using a synchronous demodulator for
all values of K. Further, there is a linear relationship between output
Vd at a fixed time of a cycle of the output (e.g., at the peak of the
cycle), the average absolute value of its positive and/or negative
demodulation (if output I2 of the current source G2 is sinusoidal), and
the ratio of the capacitances c1/c2. Thus, there is a linear-relationship
between output Vd at a fixed time of a cycle of the output, the average
extracted from the output using synchronous demodulation, and the
capacitance c1 if capacitor C2 has a fixed capacitance, or the reciprocal
of the capacitance c2 if capacitor C1 has a fixed capacitance, which is
particularly useful when measuring distances because the capacitance
between two conductive objects, e.g., two plates, is inversely
proportional to the distance between them. It should be noted that if the
current sources G1 and G2 have significant dc components in their
outputs, then bypass resistors may be placed across capacitors C1 and C2.
[0047] FIG. 3b depicts circuitry 300b employing capacitance measurement
circuitry 300al-300an to produce a linear response to changes in each of
at substantially identical voltage potentials at all times. Each of the
capacitors/capacitive transducers C11-C1n having capacitances c11-c1n,
respectively, is compared to the capacitance c2 of capacitor/capacitive
transducer C2 coupled between the non-inverting input of operational
amplifier A0 and ground (see FIG. 3b). Capacitors C11-C1n are driven by
current sources G11-G1n, respectively. Each of the respective outputs
I11-I1n of the current sources G11-G1n is equal to I2 times a respective
constant K11-K1n, in which each of the constants K11-K1n is greater than
or equal to zero. Each of the capacitance measurement circuitry
300a1-300an operates like the capacitance measurement circuitry 300a (see
FIG. 3a), with the exception that the voltage potentials at capacitors
C11-C1n are compared to the level of the output of operational amplifier
A0, which is configured as a voltage follower to produce substantially
the same voltage potential as that across capacitor C2, which is driven
by output I2 of current source G2. Therefore, there is a linear
of the output (e.g., at the peak of the cycle), the average absolute
value of its positive and/or negative cycles, or the signal extracted
from the output using synchronous demodulation (if output I2 of the
current source G2 is sinusoidal), and the ratio of the capacitances
cll/c2 through c1n/c2, respectively. As a result, there is a linear
[0048] FIG. 4a depicts a third illustrative embodiment of capacitance
measurement circuitry 400a, in accordance with the present invention. In
the illustrated embodiment, the capacitance measurement circuitry 400a
includes a periodic varying voltage source G1, a first operational
amplifier A0, a second operational amplifier A1, and a third operational
amplifier A2 configured as a difference amplifier. Each of two
capacitors/capacitive transducers C1 and C2 undergoing comparison, having
capacitances c1 and c2, respectively, has one end connected to circuit
amplifier A0 or operational amplifier A1. Capacitor C1 is connected to
the inverting input of operational amplifier A1 at node 101, and
capacitor C2 is connected to the inverting input of operational amplifier
A0 at node 102. A first feedback resistor R1 having resistance r1 is
connected between the output of operational amplifier A1 and its
inverting input. Similarly, a second feedback resistor R2 having
resistance r2 is connected between the output of operational amplifier A0
and its inverting input. The non-inverting inputs of operational
amplifiers A1 and A0 are both driven by output Vs of the periodic varying
voltage source G1, which may be a sinusoidal voltage source. Due to the
high open loop gain of operational amplifiers A1 and A0, capacitors C1
and C2 are maintained at substantially the same voltage potential as
output Vs of the voltage source G1 at all times. V0 is the output of the
operational amplifier A0 and V1 is the output of the operational
amplifier A1. (Vl-Vs) is equal to the time derivative of Vs multiplied by
the value (r1*c1), i.e., (V1-Vs)=r1*c1*dVs/dt, and is in-phase with
current I1 flowing through resistor R1 into capacitor C1. (V0-Vs) is
equal to the time derivative of Vs multiplied by the value (r2*c2), i.e.,
(V0-Vs)=r2*c2*dVs/dt, and is in-phase with current I2 flowing through
resistor R2 into capacitor C2. When output V1 of operational amplifier A1
is provided to the non-inverting input of difference amplifier A2, while
output V0 of operational amplifier A0 is provided to the inverting input
of difference amplifier A2, output Vd of difference amplifier A2 is
proportional to a signal representing the time derivative of Vs modulated
by the value (r1*c1-r2*c2), and is in-phase or out-of-phase with current
flowing through resistors R1 and R2, depending on whether (r1*c1-r2*c2)
is greater than or less than zero (the phase of output Vd reverses if the
inputs to difference amplifier A2 are interchanged). It should be noted
that the configuration of difference amplifier A2, as shown in FIG. 4a,
is described herein for purposes of illustration, and that other suitable
circuit configurations may be employed. For example, alternative
operational amplifier. If voltage Vs is sinusoidal, then the change in
the value of (r1*c1-r2*c2) can be measured using a synchronous
demodulator. At steady state, the time derivative of Vs is only a
function of time, and therefore there is a linear relationship between
output Vd at a fixed time of a cycle of the output (e.g., at the peak of
the cycle), the average absolute value of its positive and/or negative
demodulation (if output Vs of the voltage source G1 is sinusoidal), and
the value (r1*c1-r2*c2). As a result, there is a linear relationship
capacitance c1 if capacitor C2 has a fixed capacitance, or the
capacitance c2 if capacitor C1 has a fixed capacitance.
[0049] FIG. 4b depicts circuitry 400b employing capacitance measurement
circuitry 400a1-400an to produce a linear response to changes in each of
the differences between the capacitance of a capacitor/capacitive
transducer multiplied by a first constant factor, and the capacitance of
one or more other different capacitors/capacitive transducers after each
is multiplied by a respective second constant factor, while keeping all
of the capacitors/capacitive transducers at substantially identical
voltage potentials at all times. Each of capacitors/capacitive
transducers C11-C1n having capacitances of c11-c1n, respectively, is
compared to the capacitance c2 of capacitor/capacitive transducer C2.
Each of the capacitance measurement circuitry 400a1-400an in conjunction
with operational amplifier A0 feedback resistor R2, and capacitor C2,
operates like the capacitance measurement circuitry 400a (see FIG. 4a).
There is therefore a linear relationship between each of outputs Vd1-Vdn
at a fixed time of a cycle of the output (e.g., at the peak of the
its respective value (r1n*c1n-r2*c2), in which "r1n" is the resistance of
the respective feedback resistor R1n associated with the respective
operational amplifier A1n. Accordingly, there is a linear relationship
between each of outputs Vd1-Vdn at a fixed time of a cycle of the output,
the average absolute value of its positive and/or negative cycles, or the
signal extracted from the output using synchronous demodulation, and the
capacitances c11-c1n, respectively, if capacitor C2 has a fixed
[0050] By switching capacitors/capacitive transducers in and out for
subsequent measurement, each embodiment of the presently disclosed
capacitance measurement circuitry can sequentially produce linear
responses to changes in the capacitance or the reciprocal of the
capacitance of a large number of capacitors/capacitive transducers, or
compare the capacitances of a large number of capacitors/capacitive
transducers to the capacitances of a large number of sets of
capacitors/capacitive transducers. It is noted that any suitable type of
capacitive transducer may be employed in each embodiment of the
capacitance measurement circuitry described above, including but not
limited to any suitable type of capacitive transducer for sensing force,
pressure, strain, acceleration, sound, mechanical displacement, fluid
flow, etc. It is further noted that each embodiment of the capacitance
measurement circuitry described above may employ any suitable type of
double-ended power supply, or single-ended power supply, if an
appropriate circuit ground reference can be provided e.g., by a voltage
splitter circuit.
[0051] FIGS. 5a-5d depict illustrative embodiments of a front surface of a
touchless switch, in accordance with the present invention. The front
surface of the presently disclosed touchless switch can take the form of
any suitable type of container such as containers 500a-500c depicted in
FIGS. 5a-5c, respectively. As shown in FIGS. 5a-5c, each of the
containers 500a-500c includes a base portion such as base portions
502a-502c of FIGS. 5a-5c, respectively, and a brim portion such as brim
portions 504a-504c of FIGS. 5a-5c, respectively. Alternatively, the front
surface of the switch can be flat like a front surface 500d (see FIG.
5d), or any other suitable surface configuration. The presently disclosed
touchless switch includes two sensor electrodes, specifically, a center
electrode and an outer electrode, which are disposed on or behind the
front surface of the switch and are maintained at substantially the same
voltage potential. The center electrode is spaced from and at least
partly surrounded by the outer electrode. The center and outer electrodes
can be of any suitable shape, form, or size, and need not be a solid
piece, e.g., an electrode may be a wire mesh. FIGS. 6a-6d depict
illustrative arrangements and positions of the center and outer
electrodes relative to each other, and relative to the front surfaces
500a-500d of FIGS. 5a-5d, respectively. As shown in FIGS. 6a-6c, when the
front surface is in the form of a container, the center electrode is
disposed near the base of the container, and the outer electrode is
disposed near the brim of the container.
[0052] The presence of the tip of a human finger or a finger-like object
near or proximate to the center electrode of the touchless switch can be
detected using an embodiment of the presently disclosed capacitance
measurement circuitry. When detecting the presence of the human finger or
finger-like object, the capacitance measurement circuitry compares the
capacitances of the capacitors formed between the two sensor electrodes
(i.e., the center electrode and the outer electrode) and the human finger
or finger-like object, thereby substantially canceling out extraneous
common-mode disturbances, for example, the capacitance between the rest
of the human body and the sensor electrodes, the capacitance between the
human body and ground, environmental changes, electrical noise, etc. Such
extraneous common-mode disturbances tend to affect both sensor electrodes
equally due to their close proximity to one another. Additionally, the
outer electrode can be positioned in the touchless switch so that the tip
of the human finger or finger-like object is required to go past a
specified boundary before actuating the switch. In one embodiment, this
is accomplished by positioning the outer electrode a specified distance
in front of the center electrode, and configuring the spacing between the
two sensor electrodes and their relative surface areas so that when an
object is near the electrodes, but more than a specified distance away
from the center electrode, the ratio of the capacitance between the
object and the center electrode to the capacitance between the object and
the outer electrode, or the difference between the capacitance between
the object and the center electrode multiplied by a first constant and
the capacitance between the object and the outer electrode multiplied by
a second constant, is less than a preset threshold. It is understood that
the capacitance ratio and difference measurements are performed by an
embodiment of the above-described capacitance measurement circuitry, and
are facilitated by the fixed geometrical shape, size, and relative
position of the two sensor electrodes. In one embodiment, the touchless
switch is actuated when the measured capacitance ratio or difference
exceeds the preset threshold.
[0053] Accordingly, the touchless switch is not actuated by a human finger
or finger-like object until the finger passes through the specified
boundary. In the event the front surface of the touchless switch has the
form of the surface of a container (see, e.g., FIGS. 5a-5c), the
specified boundary coincides with an imaginary plane defined by the brim
of the container. As the tip of a human finger or finger-like object
moves toward the center electrode and breaks the plane of the specified
boundary, the capacitance associated with the center electrode increases
more rapidly than the capacitance associated with the outer electrode.
The touchless switch is actuated when the ratio of the capacitances
associated with the center electrode and the outer electrode, or the
difference between the capacitance associated with the center electrode
multiplied by a first constant and the capacitance associated with the
outer electrode multiplied by a second constant, exceeds the preset
[0054] FIG. 7 depicts an illustrative embodiment of a touchless switch
700, in accordance with the present invention. In the illustrated
embodiment, the touchless switch 700 includes a front surface 702 in the
form of a container, a center electrode 704, an outer electrode 706, and
a guard electrode 708 surrounding the back and the sides of the two
sensor electrodes 704, 706. All of the electrodes 704, 706, 708 are
maintained at substantially the same voltage potential. As a result, the
two sensor electrodes 704, 706 are operative to form electric fields only
between the sensor electrodes and objects disposed in front of the
switch, i.e., above the switch 700, as depicted in FIG. 7. Leads
circuitry may also be guarded using a twin-axial cable or two coaxial
inside cable conductors connected to the sensor electrodes.
[0055] FIG. 8a depicts a first illustrative circuit implementation 800a of
a touchless switch, in accordance with the present invention. As shown in
FIG. 8a, the circuit implementation 800a comprises a center electrode E1,
an outer electrode E2, a guard electrode E3, a startup delay section 203,
a switching decision section 205, a switching output section 207, and
capacitance measurement circuitry 802a, which includes a periodic varying
voltage source G1, operational amplifiers A0 and A1, resistors R1-R3, and
an operational amplifier A2 configured as a difference amplifier. The
center electrode El is connected to the inverting input of operational
amplifier A1 at node 201, and the outer electrode E2 is connected at node
202 to the non-inverting input of operational amplifier A0, which is
configured as a voltage follower to provide the voltage potential of
outer electrode E2 to the non-inverting input of operational amplifier
A1. The nodes 201 and 202 are both driven by an output Vs of the periodic
varying voltage source G1. Electrodes E1 and E2 correspond to capacitors
C1 and C2 of FIG. 2a, respectively. Guard electrode E3 is connected to
the output of operational amplifier A0, and therefore the voltage
potential of guard electrode E3 is substantially equal to the voltage
potentials of sensor electrodes E1 and E2. Guard electrode E3 may be
configured to surround the back and the sides of sensor electrodes E1 and
E2 so that capacitances can only be formed between conductive objects
disposed in front of the touchless switch and sensor electrodes E1 and
E2. It is noted that operational amplifiers A1 and A2 of FIG. 8a are like
operational amplifiers A1 and A2 of FIG. 2a, respectively, resistor R2 of
FIG. 8a is like resistor R2 of FIG. 2a, resistors R1 and R3 of FIG. 8a
are like resistors R1 and R3 of FIG. 2a, respectively, and the periodic
varying voltage source G1 of FIG. 8a is like the periodic varying voltage
source G1 of FIG. 2a. Thus, there is a linear relationship between output
Vd of difference amplifier A2 at a fixed time of a cycle of the output
(e.g., at the peak of the cycle), or the average absolute value of its
using synchronous demodulation (if output Vs of voltage source G1 is
sinusoidal), and the ratio of the capacitance associated with center
electrode E1 to the capacitance associated with outer electrode E2.
Difference amplifier A2 provides output Vd to the switching decision
section 205, which determines whether to actuate the touchless switch
based on signal Vd. For example, the switching decision section 205 can
base its decision on the phase, the amplitude, an average, and/or any
other suitable property of signal Vd. Alternatively, the switching
decision section 205 can require a specified number of consecutive
detections of the required phase and/or amplitude of signal Vd, or the
satisfaction of certain criteria, before deciding to actuate the
touchless switch. If voltage Vs is sinusoidal, then a synchronous
demodulator can be included in the switching decision section 205 so that
the change in the ratio of the capacitance associated with center
electrode E1 to the capacitance associated with outer electrode E2 can be
obtained with a high degree of accuracy, even at a high noise level. It
should be noted that the switching decision section 205 may require one
or more signals in addition to signal Vd to determine whether or not to
actuate the switch. For example, the switching decision section 205 may
require a reference signal to determine the phase of signal Vd. The
switching decision section 205 provides a logic signal 206 representing
its decision to the switching output section 207, which implements the
required switching action. It is noted that the switching output section
207 may be implemented using any suitable number of logical outputs
(normally high or low), solid state switch outputs, and/or dry contact
outputs (normally open or closed) in any suitable switching mode,
including but not limited to pulse mode, momentarily mode, toggle mode,
etc. The switching output section 207 can also be configured to produce
audio and/or visual outputs to indicate the status of the touchless
switch. Because the capacitance measurement circuitry 802a takes several
cycles of output Vs of the voltage source G1 to stabilize, the startup
delay section 203 outputs a startup signal 204 to the switching decision
section 205 during the startup period to prevent the switching decision
section 205 from inadvertently actuating the switch. When sensor
electrodes E1 and E2 are disposed at a distance away from the inputs of
operational amplifiers A0 and A1, the leads from center electrode E1 and
outer electrode E2 may be guarded using a twin-axial cable or two coaxial
cables of equal length, using the outer conductors as the guard shields
connected to guard electrode E3 and maintained at substantially the same
voltage potential as the inside conductors connected to respective sensor
electrodes E1 and E2, so that no stray capacitance is introduced and any
other unwanted effects introduced by the leads are substantially
[0056] FIG. 8b depicts a first illustrative circuit implementation 800b of
a set of touchless switches, including a periodic varying voltage source
G1, an operational amplifier A0, capacitance measurement circuitry
802a1-802an, the startup delay section 203, the switching decision
section 205, and the switching output section 207. It is noted that each
of the capacitance measurement circuitry 802a1-802an in conjunction with
operational amplifier A0 is like the capacitance measurement circuitry
802a (see FIG. 8a), and corresponds to a respective touchless switch in
the set of touchless switches. Specifically, electrode E2 coupled to the
non-inverting input of operational amplifier A0 corresponds to a common
outer electrode of the set of touchless switches, and electrode E3
coupled to the output of operational amplifier A0 corresponds to a common
guard electrode of the set of touchless switches. Each of electrodes
E11-E1n corresponds to a center electrode of a respective touchless
switch. It is noted that operational amplifier A0 of FIG. 8b is like
operational amplifier A0 of FIG. 2b, operational amplifiers A11-A1n of
FIG. 8b are like operational amplifiers A11-A1n of FIG. 2b, respectively,
difference amplifiers A2l-A2n of FIG. 8b are like difference amplifiers
A2l-A2n of FIG. 2b, respectively, resistor R2 of FIG. 8b is like resistor
R2 of FIG. 2b, resistors R1l-R1n of FIG. 8b are like resistors R1l-R1n of
FIG. 2b, respectively, resistors R3l-R3n of FIG. 8b are like resistors
R3l-R3n of FIG. 2b, respectively, and the periodic varying voltage source
G1 of FIG. 8b is like the periodic varying voltage source G1 of FIG. 2b.
Difference amplifiers A2l-A2n provide output signals Vd1-Vdn,
respectively, to the switching decision section 205, which determines
when to actuate each switch based on the respective signals Vd1-Vdn. The
switching decision section 205 provides logic signals 206 representing
its respective decisions to the switching output section 207, which
implements the required switching action for each switch. It is noted
that the switching output section 207 may be implemented using any
suitable number of logical outputs (normally high or low), solid state
switch outputs, and/or dry contact outputs (normally open or closed) in
any suitable switching mode, including but not limited to pulse mode,
momentarily mode, toggle mode, etc., for each switch. The switching
output section 207 can also be configured to produce audio and/or visual
outputs to indicate the status of each switch. The startup delay section
203 of FIG. 8b is like the corresponding section 203 described above with
reference to FIG. 8a, and each switch of circuit implementation 800b (see
FIG. 8b) basically operates like the switch of circuit implementation
800a (see FIG. 8a).
[0057] FIG. 9a depicts a second illustrative circuit implementation 900a
of a touchless switch, in accordance with the present invention. As shown
in FIG. 9a, the circuit implementation 900a comprises a center electrode
E1, an outer electrode E2, a guard electrode E3, a startup delay section
203, a switching decision section 205, a switching output section 207,
and capacitance measurement circuitry 902a, which includes periodic
varying current sources G1 and G2, operational amplifiers A0 and A1,
resistor R1, and an operational amplifier A2 configured as a difference
amplifier. The center electrode E1 is connected to the inverting input of
operational amplifier A1 at node 201, and outer electrode E2 is connected
at node 202 to the non-inverting input of operational amplifier A0, which
is configured as a voltage follower to provide the voltage potential of
A1. The node 201 is driven by an output current I1 of the periodic
varying current source G1 and node 202 is driven by an output current I2
of the periodic varying current source G2. Electrodes E1 and E2
correspond to capacitors C1 and C2 of FIG. 3a, respectively. Guard
electrode E3 is connected to the output of operational amplifier A0, and
therefore the voltage potential of guard electrode E3 is substantially
the same as the voltage potentials of sensor electrodes E1 and E2. Guard
electrode E3 may be configured to surround the back and the sides of
sensor electrodes E1 and E2 so that capacitances can be formed only
between conductive objects disposed in front of the touchless switch and
the sensor electrodes E1 and E2. It is noted that operational amplifiers
A1 and A2 of FIG. 9a are like operational amplifiers Al and A2 of FIG.
3a, respectively, resistor R1 of FIG. 9a is like resistor R1 of FIG. 3a,
and the periodic varying current sources G1 and G2 of FIG. 9a are like
the periodic varying current sources G1 and G2 of FIG. 3a, respectively.
Thus, there is a linear relationship between output Vd of difference
amplifier A2 at a fixed time of a cycle of the output (e.g., at the peak
of the output), or the average absolute value of its positive and/or
negative cycles, or the signal extracted from the output using
synchronous demodulation (if output I2 of current source G2 is
Difference amplifier A2 provides output signal Vd to the switching
decision section 205, which determines whether to actuate the touchless
switch based on signal Vd. For example, the switching decision section
205 can base its decision on the phase, the amplitude, an average, and/or
any other suitable property of signal Vd. Alternatively, the switching
satisfaction of certain criteria, before deciding to actuate the switch.
If current I2 is sinusoidal, then a synchronous demodulator can be
included in the switching decision section 205 so that the change in the
ratio of the capacitance associated with center electrode E1 to the
capacitance associated with outer electrode E2 can be obtained with a
high degree of accuracy, even at a high noise level. It should be noted
that the switching decision section 205 may require one or more signals
in addition to signal Vd to determine whether or not to actuate the
switch. For example, the switching decision section 205 may require a
reference signal to determine the phase of signal Vd. The switching
decision section 205 provides a logic signal 206 representing its
decision to the switching output section 207, which implements the
outputs (normally open or closed) in any switching mode, including but
not limited to pulse mode, momentarily mode, toggle mode, etc. The
switching output section 207 can also be configured to produce audio
and/or visual outputs to indicate the status of the switch. Because the
capacitance measuring circuitry 902a takes several cycles of the output
I2 of the periodic varying current source G2 to stabilize, a startup
delay section 203 provides a startup signal 204 to the switching decision
section 205 during the startup period to prevent it from inadvertently
actuating the switch. When sensor electrodes E1 and E2 are disposed at a
distance away from the inputs of operational amplifiers A0 and A1, the
leads from sensor electrodes E1 and E2 may be guarded using a twin-axial
cable or two coaxial cables of equal length, with the outer conductors
employed as guard shields connected to guard electrode E3 and maintained
at substantially the same voltage potential as the inside conductors
connected to respective sensor electrodes E1 and E2, so that no stray
capacitance is introduced and any other unwanted effects introduced by
the leads are substantially cancelled out.
[0058] FIG. 9b depicts a second illustrative circuit implementation 900b
of a set of touchless switches, including periodic varying current
sources G11-G1n, a periodic varying current source G2, an operational
amplifier A0, capacitance measurement circuitry 902a1-902an, the startup
delay section 203, the switching decision section 205, and the switching
output section 207. It is noted that each of the capacitance measurement
circuitry 902a1-902an, in conjunction with operational amplifier A0, is
like the capacitance measurement circuitry 902a (see FIG. 9a), and
corresponds to a respective touchless switch in the set of touchless
switches. Specifically, electrode E2 coupled to the non-inverting input
of operational amplifier A0 corresponds to a common outer electrode of
the set of touchless switches, and electrode E3 coupled to the output of
operational amplifier A0 corresponds to a common guard electrode of the
set of touchless switches. Each of electrodes E11-E1n corresponds to a
center electrode of a respective touchless switch. Further, operational
amplifier A0 of FIG. 9b is like operational amplifier A0 of FIG. 3b,
operational amplifiers A11-A1n of FIG. 9b are like operational amplifiers
A11-A1n of FIG. 3b, respectively, difference amplifiers A21-A2n of FIG.
9b are like difference amplifiers A21-A2n of FIG. 3b, respectively,
resistors R11-R1n of FIG. 9b are like resistors R11-R1n of FIG. 3b,
respectively, the periodic varying current source G2 of FIG. 9b is like
the periodic varying current source G2 of FIG. 3b, and the periodic
varying current sources G11-G1n of FIG. 9b are like the periodic varying
current sources G11-G1n of FIG. 3b. Difference amplifiers A21-A2n provide
output signals Vd1-Vdn, respectively, to the switching decision section
205, which determines when to actuate each switch based on the respective
signals Vd1-Vdn. The switching decision section 205 provides logic
signals 206 representing its respective decisions to the switching output
section 207, which implements the required switching action for each
switch. It is noted that the switching output section 207 may be
implemented using any suitable number of logical outputs (normally high
or low), solid state switch outputs, and/or dry contact outputs (normally
open or closed) in any suitable switching mode, including but not limited
to pulse mode, momentarily mode, toggle mode, etc., for each switch. The
and/or visual outputs to indicate the status of each switch. The startup
delay section 203 of FIG. 9b is like the startup delay section 203 of
FIG. 9a, and each switch of circuit implementation 900b (see FIG. 9b)
basically operates like the switch of circuit implementation 900a (see
[0059] FIG. 10a depicts a third illustrative circuit implementation 1000a
in FIG. 10a, the circuit implementation 1000a comprises a center
electrode E1, an outer electrode E2, a guard electrode E3, a startup
delay section 203, a switching decision section 205, a switching output
section 207, and capacitance measurement circuitry 1002a, which includes
a periodic varying voltage source G1, operational amplifiers A0 and A1,
resistors R1 and R2, and an operational amplifier A2 configured as a
difference amplifier. The non-inverting inputs of operational amplifier
A0 and A1 are both driven by an output Vs of the periodic varying voltage
source G1. The center electrode E1 is connected to the inverting input of
at node 202 to the inverting input of operational amplifier A0. It is
noted that sensor electrodes E1 and E2 correspond to capacitors C1 and C2
of FIG. 4a, respectively. Guard electrode E3 is connected to the output
of the periodic varying voltage source G1, and is therefore substantially
the same as the voltage potential of the two sensor electrodes E1 and E2.
Guard electrode E3 may be configured to surround the back and the sides
of the sensor electrodes E1 and E2 so that capacitances can be formed
only between conductive objects disposed in front of the touchless switch
and the sensor electrodes E1 and E2. It is noted that operational
amplifiers A0 and A1 of FIG. 10a are like operational amplifiers A0 and
A1 of FIG. 4a, respectively, resistor R1 of FIG. 10a is like resistor R1
of FIG. 4a, resistor R2 of FIG. 10a is like resistor R2 of FIG. 4a, and
the periodic varying voltage source G1 of FIG. 10a is like the periodic
varying voltage source G1 of FIG. 4a. Thus, there is a linear
relationship between output Vd of difference amplifier A2 at a fixed time
of a cycle of the output (e.g., at the peak of the cycle), or the average
the voltage source G1 is sinusoidal) and the value (r1*c1-r2*c2), in
which r1 and r2 are the respective resistances of resistors R1 and R2 and
c1 and c2 are the respective capacitances associated with sensor
electrodes E1 and E2. The difference amplifier A2 provides output signal
Vd to the switching decision section 205, which determines whether to
actuate the switch based on signal Vd. For example, the switching
decision section 205 can base its decision on the phase, the amplitude,
an average, and/or any other suitable property of signal Vd.
Alternatively, the switching decision section 205 can require a specified
number of consecutive detections of the required phase and/or amplitude
of signal Vd, or the satisfaction of certain criteria, before deciding to
actuate the switch. If voltage Vs is sinusoidal, then a synchronous
the change in the value of (r1*c1-r2*c2) can be obtained with a high
degree of accuracy, even at a high noise level. It should be noted that
the switching decision section 205 may require one or more signals in
addition to signal Vd to determine whether or not to actuate the switch.
For example, the switching decision section 205 may require a reference
signal to determine the phase of output signal Vd. The switching decision
section 205 provides a logic signal 206 representing its decision to the
switching output section 207, which implements the required switching
action. It is noted that the switching output section 207 may be
open or closed) in any switching mode, including but not limited to pulse
mode, momentarily mode, toggle mode, etc. The switching output section
207 can also be configured to produce audio and/or visual outputs to
indicate the status of the switch. Because the capacitance measuring
circuit 1002a takes several cycles of output Vs of the periodic varying
voltage source G1 to stabilize, a startup delay section 203 provides a
startup signal 204 to the switching decision section 205 during the
startup period to prevent it from inadvertently actuating the switch.
When sensor electrodes E1 and E2 are disposed at a distance away from the
inputs of operational amplifiers A0 and A1, the leads from sensor
electrodes E1 and E2 may be guarded using a twin-axial cable or two
coaxial cables of equal length, with the outer conductors employed as
guard shields connected to the guard electrode E3 and maintained at
substantially the same voltage potential as the inside conductors
[0060] FIG. 10b depicts a third illustrative circuit implementation 1000b
of a set of touchless switches, including a periodic varying voltage
source G1, an operational amplifier A0, capacitance measurement circuitry
1002a1-1002an, the startup delay section 203, the switching decision
of the capacitance measurement circuitry 1002a1-1002an, in conjunction
with operational amplifier A0, is like the capacitance measurement
circuitry 1002a (see FIG. 10a), and corresponds to a respective touchless
switch in the set of touchless switches. Specifically, electrode E2
coupled to the inverting input of operational amplifier A0 corresponds to
a common outer electrode of the set of touchless switches, and electrode
E3 coupled to the output Vs of voltage source G1 corresponds to a common
switch. Further, operational amplifier A0 of FIG. 10b is like operational
amplifier A0 of FIG. 4b, operational amplifiers A11-A1n of FIG. 10b are
like operational amplifiers A11-A1n of FIG. 4b, respectively, difference
amplifiers A21-A2n of FIG. 10b are like difference amplifiers A21-A2n of
FIG. 4b, respectively, resistor R2 of FIG. 10b is like resistor R2 of
FIG. 4b, resistors R11-R1n of FIG. 10b are like resistors R11-R1n of FIG.
4b, respectively, and the periodic varying voltage source G1 of FIG. 10b
is like the periodic varying voltage source G1 of FIG. 4b. The difference
amplifiers A21-A2n provide output signals Vd1-Vdn, respectively, to the
switching decision section 205, which determines when to actuate each
switch based on the respective signals Vd1-Vdn. The switching decision
section 205 provides logic signals 206 representing its respective
decisions to the switching output section 207, which implements the
required switching action for each switch. It is noted that the switching
output section 207 may be implemented using any suitable number of
logical outputs (normally high or low), solid state switch outputs,
and/or dry contact outputs (normally open or closed) in any suitable
switching mode, including but not limited to pulse mode, momentarily
mode, toggle mode, etc., for each switch. The switching output section
indicate the status of each switch. The startup delay section 203 of FIG.
10b is like the startup delay section 203 of FIG. 10a, and each switch of
circuit implementation 1000b (see FIG. 10b) operates like the switch of
circuit implementation 1000a (see FIG. 10a).
[0061] Having described the above illustrative embodiments, other
alternative embodiments or variations may be made. For example, each of
the presently disclosed circuit implementations of touchless switches can
be scaled up to detect the proximity of a larger human appendage or other
conductive object, e.g., the palm of a human hand. The above illustrative
embodiments can also be adapted to detect the position or movement of a
human appendage or conductive object by measuring the capacitances
between the object and each of an array of sensor electrodes, using one
of the capacitance measuring techniques described above and analyzing the
results using suitable electronic circuitry or a suitably programmed
[0062] It is noted that the outer electrode of the touchless switch may be
positioned in front of, behind, or at any other suitable position
relative to the center electrode or set of center electrodes, depending
upon the specific application. Also, as discussed above, the electrodes
piece. For example, in an application for a proximity sensor in which the
sensor is used in an outdoor environment, the proximity sensor can have
an outer electrode in the form of an insulated conductive mesh disposed
on the outer surface of the proximity sensor, and a center electrode
placed behind the outer electrode on the inner side of the surface of the
proximity sensor. In such an application, the presence of a human
appendage or other conductive object near or proximate to the proximity
sensor in front of the center electrode causes the capacitance associated
with the center electrode to increase more than the capacitance
associated with the outer electrode. The condition in which the relative
changes in the capacitance associated with the center electrode and the
capacitance associated with the outer electrode exceed a preset threshold
can then be detected using one of the above-described embodiments of the
capacitance measurement circuitry. When water or moisture is deposited on
the outer surface of the proximity sensor, it is essentially deposited on
the insulated conductive mesh of the outer electrode. As a result, the
water or moisture has a greater effect on the capacitance associated with
the outer electrode than the capacitance associated with the center
electrode for the same amount of surface area due to fact that it is much
closer to the outer electrode, being separated by just the thickness of
the insulation of the outer electrode. Since the outer electrode is in
the form of a mesh, the surface area of the outer electrode in contact
with the water or moisture is usually much smaller than the area
projected onto the surface of the center electrode by the water or
moisture. By properly designing the relative sizes of the surface areas
and the spacing between the two electrodes of the proximity sensor, the
relative changes in the capacitance associated with the two electrodes
due to the presence of water or moisture on the surface of the proximity
sensor can be made never to exceed the preset threshold, thereby enabling
the proximity sensor to operate outdoors under inclement weather
conditions. It is possible to have the insulated conductive mesh of the
outer electrode reside in grooved or recessed areas of the surface of the
proximity sensor so that the insulated conductive mesh can come in
contact with water or moisture deposited on the surface of the proximity
sensor, but not a human appendage or other conductive object near the
proximity sensor. In this way, the relative changes in the capacitance
associated with the two electrodes due to a human appendage or conductive
object touching the surface of the proximity sensor can be made to exceed
the preset threshold, while the mere presence of water or moisture on the
surface of the proximity sensor does not cause the relative changes in
capacitance to exceed the preset threshold.
[0063] In addition, while the present invention may be embodied using
hardware components, it is appreciated that one or more functions
necessary to implement the invention may alternatively be embodied in
whole or in part using hardware or software or some combination thereof
using micro-controllers, microprocessors, digital signal processors,
programmable logic arrays, or any other suitable hardware and/or
[0064] It will be appreciated by those of ordinary skill in the art that
further modifications to and variations of the above-described linear
capacitance measurement and touchless switch may be made without
departing from the inventive concepts disclosed herein. Accordingly, the
invention should not be viewed as limited except as by the scope and
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Patent InfoApplication # US 20070164756 A1Publish Date 07/19/2007 Document # 11654495 File Date 01/17/2007 USPTO Class 324662000 Other USPTO Classes International Class 01R27/26 Drawings 16 Negative Feedback