Source: https://patents.google.com/patent/US8838033
Timestamp: 2018-07-17 02:22:35
Document Index: 448261943

Matched Legal Cases: ['Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60', 'Application No. 60']

US8838033B2 - System and method for signal amplification - Google Patents
System and method for signal amplification Download PDF
US8838033B2
US8838033B2 US11005837 US583704A US8838033B2 US 8838033 B2 US8838033 B2 US 8838033B2 US 11005837 US11005837 US 11005837 US 583704 A US583704 A US 583704A US 8838033 B2 US8838033 B2 US 8838033B2
US11005837
US20050100084A1 (en )
Methods and systems for processing a signal with a corresponding noise profile are disclosed. Aspects of the method may comprise analyzing spectral content of the noise profile. At least one noise harmonic within the signal may be filtered based on said analyzed spectral content. The filtered signal may be amplified. The noise profile may comprise a phase noise profile. The signal may comprise at least one of a sinusoidal signal and a noise signal. At least one filter coefficient that is used to filter the at least one noise harmonic may be determined. The filtering may comprise low pass filtering. The signal may be modulated prior to filtering. The amplifying may comprise buffering. A non-linearity characteristic of the signal may be determined and a noise harmonic may be low-pass filtered within the signal based on the determined non-linearity characteristic.
The present application is a continuation-in-part of U.S. patent application Ser. No. 09/634,552, filed Aug. 8, 2000 which claims benefit from and priority to U.S. Patent Application Ser. No. 60/160,806, filed Oct. 21, 1999; Application No. 60/163,487, filed Nov. 4, 1999; Application No. 60/163,398, filed Nov. 4, 1999; Application No. 60/164,442, filed Nov. 9, 1999; Application No. 60/164,194, filed Nov. 9, 1999; Application No. 60/164,314, filed Nov. 9, 1999; Application No. 60/165,234, filed Nov. 11, 1999; Application No. 60/165,239, filed Nov. 11, 1999; Application No. 60/165,356; filed Nov. 12, 1999; Application No. 60/165,355, filed Nov. 12, 1999; Application No. 60/172,348, filed Dec. 16, 1999; Application No. 60/201,335, filed May 2, 2000; Application No. 60/201,157, filed May 2, 2000; Application No. 60/201,179, filed May 2, 2000; Application No. 60/202,997, filed May 10, 2000; Application No. 60/201,330, filed May 2, 2000. The above referenced applications are hereby incorporated herein by reference in their entireties.
The present application is also a continuation-in-part of U.S. patent application Ser. No. 10/409,213, filed Apr. 3, 2003 and entitled “Phase Locked Loop That Avoids False Locking,” and U.S. patent application Ser. No. 10/957,043, filed Oct. 1, 2004 and entitled “System And Method For Signal Limiting,” the complete subject matters of which are hereby incorporated herein by reference in their entireties.
U.S. patent application Ser. No. 10/409,213, filed Apr. 3, 2003;
U.S. patent application Ser. No. 09/634,552, filed Aug. 8, 2000;
U.S. patent application Ser. No. 10/813,486, filed Mar. 30, 2004; and
U.S. patent application Ser. No. 10/957,043, filed Oct. 1, 2004.
Spectral purity and reduced phase noise are becoming an inseparable requirement of signal generation and amplification circuits. Most modern communication systems, in particular, employ amplifiers that may be implemented as buffers and/or low-noise amplifiers (LNAs), for example, and are characterized with corresponding performance specifications. Those specifications normally dictate the performance of individual blocks, including voltage controlled oscillators (VCOs), dividers, etc. Traditionally, the noise and spectral profile of different blocks are included in a linear, phase domain AC-type analysis, or simulation, to estimate final spectral performance. Such analysis, however, ignores the nonlinear effects in the signal generation path, including an amplifying action by an amplifier, for example.
During operation of a conventional VCO, the VCO output is buffered before it is applied to the next stage. The buffer can be implemented as a power amplifier designed to deliver the signal to an off-chip load, or it may also be implemented as a simple tuned stage that sits between the VCO and a divider, for example. Because of the non-linear effect in the signal amplifier/buffer within an electric circuit containing a conventional VCO, for example, and the resulting phase noise profile, as outlined below, an amplifying action by an amplifier may substantially increase the phase noise profile of the generated signal at the output of the amplifier.
Aspects of the present invention may be found in a method and system for processing a signal with a corresponding noise profile. Aspects of the method may comprise analyzing spectral content of the noise profile. At least one noise harmonic within the signal may be filtered based on said analyzed spectral content. The filtered signal may be amplified. The noise profile may comprise a phase noise profile. The signal may comprise at least one of a sinusoidal signal and a noise signal. At least one filter coefficient that is used to filter the at least one noise harmonic may be determined.
The filtering may comprise low pass filtering. The signal may be modulated prior to filtering. The amplifying may comprise buffering. A non-linearity characteristic of the signal may be determined and a noise harmonic may be low-pass filtered within the signal based on the determined non-linearity characteristic. The non-linearity characteristic may comprise a noise harmonic frequency and/or a noise harmonic amplitude. The spectral content may comprise an input noise spectrum and/or an output noise spectrum.
Aspects of the system may comprise a processor that analyzes spectral content of the noise profile. A filter may filter at least one noise harmonic within the signal based on the analyzed spectral content. An amplifier may amplify the filtered signal. The noise profile may comprise a phase noise profile and the signal may comprise a sinusoidal signal and/or a noise signal. The processor may determine a filter coefficient that is used to filter the noise harmonic. The filter may comprise a low-pass filter. The system may further comprising a modulator that modulates the signal prior to the filtering.
The amplifier may buffer the filtered signal. The processor may determine a non-linearity characteristics of the signal and the filter may low-pass filters the noise harmonic within the signal based on the determined non-linearity characteristic. The non-linearity characteristic may comprise a noise harmonic frequency and/or a noise harmonic amplitude. The spectral content may comprise an input noise spectrum and/or an output noise spectrum.
FIG. 21 is a schematic block diagram of a CMOS tuned amplifier that may be utilized in accordance with an embodiment of the invention.
FIG. 22 is a graphical representation of an exemplary amplifier input spectrum, in accordance with an embodiment of the invention.
FIG. 23 is a graphical representation of an exemplary amplifier output spectrum including output current, in accordance with an embodiment of the invention.
FIG. 24 is a schematic block diagram of a CMOS tuned amplifier utilizing noise spectrum analysis, in accordance with an embodiment of the invention.
FIG. 25 is a flow diagram of an exemplary method for processing a sinusoidal wave signal with a phase noise profile, in accordance with an embodiment of the invention.
An amplifier, or a buffer, may comprise an amplifying active device with a certain level of nonlinearity, followed by a tuned stage, for example. In one aspect of the invention, a general technique for analyzing a mildly nonlinear buffer/amplifier may be developed through solving a typical implementation with long channel CMOS devices. In a different aspect of the invention, a resulting technique for reducing phase noise prior to amplification may be implemented. For example, a processor may be utilized prior to amplification by the amplifier to analyze one or more noise characteristics of an incoming signal. A filter, such as a low-pass filter, may then be utilized to filter one or more noise characteristics, such as a noise harmonic signal, from the analyzed signal prior to amplification.
Nonlinear operations within an electric circuit, such as amplifying, may cause distortion and aliasing in the signal and noise spectrum. In particular, it may be established for a hard-limiter, for example, that a limiting action by the hard-limiter may cause infinite folding and generation of harmonics at the output of the signal limiter. Similarly, it may be established that an amplifying action by an amplifier may also cause distortion and aliasing in the signal and noise spectrum.
x ⁡ ( t ) ≈ A 0 ⁢ ⁢ cos ⁢ ⁢ 2 ⁢ ⁢ π ⁢ ⁢ f c ⁢ t + A 0 ⁢ m 2 ⁡ [ cos ⁢ ⁢ 2 ⁢ ⁢ π ⁡ ( f c + f m ) ⁢ t - cos ⁢ ⁢ 2 ⁢ ⁢ π ⁡ ( f c - f m ) ⁢ t ] ( 3 )
x ⁡ ( t ) = A 0 ⁢ ⁢ cos ⁢ ⁢ 2 ⁢ ⁢ π ⁢ ⁢ f c ⁢ t + ( a 0 2 + A 0 ⁢ m 2 ) ⁢ cos ⁢ ⁢ 2 ⁢ ⁢ π ⁡ ( f c + f m ) ⁢ t + ( a 0 2 - A 0 ⁢ m 2 ) ⁢ cos ⁢ ⁢ 2 ⁢ ⁢ π ⁡ ( f c - f m ) ⁢ t ( 6 )
x ⁡ ( t ) = A 0 ⁢ ⁢ cos ⁢ ⁢ 2 ⁢ ⁢ π ⁢ ⁢ f c ⁢ t + ( A 2 2 + A 0 ⁢ A 2 2 ⁢ A 0 ) ⁢ cos ⁢ [ 2 ⁢ ⁢ π ( f c + ( f 2 - f c ) ] ⁢ t + ( A 2 2 - A 0 ⁢ A 2 2 ⁢ A 0 ) ⁢ cos ⁢ [ 2 ⁢ ⁢ π ( f c - ( f 2 - f c ) ] ⁢ t ( 8 )
x ⁡ ( t ) = ( A 0 + a 0 ⁢ ⁢ cos ⁢ ⁢ 2 ⁢ ⁢ π ⁢ ⁢ f m ⁢ t ) · cos ⁡ ( 2 ⁢ π ⁢ ⁢ f c ⁢ t + m ⁢ ⁢ sin ⁢ ⁢ 2 ⁢ π ⁢ ⁢ f m ⁢ t ) , ⁢ ⁢ a 0 = A 2 , ⁢ ⁢ m = A 2 A 0 , ⁢ ⁢ f m = f 2 - f c ( 9 )
y ⁡ ( t ) = A 0 ⁢ ⁢ cos ⁢ ⁢ 2 ⁢ ⁢ π ⁢ ⁢ f c ⁢ t + A 2 2 ⁢ cos ⁢ ⁢ 2 ⁢ ⁢ π ⁡ ( f c + f m ) ⁢ t - A 2 2 ⁢ cos ⁢ ⁢ 2 ⁢ ⁢ π ⁡ ( f c - f m ) ⁢ t ( 10 )
V in(t)=A sin(2πf 1 t)+A sin(2πf 2 t+θ) (11)
{ 2 ⁢ π ⁢ ⁢ f 1 ⁢ t = - 2 ⁢ ⁢ π ⁢ ⁢ f 2 ⁢ t - θ + 2 ⁢ k ⁢ ⁢ π 2 ⁢ ⁢ π ⁢ ⁢ f 1 ⁢ t = π + 2 ⁢ ⁢ π ⁢ ⁢ f 2 ⁢ t + θ + 2 ⁢ k ⁢ ⁢ π ⁢ → { 2 ⁢ π ⁢ ⁢ ( f 1 + f 2 ) ⁢ t = 2 ⁢ k ⁢ ⁢ π ⁢ - θ ⁢ 2 ⁢ ⁢ π ⁢ ⁢ ( f 1 - f 2 ) ⁢ t = 2 ⁢ k ⁢ ⁢ π + π + θ ( 13 )
A 1 ⁢ ⁢ sin ⁡ ( 2 ⁢ π ⁢ ⁢ f 1 ⁢ Δ 2 ) · A = V m ( 19 ) Δ = 1 π ⁢ ⁢ f 1 ⁢ ⁢ sin - 1 ⁡ ( V m AA 1 ) ( 20 )
Δ ≈ 1 π ⁢ ⁢ f 1 · V m AA 1 ( 21 )
V out(t)=V out1(t)+V out2(t)=V out2(t)+V P(t)×V S(t) (22)
V out ⁢ ⁢ 1 ⁡ ( f ) = ∑ k = - ∞ ∞ ⁢ a k ⁢ δ ⁡ ( f - kf 1 ) ( 24 )
a k = { 0 if ⁢ ⁢ k = even 1 2 ⁢ j · ( 2 ⁢ f 1 ⁢ AA 1 ⁡ ( Δ - sin ⁡ ( 2 ⁢ π ⁢ ⁢ f 1 ⁢ Δ ) 2 ⁢ π ⁢ ⁢ f 1 ) + 4 ⁢ V m π ⁢ cos ⁡ ( π ⁢ ⁢ f 1 ⁢ ⁢ Δ ) ) if ⁢ ⁢ k = 1 1 2 ⁢ j · ( 2 ⁢ f 1 ⁢ AA 1 ⁡ ( sin ⁡ ( ( k - 1 ) ⁢ π ⁢ ⁢ f 1 ⁢ Δ ) ( k - 1 ) ⁢ π ⁢ ⁢ f 1 - sin ⁡ ( ( k + 1 ) ⁢ π ⁢ ⁢ f 1 ⁢ Δ ) ( k + 1 ) ⁢ π ⁢ ⁢ f 1 ) + 4 ⁢ V m π ⁢ ⁢ k ⁢ cos ⁡ ( k ⁢ ⁢ π ⁢ ⁢ f 1 ⁢ ⁢ Δ ) ) otherwise ( 25 )
Equation (26) may be a very close approximation as ak(Δ) is flat around A=0, when
V S ⁡ ( f ) = ∑ k = - ∞ ∞ ⁢ b k ⁢ δ ⁡ ( f - k ⁡ ( 2 ⁢ ⁢ f 1 ) ) ( 27 ) b 0 = 2 ⁢ ⁢ Af 1 ⁢ Δ ⁢ ⁢ and ⁢ ⁢ b k = A k ⁢ ⁢ π ⁢ ⁢ sin ⁡ ( 2 ⁢ k ⁢ ⁢ π ⁢ ⁢ f 1 ⁢ Δ ) ⁢ ⁢ ( k > 0 ) ( 28 )
b k ≈ A k ⁢ ⁢ π ⁢ 2 ⁢ ⁢ k ⁢ ⁢ π ⁢ ⁢ f 1 ⁢ Δ = 2 ⁢ ⁢ Af 1 ⁢ Δ ( 29 )
b k ≈ 2 ⁢ ⁢ Af 1 · 1 π ⁢ ⁢ f 1 · V m AA 1 = 2 ⁢ V m π ⁢ ⁢ A 1 ( 30 )
V out ⁡ ( f ) = ∑ k = odd ⁢ a k ⁢ δ ⁡ ( f - kf 1 ) + V P ⁡ ( f ) * ∑ k = - ∞ ∞ ⁢ b k ⁢ δ ⁡ ( f - k ⁡ ( 2 ⁢ f 1 ) ) ( 31 ) V out ⁡ ( f ) = ∑ k = odd ⁢ a k ⁢ δ ⁡ ( f - kf 1 ) + ∑ k = - ∞ ∞ ⁢ b k ⁢ V P ⁡ ( f - k ⁡ ( 2 ⁢ f 1 ) ) ( 32 )
V out ⁡ ( f ) = ∑ k = odd ⁢ 4 ⁢ V m 2 ⁢ ⁢ jk ⁢ ⁢ π ⁢ δ ⁡ ( f - kf 1 ) + 2 ⁢ V m π ⁢ ⁢ A 1 ⁢ ∑ k = - ∞ ∞ ⁢ V P ⁡ ( f - k ⁡ ( 2 ⁢ f 1 ) ) ( 33 )
V OSC ⁡ ( t ) ≈ A 0 ⁢ ⁢ cos ⁢ ⁢ 2 ⁢ ⁢ π ⁢ ⁢ f c ⁢ t + A 0 ⁢ m 2 ⁡ [ cos ⁢ ⁢ 2 ⁢ ⁢ π ⁡ ( f c + f m ) ⁢ t - cos ⁢ ⁢ 2 ⁢ ⁢ π ⁡ ( f c - f m ) ⁢ t ] ( 38 )
Where the termφ(t) may reflect the phase variation due to the noise sources in the VCO. Referring to FIG. 19, there is illustrated the power spectral density of phase noise Sφφ(f). A VCO by definition is a phase integrator and, therefore, the power spectral density of the VCO output phase, in terms of the input modulating process, may be represented by:
N Th = 4 ⁢ N Th ⁢ ⁢ 1 · ( 2 ⁢ V m π ⁢ ⁢ A 1 ) 2 + ( M - 2 ) ⁢ N Th ⁢ ⁢ 1 · ( 2 ⁢ V m π ⁢ ⁢ A 1 ) 2 ( 50 )
N Th = ( M + 2 ) ⁢ N Th ⁢ ⁢ 1 · ( 2 ⁢ V m π ⁢ ⁢ A 1 ) 2 ( 51 )
In one aspect of the invention, a filter may be utilized in accordance with an amplifier in order to filter out phase noise prior to amplifying the signal and folding a phase noise harmonic on top of itself.
FIG. 21 is a schematic block diagram of a CMOS tuned amplifier that may be utilized in accordance with an embodiment of the invention. Referring to FIG. 21, the CMOS tuned amplifier 2100 may comprise a C-R-L circuit 2102 and a transistor 2104. The transistor 2104 may comprise an NMOS transistor, for example. The input voltage Vin, output voltage Vout, and the output current Iout for the CMOS tuned amplifier 2100 may be characterized by the respective graphs on diagram 2150.
In operation, the voltage to be amplified may be applied to the input Vin. When the input voltage, or the threshold voltage of the NMOS transistor 2104, is smaller than VT, the current at the output Iout is zero. The output current may be calculated utilizing the following equation:
I o = k 2 ⁢ ( V in - V T ) 2
The output current, therefore, may correspond to a second order periodic signal at the same frequency as the input voltage, as reflected on diagram 2150. The current may then be applied to a resonance tank, such as the C-R-L circuit 2102, that may extract the fundamental frequency of the current, which may correspond to the amplified version of the input sine wave into the amplifier 2100. The DC bias voltage of the input may determine the conduction angle, or the fraction of period, in radians, when the transistor 2104 conducts and the output current is not zero. In order to achieve higher linearity within the CMOS tuned amplifier 2100, the conduction angle may be increased. However, for higher power efficiency, the conduction angle may be decreased.
In one aspect of the invention, the CMOS tuned amplifier 2100 design parameters, such as conduction angle, may be determined utilizing linearity and gain requirements. Once a conduction angle is determined, the output current may be calculated utilizing the following equation:
I o = ∏ ( T c , T 1 ) · k 2 ⁢ ( V i - V T ) 2
In the above equation, the gate function, Π, may correspond to a square wave with width Tc and period T1, where the conduction time Tc may correspond to the time that the transistor 2104 is on in each cycle and T1 may correspond to the period of the input signal. The Fourier Transform of this signal may comprise impulses spaced by f1, or the fundamental frequency of the signal. The relative amplitude of the impulses may depend on the conduction angle. Π may, therefore, be decomposed into a series of impulses in the frequency domain utilizing the following equations:
∏ ( T c , T 1 ) = ∑ k = - ∞ ∞ ⁢ b k ⁢ δ ⁡ ( f - k T 1 ) , where b 0 = T c T 1 ⁢ ⁢ and ⁢ ⁢ b k = 1 k ⁢ ⁢ π ⁢ sin ⁡ ( k ⁢ ⁢ π ⁢ T c T 1 ) ⁢ ⁢ ( k > 0 )
The output current Iout may then be determined in frequency domain utilizing the following equations:
I out ⁡ ( f ) = F ⁢ { ∏ ( T c , T 1 ) } * F ⁢ { k 2 ⁢ ( V in - V T ) 2 } ⁢ I out ⁡ ( f ) = { ∑ k = - ∞ ∞ ⁢ b k ⁢ δ ⁡ ( f - k T 1 ) } * F ⁢ { k 2 ⁢ ( V in - V T ) 2 }
Depending on the design parameters of the CMOS tuned amplifier 2100, the output current spectrum may be characterized by various shapes. In an exemplary aspect of the invention, the transistor 2104 may be always on. In this case, Tc is equal to T1. The output current Iout may then be determined utilizing the following equation:
I out ⁡ ( f ) = F ⁢ { k 2 ⁢ ( V in - V T ) 2 }
FIG. 22 is a graphical representation 2200 of an exemplary amplifier input spectrum, in accordance with an embodiment of the invention. Referring to FIGS. 21 and 22, the input to amplifier 2100 may be represented as a DC level voltage, a sinusoid and a small perturbation. The input voltage Vin may then be determined from the following equations:
V in(t)=V bias +V 1 cos 2πf 1 t+V P(t)
V in(t)−V T =V bias −V T +V 1 cos 2πf 1 t+V P(t)
And using a more compressed notation, such as Vin(t)−VT=V0+V1 cos 2πf1t+VP(t), the following equation may be derived:
k 2 ⁢ ( V in ⁡ ( t ) - V T ) 2 = k 2 ⁢ ( V 0 + V 1 ⁢ cos ⁢ ⁢ 2 ⁢ π ⁢ ⁢ f 1 ⁢ t + V p ⁡ ( t ) ) 2
If the perturbation term is small, the second power of the perturbation may be ignored and the perturbation voltage in frequency domain, as well as the input voltage Vin, may be represented by the graphical depiction 2200. The following equations may then be derived:
k 2 ⁢ ( V in ⁡ ( t ) - V T ) 2 ≈ k 2 ⁢ ( V 0 2 + V 1 2 ⁢ cos 2 ⁢ 2 ⁢ πf 1 ⁢ t + 2 ⁢ V 0 ⁢ V 1 ⁢ cos ⁢ ⁢ 2 ⁢ π ⁢ ⁢ f 1 ⁢ t + 2 ⁢ V 0 ⁢ V p ⁡ ( t ) + 2 ⁢ V 1 ⁢ cos ⁢ ⁢ 2 ⁢ π ⁢ ⁢ f 1 ⁢ tV p ⁡ ( t ) ) k 2 ⁢ ( V in ⁡ ( t ) - V T ) 2 = k 2 ⁢ ( V 0 2 + V 1 2 2 + V 1 2 2 ⁢ cos ⁢ ⁢ 4 ⁢ π ⁢ ⁢ f 1 ⁢ t + 2 ⁢ V 0 ⁢ V 1 ⁢ cos ⁢ ⁢ 2 ⁢ π ⁢ ⁢ f 1 ⁢ t + 2 ⁢ V 0 ⁢ V p ⁡ ( t ) + 2 ⁢ V 1 ⁢ cos ⁢ ⁢ 2 ⁢ π ⁢ ⁢ f 1 ⁢ tV p ⁡ ( t ) )
To avoid the complications arising from aliasing, it may be assumed that the perturbation is characterized by a relatively limited bandwidth. The output current Iout in frequency domain may be determined utilizing the following equation:
I 0 ⁡ ( f ) = F ⁢ { k 2 ⁢ ( V in ⁡ ( t ) - V T ) 2 } = k 2 ⁢ { ( V 0 2 + V 1 2 2 ) ⁢ δ ⁡ ( f ) + V 1 2 2 ⁢ { 1 2 ⁢ δ ⁡ ( f - 2 ⁢ f 1 ) + 1 2 ⁢ δ ⁡ ( f + 2 ⁢ f 1 ) } + 2 ⁢ V 0 ⁢ V 1 ⁢ { 1 2 ⁢ δ ⁡ ( f - f 1 ) + 1 2 ⁢ δ ⁡ ( f + f 1 ) } + 2 ⁢ V 0 ⁢ V p ⁡ ( f ) + 2 ⁢ V 1 ⁢ { 1 2 ⁢ V p ⁡ ( f - f 1 ) + 1 2 ⁢ V p ⁡ ( f + f 1 ) } }
FIG. 23 is a graphical representation of an exemplary amplifier output spectrum including output current, in accordance with an embodiment of the invention. Referring to FIG. 23, the graphical representation 2300 may depict the output current Iout in frequency domain, as determined by the above equation. From the graph 2300 it may be determined that the noise spectrum spreads over at least twice the bandwidth of the input signal. Such noise spectrum may be used to characterize amplifiers with polynomial nonlinearity. The ratio of signal spectrum around the carrier signal to the perturbation is at a constant level and is equal to that of input. Both the fundamental frequency f1 and the perturbation may be amplified with a gain of gm, which may be achieved with an exemplary amplifier characterized by second order nonlinearity, for example. In more nonlinear amplification devices, the perturbation may be amplified at a larger gain than the carrier.
In a different aspect of the invention, if the conduction angle is less than a full period, the output current Iout may be determined by the equation:
I out ⁡ ( f ) = { ∑ k = - ∞ ∞ ⁢ ⁢ b k ⁢ δ ⁡ ( f - kf 1 ) } * F ⁢ { k 2 ⁢ ( V in - V T ) 2 }
The output current, therefore, may be characterized by replicas of the spectrum of FIG. 23 spaced by f1 in frequency. If the noise spectrum of FIG. 23 is not bandlimited to f1/2, there may be considerable folding of spectra into the amplified signal. In order to avoid the folding of noise into the amplified signal, the noise profile of the input signal may be analyzed and one or more noise characteristics may be filtered prior to the amplification action.
FIG. 24 is a schematic block diagram of a CMOS tuned amplifier 2400 utilizing noise spectrum analysis, in accordance with an embodiment of the invention. Referring to FIG. 24, the CMOS tuned amplifier 2400 may comprise a C-R-L circuit 2402, a transistor 2404, a processor 2408, and a low-pass filter 2406. The transistor 2404 may comprise an NMOS transistor, for example. The amplifier 2400 may be characterized by input voltage Vin, output voltage Vout, and the output current Iout.
In operation, the amplifier 2400 may fold one or more spectrum noise characteristics of the incoming signal. In one aspect of the invention, the amplifier 2400 may utilize the processor 2408 to analyze the noise profile of the incoming signal 2410. The processor 2408 may comprise on-chip processor and may be configured to analyze spectral content of noise profile. In this manner, the processor 2408 may configure the low-pass filter 2406 by a control signal 2412 so that the low-pass filter 2406 may filter out one or more noise harmonics and avoid folding of those harmonics in the output signal after the amplifier 2400. The processor 2408 may also be a part of a portable analyzing device utilizing spectral analysis hardware, firmware and/or software, for example.
FIG. 25 is a flow diagram of an exemplary method 2500 for processing a sinusoidal wave signal with a phase noise profile, in accordance with an embodiment of the invention. At step 2502, spectral content of the noise profile of the modulated sinusoidal signal may be analyzed. For example, an on-chip processor, or a removable device, may be utilized to analyze the spectral content and one or more noise characteristics. At step 2504, the sinusoidal signal may be low-pass filtered to remove phase noise characteristics, prior to the signal being amplified. The spectral content analysis may be utilized to determine one or more filter coefficients which configure the low-pass filter. At step 2506, the filtered modulated signal may be amplified by an amplifier.
While the invention contemplates the application of a filter in accordance with a CMOS tuned amplifier, the invention is not limited in this way. A filter in accordance with an amplifier may also be applied to other circuits or arrangements with one or more different types of amplifiers so that a phase-noise profile of a signal may be reduced prior to the signal being amplified by an amplifier.
1. A method for processing a signal with a corresponding noise profile, the method comprising:
amplifying said filtered signal, wherein the noise profile comprises a phase noise profile.
2. The method according to claim 1, further comprising determining at least one filter coefficient that is used to filter said at least one noise harmonic.
analyzing spectral content of the signal;
predicting spectral folding based on the spectral analysis of the signal; and
determining the at least one filter coefficient to reduce the predicted spectral folding.
4. The method according to claim 1, wherein said filtering comprises low pass filtering.
5. The method according to claim 1, further comprising determining at least one non-linearity characteristic of the signal.
6. The method according to claim 5, further comprising low-pass filtering at least one noise harmonic within the signal based on said determined at least one non-linearity characteristic.
7. The method according to claim 5, wherein said at least one non-linearity characteristic comprises at least one of a noise harmonic frequency and a noise harmonic amplitude.
8. A system for processing a signal with a corresponding noise profile, the system comprising:
analyze spectral content of the noise profile;
an amplifier that amplifies said filtered signal,
wherein the noise profile comprises a phase noise profile.
9. The system according to claim 8, wherein said processor determines at least one non-linearity characteristic of the signal.
10. The system according to claim 9, wherein said filter low-pass filters at least one noise harmonic within the signal based on said determined at least one non-linearity characteristic.
11. The system according to claim 9, wherein said at least one non-linearity characteristic comprises at least one of a noise harmonic frequency and a noise harmonic amplitude.
12. The system of claim 8, wherein the amplifier comprises a CMOS transistor with a nonlinear response characterized by a threshold voltage.
a tank circuit to tune the amplifier.
predict spectral folding at an output of the amplifier; and
configure the filter to reduce the predicted spectral folding.
US11005837 1999-10-21 2004-12-07 System and method for signal amplification Active 2029-12-12 US8838033B2 (en)
US20050100084A1 true US20050100084A1 (en) 2005-05-12
US8838033B2 true US8838033B2 (en) 2014-09-16
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US11005837 Active 2029-12-12 US8838033B2 (en) 1999-10-21 2004-12-07 System and method for signal amplification
US (1) US8838033B2 (en)
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US20050100084A1 (en) 2005-05-12 application
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