Source: https://patents.google.com/patent/US6671188?oq=5%2C832%2C511
Timestamp: 2018-05-26 21:37:03
Document Index: 768817807

Matched Legal Cases: ['art 7', 'art 30', 'art 5', 'art 30', 'art 7', 'arts 4']

US6671188B2 - Soft-switching power supply - Google Patents
Soft-switching power supply Download PDF
US6671188B2
US6671188B2 US10222391 US22239102A US6671188B2 US 6671188 B2 US6671188 B2 US 6671188B2 US 10222391 US10222391 US 10222391 US 22239102 A US22239102 A US 22239102A US 6671188 B2 US6671188 B2 US 6671188B2
US10222391
US20030043600A1 (en )
A switching power supply includes a transformer having a primary winding connected to a pair of a.c. input terminals via a rectifier circuit, and a secondary winding connected to a pair of d.c. output terminals via a rectifying and smoothing circuit. Connected between the pair of outputs of the rectifier circuit via an inductor and part or whole of the transformer primary, a primary switch is turned on and off at a repetition frequency higher than the frequency of the a.c. input voltage. A soft-switching circuit is provided which comprises a transformer tertiary, an ancillary switch, a capacitor, and diodes for zero-voltage switching of the primary switch.
This invention relates to electric power supplies, and particularly to a switching power supply capable of a.c. to d.c. voltage conversion, featuring provisions for attainment of closer approximation of the input current waveform to a sinusoidal wave, and a higher power factor.
Despite its undisputable advantages, this prior art switching power supply has proved to be not so satisfactory as can be desired in terms of efficiency.
The present invention seeks to improve the switching power supply of the noted prior art type for still higher efficiency without impairment of its inherent advantages.
Briefly, the invention may be summarized as a switching power supply capable of translating a.c. voltage into d.c. voltage. Included is a transformer connected to a pair of a.c. input terminals via a rectifier circuit, and to a pair of d.c. output terminals via a rectifying and smoothing circuit. The rectifier circuit has a first and a second output for providing a rectifier output voltage, the second being grounded in the preferred embodiments disclosed herein. A smoothing capacitor is connected between a first extremity of the primary winding of the transformer and the second output of the rectifier circuit, and an inductor between the first output of the rectifier circuit and the smoothing capacitor via at least part of the transformer primary. A primary switch is connected between a second extremity of the transformer primary and the second output of the rectifier circuit. The primary switch is provided with soft-switching capacitance means which can take the form of either a discrete capacitor connected in parallel therewith or parasitic capacitance of its own.
The invention particularly features a soft-switching circuit incorporated with the switching power supply of the foregoing general configuration. The soft-switching circuit comprises an additional winding electromagnetically coupled to the transformer primary, an ancillary switch connected in series therewith, and current supply means connected to the additional transformer winding for supplying thereto a current of sufficient magnitude to cause the transformer primary to develop a voltage that enables the soft-switching capacitance means to discharge. A switch control circuit is connected both to the primary switch for on-off control of the primary switch at a repetition frequency higher than the frequency of the a.c. input voltage, and to the ancillary switch in order to initiate conduction through the ancillary switch earlier than the beginning of each conducting period of the primary switch and to terminate conduction through the ancillary switch not later than the end of each conducting period of the primary switch.
Such being the improved construction of the switching power supply according to the invention, a current will flow through the inductor during the conducting periods of the primary switch. Improvements in power factor and input waveform are accomplished as the inductor current varies in amplitude in proportion with that of the a.c. input voltage.
The conduction of the ancillary switch, on the other hand, will result in current flow through the additional transformer winding, which is a tertiary in the preferred embodiments. Since the transformer primary and tertiary are electromagnetically coupled together, the current flow through the transformer tertiary will result in the discharge of the soft-switching capacitance means, with a consequent drop in the voltage across the primary switch. A zero-voltage turn-on of the primary switch is accomplished for reduction of switching loss and noise as the primary switch is turned on when the voltage across the same is reduced as above.
The primary switch operates both for improvements in power factor and input waveform and for d.c.-to-d.c. conversion. The objectives of improved power factor and improved input waveform in view are thus attained with little or no addition to the size or manufacturing cost of the switching power supply.
It will also be appreciated that the winding included in the soft switching circuit is incorporated with the transformer as a tertiary in the preferred embodiments. This feature also contributes to the compactness of the device.
A further feature of the invention resides in an ancillary charging circuit connected between a third output of the rectifier circuit and the smoothing capacitor. The third output of the rectifier circuit puts out substantially the same rectifier output voltage between itself and the noted second output of the rectifier circuit as that between the first and the second output thereof. The ancillary charging circuit comprises another additional winding of the transformer. Various specific designs will be proposed for the ancillary charging circuit.
The ancillary charging circuit is well calculated to charge the smoothing capacitor to the required degree even if the current through the primary inductor, which is for improvements in power factor and input waveform, is lessened in magnitude. The smoothing capacitor is charged both via the primary inductor and via the ancillary charging circuit. As a result, the current charging the smoothing capacitor via the primary inductor can be made less by an amount equal to the current charging the smoothing capacitor via the ancillary charging circuit than if the smoothing capacitor were charged via the primary inductor only. Power loss at the primary inductor is thus decreased, and its size can be reduced.
The ancillary charging circuit may also be utilized to make higher the voltage under which the smoothing capacitor is charged. This will serve to prevent the flow of overcurrent into the smoothing capacitor via the primary inductor at or adjacent the peaks of the a.c. input voltage. The result will be the reduction of higher harmonics of the a.c. input voltage.
An additional advantage of the ancillary charging circuit is that it makes use of another additional winding of the transformer. The ancillary charging circuit can therefore be most simplified in construction and reduced in size.
FIG. 3, consisting of (A) through (C), is a series of diagrams showing the voltage and current waveforms appearing in various parts of the FIG. 2 switch control circuit;
FIG. 4, consisting of (A) through (I), is a series of diagrams showing the voltage and current waveforms appearing in various parts of the FIG. 1 switching power supply;
FIG. 5, consisting of (A) through (E), is a series of diagrams showing on a different time scale the voltage and current waveforms appearing in various parts of the FIG. 1 device;
FIG. 8, consisting of (A) through (I), is a series of diagrams showing the voltage and current waveforms appearing in various parts of the FIG. 7 switching power supply;
FIG. 9 is a view similar to FIG. 1 but showing a fourth preferred form of switching power supply according to the invention;
FIG. 11, consisting of (A) through (E), is a series of diagrams showing the voltage and current waveforms appearing in various parts of the FIG. 10 switching power supply;
FIG. 12, consisting of (A) through (H), is a series of diagrams showing the voltage and current waveforms appearing in various parts of the FIG. 9 device on a different time scale;
FIG. 13 is a view similar to FIG. 1 but showing a sixth preferred form of switching power supply according to the invention;
FIG. 14 is also a view similar to FIG. 1 but showing a seventh preferred form of switching power supply according to the invention;
FIG. 15 is also a view similar to FIG. 1 but showing an eighth preferred form of switching power supply according to the invention;
FIG. 16 is also a view similar to FIG. 1 but showing a ninth preferred form of switching power supply according to the invention;
FIG. 17 is also a view similar to FIG. 1 but showing a tenth preferred form of switching power supply according to the invention;
FIG. 18 is also a view similar to FIG. 1 but showing an eleventh preferred form of switching power supply according to the invention;
FIG. 19 is also a view similar to FIG. 1 but showing a twelfth preferred form of switching power supply according to the invention;
FIG. 20 is also a view similar to FIG. 1 but showing a thirteenth preferred form of switching power supply according to the invention; and
FIG. 21 is also a view similar to FIG. 1 but showing a fourteenth preferred form of switching power supply according to the invention;
The switching power supply shown in FIG. 1 by way of a preferable embodiment of the invention has a pair of input terminals 1 and 2 which are to be connected to a source, not shown, of commercial a.c. voltage Vac with a frequency of, for instance, 50 Hz. A noise filter 3 is connected to this pair of input terminals 1 and 2. The noise filter 3 can be of the conventional make comprising inductors and capacitors for removal of high-frequency noise from the incoming fixed-frequency alternating current.
The noise filter 3 is connected to a rectifier circuit 4 having a first 41 and a second 42 input, four diodes D1, D2, D3 and D4, and a first 43 and a second 45 output. The pair of rectifier inputs 41 and 42 are connected to the pair of a.c. inputs 1 and 2, respectively, via the noise filter 3. The first diode D1 has its anode connected to the first rectifier input 41, to which there is also connected the cathode of the second diode D2. The third diode D3 has its anode connected to the second rectifier input 42, to which there is also connected the cathode of the fourth diode D4. The first D1 and the third D3 diode have their cathodes interconnected, and the junction 48 therebetween is connected to the first rectifier output 43. The second D2 and the fourth D4 diode have their anodes interconnected, and the junction 49 therebetween is connected to the second rectifier output 45. It will be observed that the second rectifier output 45 is grounded and will therefore be sometimes referred to as the grounded output conductor hereinafter in this specification.
At 5 is shown a transformer having a primary winding N1, a secondary winding N2, and, according to a feature of this invention, a tertiary or ancillary winding N3, all wound around a magnetic core 9 and electro-magnetically coupled together. The transformer primary N1 is tapped at 10 and thereby divided into two parts N1a and N1b. The three transformer windings N1, N2 and N3 are polarized as marked with the dots in FIG. 1. It will be seen from the markings that the transformer primary N1 and secondary N2 are opposite in polarity.
A smoothing capacitor C1, preferably an electrolytic capacitor, is connected between one extremity of the transformer primary N1 and the grounded second output conductor 45 of the rectifier circuit 4. An inductor L1 has one extremity thereof connected to the first rectifier output 43, and the other extremity to the tap 10 on the transformer primary N1, so that the inductor is connected to the smoothing capacitor C1 via the first part N1a of the transformer primary N1 in this particular embodiment of the invention. The position of the tap 10 on the transformer primary N1 is variable as required or desired. Generally, the smaller the ratio of the transformer primary first part N1a to the transformer primary second part N1b, the higher will the resulting power supply be in efficiency, but, at the same time, the less in power factor.
Shown as an insulated-gate field-effect transistor, a primary switch Q1 is connected between the other extremity of the transformer primary N1 and the second rectifier output conductor 45. The smoothing capacitor C1 is in parallel with this primary switch Q1 via the transformer primary N1.
Also connected in parallel with the primary switch Q1 is a soft-switching capacitor Cq1 which is less in capacitance than the smoothing capacitor C1. Although the soft-switching capacitor Cq1 is shown as a discrete unit, the functions intended therefor could be served by the parasitic capacitance between the drain and source of the primary switch Q1. The provision of the discrete soft-switching capacitor Cq1 is not a requirement in cases where the primary switch Q1 has a sufficient amount of such parasitic capacitance.
A diode Dq1 is connected reversely in parallel with the primary switch Q1 for its protection. The showing of this primary switch protection diode Dq1 as a discrete unit is also by way of example only. The functions intended for this diode could be served by the so-called body diode of the primary switch Q1.
The transformer secondary N2 has its opposite extremities connected respectively to the pair of output terminals 11 and 12 via a rectifying and smoothing circuit 6. The rectifying and smoothing circuit 6 comprises a rectifying diode D0 and a smoothing capacitor C0. Connected between one extremity of the transformer secondary N2 and the output terminal 11, the rectifying diode D0 is so oriented as to be conductive when the primary switch Q1 is off, and nonconductive when the primary switch Q1 is on. The capacitor C0 is connected in parallel with the transformer secondary N2 via the diode D0. A unidirectional output voltage is thus obtained between the pair of output terminals 11 and 12 for feeding a load 20 connected thereto.
The present invention particularly features a soft-switching circuit 7 for zero-voltage switching of the primary switch Q1. The soft-switching circuit 7 comprises an ancillary switch Q2, two ancillary diodes Da and Db, an ancillary capacitor Ca, and an ancillary switch protection diode Dq2, in addition to the noted transformer tertiary N3. This transformer tertiary is so polarized that the current flowing therethrough may cause a voltage to be developed across the transformer primary N1 with consequent discharge of the soft-switching capacitor Cq1. The transformer tertiary N3 has leakage inductance La. An additional inductor, not shown, may be connected in series with the transformer tertiary N3 in cases where the leakage inductance Na fails to provide a required amount of inductance.
The transformer tertiary N3, ancillary diode Da and ancillary switch Q2 are connected in series with one another. This serial connection is connected in parallel with the serial connection of the transformer primary second part N1b and primary switch Q1 via the ancillary capacitor Ca. The transformer tertiary N3 has one of its opposite extremities connected to the tap 10 on the transformer primary N1 via the ancillary capacitor Ca, and the other extremity to the source of the primary switch Q1 via the first ancillary diode Da and ancillary switch Q2. Shown as a field-effect transistor, the ancillary switch Q2 has the diode Dq2 built into it, which is so oriented as to permit reverse current flow. The ancillary switch Q2 is understood to possess parasitic capacitance between its drain and source. The ancillary diode Da is so oriented as to be forward-biased by the voltage induced across the transformer tertiary N3 during the nonconducting periods of the primary switch Q1. The second ancillary diode Db is connected in parallel with the serial circuit of the transformer tertiary N3, first ancillary diode Da, and ancillary switch Q2. The ancillary capacitor Ca and second ancillary diode Db function to cause current flow through the transformer tertiary N3, as will be detailed presently.
As shown also in FIG. 1, a switch control circuit 8 has inputs connected to the pair of output terminals 11 and 12 by way of conductors 13 and 14, respectively, an output connected to the control input of the primary switch Q1 by way of a conductor 15, and another output connected to the control input of the ancillary switch Q2 by way of a conductor 16. It is understood that the switch control circuit 8 is additionally connected to the sources of both switches Q1 and Q2, such connections being not specifically indicated in the drawings because of their impertinence to the instant invention. The switch control circuit 8 delivers a primary switch control signal Vg1 over the conductor 15 for on-off control of the primary switch Q1, and an ancillary switch control signal Vg2 over the conductor 16 for on-off control of the ancillary switch Q2.
FIG. 2 is a more detailed illustration of the switch control circuit 8. Included is a output voltage detector circuit 21 which is connected to the pair of input conductors 11 and 12, FIG. 1, by way of the conductors 13 and 14 for putting out a voltage proportional to the output voltage Vo of this power supply. The output of the output voltage detector circuit 21 is connected to one input of a differential amplifier 22, the other input of which is connected to a reference voltage source 23. The output of the differential amplifier 22 is connected both to one input of a first comparator 25 and, via a level-setting voltage source 26, to one input of a second comparator 27. The other inputs of the comparators 25 and 27 are both connected to a sawtooth generator circuit 24. The output of the first comparator 25 is connected to the control terminal of the primary switch Q1, FIG. 1, by way of the output conductor 15. The output of the second comparator 27 is connected to the control input of the of the ancillary switch Q2 by way of the output conductor 16.
As required or desired, the differential amplifier 22 could be coupled photoelectrically to both comparator 25 and level-setting voltage source 26. The output voltage detector circuit 21 could also be coupled photoelectrically to the differential amplifier 22.
Inputting the output voltage of the output voltage detector circuit 21 and the reference voltage from its source 23, the differential amplifier 22 puts out the difference Vr1 therebetween for delivery to the first comparator 25. This first comparator 25 compares the incoming difference voltage Vr1 with the sawtooth voltage Vt from the sawtooth generator 24, as indicated at (A) in FIG. 3, and puts out a series of duration-modulated primary switch control pulses Vg1 shown at (B) in the same figure, as well as at (A) in both FIGS. 4 and 5. The duration-modulated switch control pulses Vg1 are delivered over the first output conductor 15 to the gate of the primary switch Q1, making its on-off control accordingly.
The level-setting power supply 26 has a voltage set at −Vd, such that the voltage Vr2 on its output side is less than the differential amplifier output voltage Vr1 by Vd. The second comparator 27 compares this voltage Vr2 with the sawtooth voltage Vt, as indicated also at (A) in FIG. 3, and puts out another series of duration-modulated switch control pulses Vg2 of greater duration than the primary switch control pulses Vg1, as shown at (C) in FIG. 3 as well as at (B) in FIG. 4. The second series of duration-modulated switch control pulses Vg2 are delivered over the second output conductor 16 of the switch control circuit 8 to the gate of the ancillary switch Q2.
It will be observed from (A) and (B) in FIG. 4 that each ancillary switch control pulse Vg2 rises, as at t1, to initiate conduction through the ancillary switch Q2 when the primary switch Q1 is off. Each primary switch control pulse Vg1 rises, as at t2, to initiate conduction through the primary switch Q1 shortly after the conduction of the ancillary switch Q2. The time interval from t1 to t2 is so determined as to minimize switching loss when the primary switch Q1 is turned on.
Both primary switch Q1 and ancillary switch Q2 are shown to turn off at the same moment, as t5 in both FIGS. 3 and 4. This showing is not mandatory, however. The ancillary switch Q2 could be turned off at any moment from t4, when the current Iq2 through the ancillary switch Q2 drops to zero as at (F) in FIG. 4, and t5, when the primary switch Q1 goes off. For turning off the ancillary switch Q2 at t4, for instance, as indicated by the broken line at (B) in FIG. 4, a monostable multivibrator may be connected to the output of the second comparator 27 of the switch control circuit 8, as indicated by the dashed outline labeled 28 in FIG. 2. This MMV 28 may be caused to produce pulses each lasting as from t1 to t4 in FIG. 4.
In use of the FIG. 1 power supply the pair of a.c. input terminals 1 and 2 are to be connected to an unshown source of a.c. power, and the pair of d.c. output terminals 11 and 12 to the desired load 20. The smoothing capacitor C1 will be charged to the desired d.c. voltage Vc1 as the primary switch Q1 is turned on and off by the switch control circuit 8. The resulting steady-state operation of this representative switching power supply will be discussed hereinbelow with reference to FIGS. 4 and 5 which show the voltage and current waveforms appearing in various parts of the FIG. 1 circuitry.
FIG. 5 is explanatory of how the invention achieves improvements in power factor and input waveform. At (A) in this figure are shown the series of primary switch control pulses Vg1 applied by the switch control circuit 8 to the primary switch Q1. The primary switch Q1 will be turned on and off during each cycle T of the switch control signal consisting of one pulse, as from t2 to t3, and one space between such pulses, as from t3 to t5. The repetition rate of these primary switch control pulses Vg1 is now assumed to be 20 kHz. It is also understood that the 50 Hz sinusoidal a.c. voltage Vac, (E) in FIG. 5, is now applied between the pair of a.c. input terminals 1 and 2.
As the primary switch Q1 is repeatedly turned on and off, the amplitudes or peak values of the output current I4 of the rectifier circuit 4 and the current Iq1 through the primary switch Q1 will change, as at (B) and (C) in FIG. 5, in conformity with the amplitude of the a.c. input voltage Vac. Thus the a.c. input current Iac shown at (D) in FIG. 5 will closely approximate a sinusoidal wave, with consequent improvement in power factor and waveform. It is to be noted that the primary inductor L1 is connected to the tap 10 on the transformer primary N1. As a result, there will be no flow of the primary inductor current IL1 unless the rectifier output voltage V4 becomes higher than the tap voltage due to the voltage Vc1 across the smoothing capacitor C1. The rectifier output current I4 and a.c. input current Iac are both shown to flow from t1 to t6 and from t8 to t9 at (B) and (D) in FIG. 5.
The operation of the FIG. 1 power supply exclusive of the soft-switching circuit 7 will be explained in some more detail. A current will flow through the path comprising the rectifier circuit 4, primary inductor L1, second part N1b of the transformer primary N1, and primary switch Q1 during the conducting periods Ton of the primary switch Q1, as from t2 to t3 in FIG. 5. A current will flow at the same time through the path comprising the smoothing capacitor C1, transformer primary N1, and primary switch Q1. The voltage developing across the transformer secondary N2 during these primary switch conducting periods Ton will be oriented to reverse-bias the diode D0, holding the same nonconductive. Energy will therefore be stored on the transformer 5 during the primary switch conducting periods Ton, as well as on the primary inductor L1.
On the other hand, during the nonconducting periods Toff of the primary switch Q1, as from t3 to t5 in FIG. 5, both primary inductor L1 and transformer 5 will release the energy they have stored during the previous primary switch conducting period Ton, causing the flow of the current I4 through the path comprising the rectifier circuit 4, primary inductor L1, first part N1a of the transformer primary N1, and smoothing capacitor C1. This capacitor C1 will therefore be charged.
The current I4 charging the smoothing capacitor C1 during each primary switch nonconducting period Toff will diminish with time and become zero as at t4 in FIG. 5. During these nonconducting periods Toff, due to energy release from the transformer 5, there will be induced across the transformer secondary N2 a voltage oriented to cause conduction through the diode D0 of the rectifying and smoothing circuit 6. The capacitor C0 and load 18 will both be powered through the diode D0.
One cycle of switching operation, lasting as from t2 to t5 in FIG. 5, has now come to an end. The same cycle will repeat itself after t5 when the primary switch Q1 is closed again. The power supply output voltage V0 may exceed a predefined limit in the course of such repetition of switching cycles. Thereupon the switch control circuit 8 will respond by shortening the primary switch conducting periods Ton to an extent necessary to return the power supply output voltage V0 to normal. The switch control circuit 8 will also respond to a drop in the power supply output voltage V0, by making the primary switch conducting periods Ton longer until the output voltage returns to normal.
Reference is now invited to FIG. 4 again for a study of how the soft-switching circuit 7 functions for the soft switching of the primary switch Q1. As will be noted from (A) and (B) in FIG. 4, which show the primary switch control pulses Vg1 and ancillary switch control pulses Vg2, both primary switch Q1 and ancillary switch Q2 have been off before t1 in FIG. 4. Therefore, as in the t3-t5 period in FIG. 5, the diode current Ido will have flowed as at (I) in FIG. 4, and the voltage Vq1 across the primary switch Q1 and the voltage Vq2 across the ancillary switch Q2 will have been both held high before t1 as at (C) and (E) in FIG. 4.
The ancillary switch Q2 is shown turned on at t1 by the ancillary switch control signal Vg2, (B) in FIG. 4. The current Iq2 will then flow as at (F) in FIG. 4 through the path comprising the smoothing capacitor C1, transformer primary first part N1a, ancillary capacitor Ca, transformer tertiary N3, first ancillary diode Da, and ancillary switch Q2. The flow of the current Iq2 through the transformer primary N1a and transformer tertiary N3 will cause a voltage to be developed across the transformer secondary N2, such that the output rectifier diode D0 will be reverse-biased. The current Id0 through the diode D0 will become zero, as at (I) in FIG. 4, upon non-conduction of the diode D0. The clamping of the transformer primary N1 via the transformer secondary N2 will thus be eliminated.
Thereupon the soft-switching capacitor Cq1 will start discharging along the path comprising the transformer primary second part N1b, ancillary capacitor Ca, transformer tertiary N3, ancillary diode Da, and ancillary switch Q2. The voltage Vq1 across the primary switch Q1 will diminish until it becomes zero at t2, as at (C) in FIG. 4. Since the transformer tertiary N3 connected in series with the ancillary switch Q2 possesses inductance La, the current discharged by the soft-switching capacitor Cq1 will flow by the resonance of the capacitance of the capacitor Cq1 and the inductance La of the transformer tertiary N3. The current Iq2 through the ancillary switch Q2 will start rising from t1, as at (F) in FIG. 4. It is thus seen that the zero-current switching of the ancillary switch Q2 is accomplished at t1, with little or no power loss when this switch is turned on.
Upon completion of discharge by the soft-switching capacitor Cq1 at t2, the current Iq1 shown at (D) in FIG. 4 will start flowing along the path comprising the transformer tertiary N3, first ancillary diode Da, ancillary switch Q2, primary switch protection diode Dq1, and transformer primary second part N1b, and ancillary capacitor Ca due to the liberation of the energy that has been stored on the transformer primary second part N1b and transformer tertiary N3. The current Iq1 is shown at (D) in FIG. 4 as the sum of the current through the primary switch Q1 and the current through its protection diode Dq1. However, this current Iq1 will be collectively referred to as the primary switch current for simplicity.
The energy due to the inductance La of the transformer tertiary N3 will also be released along the path comprising the first ancillary diode Da, ancillary switch Q2, smoothing capacitor C1, transformer primary first part N1a, and ancillary capacitor Ca. The current Iq2 through the ancillary switch Q2 will diminish with the progress of energy release from the transformer tertiary N3. The current Iq1 through the primary switch protection diode Dq1, that is, through the primary switch Q1 will become zero at t3. Since the primary switch protection diode Dq1 is conductive from t2 to t3 in FIG. 4, the voltage Vq1 across the primary switch Q1 will be approximately zero during this period. The primary switch Q1 may therefore be turned on at zero voltage during the t2-t3 period. The primary switch control signal Vg1 is shown to go high at t2 at (A) in FIG. 4. However, in consideration of possible fluctuations in the moment the primary switch Q1 is turned on in practice, this switch may preferably be turned on midway between t2 and t3.
Since FIG. 4 is explanatory of what is taking place from t1 to t6 in FIG. 5, the primary switch protection diode Dq1 is understood to conduct at t2 in FIG. 4. Then the current IL1 through the primary inductor L1 will start increasing as at (G) in FIG. 4.
At t3, when the primary switch protection diode Dq1 becomes incapable of being held conductive, the current Iq1 through the primary switch Q1 will become zero and thereafter start flowing positive as at (D) in FIG. 4. Thus, during the ensuing t3-t4 period, the primary switch current Iq1 will flow along both the path comprising the rectifier circuit 4, primary inductor L1, transformer primary second part N1b, and primary switch Q1, and the path comprising the smoothing capacitor C1, transformer primary N1, and primary switch Q1.
It is understood that energy release from the inductance La of the transformer tertiary N3 comes to an end at t4, rather than at t3, in this particular embodiment. For this reason, during the t3-t4 period, the current Iq2 will flow as at (F) in FIG. 4 along the path comprising the transformer tertiary N3, first ancillary diode Da, ancillary switch Q2, smoothing capacitor C1, transformer primary first part N1a, and ancillary capacitor Ca. The rectifying diode D0 will be reverse biased by the voltage building up across the transformer secondary N2 when the primary switch current Iq1 is going positive as from t3 to t4, and from t4 to t5, in FIG. 4. The current Ido through this diode D0 will therefore remain zero as at (I) in FIG. 4.
The current Iq2 through the ancillary switch Q2 will be zero as at (F) in FIG. 4, whereas the current Iq1 through the primary switch Q1 will flow as at (D) in FIG. 4, from t4 to t5. During this period, as from t3 to t4, there will be current flow both along the first path comprising the first a.c. input terminal 1, filter 3, first rectifier diode D1, primary inductor L1, transformer primary second part N1b, primary switch Q1, fourth rectifier diode D4, filter 3, and second a.c. input terminal 2, and along the second path comprising the smoothing capacitor C1, transformer primary N1, and primary switch Q1. Additionally, current will also flow along a third path comprising the ancillary capacitor Ca, transformer primary second part N1b, primary switch Q1, and second ancillary diode Db. The current flowing along the first path is equivalent to the first inductance coil current IL1 shown at (G) in FIG. 4. The current flowing along the third path is equivalent to the ancillary capacitor current Ica during the t4-t5 period at (H) in FIG. 4. The ancillary capacitor current Ica during the t4-t5 period is opposite in direction to that during the t1-t4 period, so that the ancillary capacitor Ca will be charged from t1 to t4 and discharge from t4 to t5. The voltage across the ancillary capacitor Ca will drop upon its discharge from t4 to t5 and thereafter remain unchanged until t8 when the ancillary switch Q2 is turned on.
During this t4-t5 period the voltage Vq2 across the ancillary switch Q2 will be zero as at (E) in FIG. 4, and so will be the current Iq2 therethrough as at (F) in the same figure. The ancillary switch Q2 may therefore be turned off at any moment during this t4-t5 period for both zero-voltage and zero-current switching.
The conduction control of the ancillary switch Q2 is shown to end at t5 when that of the primary switch Q1 also ends, in this particular embodiment of the invention. The zero-voltage and zero-current switching of the ancillary switch Q2 is thus accomplished for reduction of switching loss when the ancillary switch is turned off. In practice, of course, the ancillary switch Q2 may be turned off at t4, as indicated by the broken line at (B) in FIG. 4, or at any other moment from t4 to t5.
As plotted at (D) in FIG. 4, the current Iq1 through the primary switch Q1 will drop to zero at t5 when this switch is turned off. The current will flow instead into the soft-switching capacitor Cq1 thereby charging the same. The voltage Vq1 across the primary switch Q1 will rise with a gradient as at (C) in FIG. 4. The zero-voltage turnoff of the primary switch Q1 has thus been achieved. The current charging the soft switching capacitor Cq1 as above will flow both along the path comprising the rectifier circuit 4, primary inductor L1, transformer primary second part N1b, and soft-switching capacitor Cq1, and along the path comprising the smoothing capacitor C1, transformer primary N1, and soft-switching capacitor Cq1.
The primary inductor L1 and transformer 5 will both release their energy during the ensuing t6-t7 period. The diode D0 of the rectifying and smoothing circuit 6 will then be forward biased by the voltage across the transformer secondary N2, so that the diode current Ido will flow as at (I) in FIG. 4. The smoothing capacitor C1 will be charged through the current path comprising the rectifier circuit 4, primary inductor L1, transformer primary first part N1a, and smoothing capacitor C1.
As indicated at (G) in FIG. 4, the current IL1 through the primary inductor L1 will drop to zero at t7. The diode D0 will stay conductive thereafter thanks to energy liberation by the transformer 5. The t7-t8 period is a repetition of the pre-t1 period in FIG. 4. One cycle of operation comes to an end at t8. Another similar cycle will restart at t8 when the ancillary switch Q2 is turned on again.
1. The primary switch Q1 is turned both on and off at zero voltage, assuring less switching loss and higher efficiency.
2. The ancillary switch Q2 is turned on at zero current and off at zero voltage and zero current, resulting in less switching loss due to this ancillary switch.
3. The current IL1 through the primary inductor L1 changes in peak value with the amplitude of the a.c. input voltage Vac, with consequent improvements in a.c. input power factor and waveform. Such improvements in power factor and waveform are accomplished with the aid of the primary switch Q1 in the d.c.-to-d.c. converter circuit comprising the smoothing capacitor C1, transformer 5, primary switch Q1, and rectifying and smoothing circuit 6. The objectives of improved power factor, improved waveform, and output voltage control are realized with the simple circuitry.
4. The soft switching circuit 7 for reduction of switching loss by the primary switch Q1 makes use of the winding N3 which is incorporated with the transformer 5 as its tertiary, thereby avoiding too much increase in the size and cost of the power supply.
5. By reason of the connection of the primary inductor L1 to the tap 10 on the transformer primary N1, the current IL1 does not flow through this inductor even if the primary switch is closed, unless the potential at the first rectifier output 43 grows higher than that at the tap 10. The first inductor current IL1 does not flow for this reason during the t0-t1, t6-t8, and t9-t10 periods in FIG. 5. Although this may seem disadvantageous from the standpoints of waveform and power factor improvements, it should also be taken into account that power loss does not occur at the primary inductor L1 as long as there is no current flow therethrough. Higher efficiency may therefore be attained without sacrifice in waveform and power factor through adjustment of the tap position on the transformer primary N1.
6. The ancillary capacitor Ca expedites current flow through the transformer tertiary N3.
The primary inductor L1 is connected to the junction between transformer primary N1 and primary switch Q1 in this alternative form of switching power supply, instead of to the tap 10 on the transformer primary N1 as in the FIG. 1 embodiment. The primary inductor L1 is therefore coupled to the smoothing capacitor C1 via the whole of the transformer primary N1 and directly to the primary switch Q1. All the other details of construction are as previously set forth with reference to FIGS. 1 and 2.
The current IL1 will flow upon conduction of the primary switch Q1 along the path comprising the first rectifier output 43, primary inductor L1, primary switch Q1 and second rectifier output conductor 45 even when the a.c. input voltage Vac is low in amplitude, as from t0 to t1, t6 to t8, and t9 to t10 in FIG. 5. For this reason the FIG. 6 power supply is preferable to its FIG. 1 counterpart from the viewpoints of improvements in input waveform and power factor. Offsetting these advantages is higher power loss as a result of the flow of the current IL1 through the primary inductor L1 practically throughout the complete cycle of the a.c. input voltage Vac. A compromise may be made by employing the FIG. 1 device where efficiency matters, and that of FIG. 6 where better input waveform and higher power factor are more important. A reduction of switching loss represents a feature common to both embodiments as they both incorporate the soft-switching circuit 7.
Another preferred form of switching power supply shown employs a modified soft-switching circuit 7 a in substitution for its FIG. 1 counterpart 7. All the other details of construction are as above stated with reference to FIGS. 1 and 2. The modified soft-switching circuit 7 a includes a quaternary winding N4 forming a part of the transformer designated 5 a in FIG. 7, in addition to all the components of the FIG. 1 soft-switching circuit 7.
A closer study of FIG. 7 will show that the ancillary capacitor Ca is connected in parallel with the transformer quaternary N4 via the second ancillary diode Db which functions as rectifier. Electromagnetically coupled to the transformer primary N1 and secondary N2, the transformer quaternary N4 is polarized to forward-bias the second ancillary diode Db during the conducting periods of the primary switch Q1. The ancillary capacitor Ca is therefore charged by the voltage developing across the transformer quaternary N4 during the conducting periods of the primary switch Q1. Charged with the polarity indicated by the plus and minus signs in FIG. 7, the ancillary capacitor Ca functions as d.c. power supply for this soft-switching circuit 7 a. The serial circuit of the transformer tertiary N3, first ancillary diode Da, and ancillary switch Q2 is connected in parallel with the ancillary capacitor Ca via the transformer quaternary N4.
The operation of the FIG. 7 embodiment exclusive of the soft-switching circuit 7 a is similar to that of the FIG. 1 embodiment exclusive of its soft-switching circuit 7. The operation of the FIG. 7 soft-switching circuit 7 a is essentially similar to that of the FIG. 1 soft-switching circuit 7. The operation of the FIG. 7 device is therefore explainable in reference to the waveform diagram of FIG. 8 which is very much similar to FIG. 4.
Both ancillary switch current Iq2, (F) in FIG. 8, and ancillary capacitor current Ica, (H) in FIG. 8, will start flowing along the path comprising the ancillary capacitor Ca, transformer quaternary N4, transformer tertiary N3, first ancillary diode Da, and ancillary switch Q2 at t1 in FIG. 8 when the ancillary switch Q2 is turned on. The current flow through the transformer tertiary N3 and quaternary N4 will cause the transformer secondary N2 to develop a voltage that is oriented to reverse-bias the diode D0 of the rectifying and smoothing circuit 6, causing nonconduction therethrough as at (I) in FIG. 8. The transformer primary N1 will be released from clamping by the voltage V0 across the capacitor C0 via the transformer secondary N2. The soft switching capacitor Cq1 will start discharging along the path comprising the transformer primary N1 and smoothing capacitor C1. Consequently, as indicated at (C) in FIG. 8, the voltage Vq1 across the primary switch Q1 will start diminishing at t1 and drop to zero at t2. The zero-voltage switching of the primary switch Q1 will thus be accomplished by turning the same on at t2 or any moment from t2 to t3. Since the transformer tertiary N3 and quaternary N4 have both inductances, the current flowing through the ancillary switch Q2 during the t1-t2 period will rise with a gradient as at (F) in FIG. 8.
At t2, when the soft switching capacitor Cq1 completes its discharge and when the primary switch Q1 is turned on as at (A) in FIG. 8, the transformer 5 a will start liberating the energy that has been stored thereon during the t1-t2 period. The primary switch current Iq1 will flow in the negative direction as at (D) in FIG. 8 along the path comprising the transformer primary N1, smoothing capacitor C1 and primary switch protection diode Dq1. The ancillary switch current Iq2 will start diminishing at t2 and drop to zero at t4, as at (F) in FIG. 8. Then, during the following t4-t5 period, the second ancillary diode Db will be forward biased by the voltage across the transformer quaternary N4, with the consequent flow of a current charging the ancillary capacitor Ca. This charge current is shown as a negative-going current at (H) in FIG. 8. A zero-voltage turnoff of the primary switch Q1 is therefore accomplished at t5 as the soft-switching capacitor Cq1 has been charged gradually as in the FIG. 1 embodiment.
The soft-switching circuit 7 a and primary inductor L1 of the FIG. 7 switching power supply function just like their counterparts of the FIG. 1 device. The FIG. 7 embodiment gains the same advantages as does that of FIG. 1.
The FIG. 9 embodiment is similar to that of FIG. 7 except that the primary inductor L1 is connected to the junction between transformer primary N1 and primary switch Q1 as in FIG. 6. The transformer used in this FIG. 9 embodiment is generally designated 5 b because, unlike the transformers 5 and 5 a disclosed above, its primary winding N1 is untapped. The operation of the FIG. 9 power supply exclusive of the soft-switching circuit 7 a is analogous with that of the FIG. 6 counterpart. The soft-switching circuit 7 a is of the same construction and connections as that of FIG. 7, so that the zero-voltage switching of the primary switch Q1 is possible as in FIG. 7.
The FIG. 10 embodiment represents an addition of an ancillary charging circuit 30 and reverse-blocking diode D5 to the FIG. 1 embodiment, all the other details of construction being as previously set forth in conjunction with FIGS. 1 and 2. The ancillary charging circuit 30 comprises an ancillary winding N5 of the transformer 5 c, a capacitor C2, two diodes D6 and D7, and an inductor L2. Connected between an additional output 44 of the rectifier circuit 4 and the smoothing capacitor C1, the ancillary charging circuit 30 provides a voltage for addition to the rectifier output voltage V4 between the additional rectifier output 44 and the grounded rectifier output 45. The reverse-blocking diode D5 is connected in series with the primary inductor L1.
The transformer ancillary winding N5 of the ancillary charging circuit 30 has one of its opposite extremities connected to both transformer primary N1 and smoothing capacitor C1. The other extremity of the transformer ancillary winding N5 is connected to the additional rectifier output 44 via a serial connection of the ancillary charging capacitor C2, ancillary charging inductor L2, and second ancillary charging diode D7. The first ancillary charging diode D6 is connected in parallel with the transformer ancillary winding N5 via the ancillary charging capacitor C2. The ancillary charging capacitor C2 is connected between ancillary charging inductor L2 and transformer ancillary winding N5. The first ancillary charging diode D6 has its anode connected to the junction between ancillary charging capacitor C2 and ancillary charging inductor L2. The additional rectifier output 44, to which is connected the ancillary charging circuit 30 as above, is connected to the junction 48 between the diodes D1 and D3 of the rectifier circuit 4.
The transformer 5 c of the FIG. 10 power supply is similar in construction to the FIG. 1 transformer 5 except for the addition of the ancillary winding N5, which of course is electromagnetically coupled to the other windings of the transformer.
The operation of the FIG. 10 embodiment except the ancillary charging circuit 30 is akin to that of the FIG. 1 embodiment. The smoothing capacitor C1 will be charged to the desired d.c. voltage Vc1 as the primary switch Q1 is turned on and off by the switch control circuit 8. The ancillary charging capacitor C2 will be charged to a voltage Vc2 by the voltage across the transformer ancillary winding N5. The resulting steady-state operation of this switching power supply, particularly of its ancillary charging circuit 30, will be discussed hereinbelow with reference to FIGS. 11 and 12 which show the voltage and current waveforms appearing in various parts of the FIG. 10 circuitry.
At (A) in this FIG. 11 is shown the series of switch control pulses Vg1 applied by the switch control circuit 8 to the primary switch Q1. The primary switch Q1 will be turned on and off during each cycle T of the switch control signal consisting of one pulse, as from t2 to t3, and one space between such pulses, as from t3 to t4. The repetition rate of the switch control signal Vg1 is assumed to be 20 kHz as in all the foregoing embodiments. At (E) in FIG. 11 is shown the 50 Hz sinusoidal a.c. voltage Vac being applied between the pair of a.c. input terminals 1 and 2.
As the primary switch Q1 is repeatedly turned on and off by the switch control pulses Vg1, the amplitudes or peak values of the output current I4 of the rectifier circuit 4 and the current Iq1 through the primary switch Q1 will change, as at (B) and (C) in FIG. 11, in conformity with the amplitude of the a.c. input voltage Vac. Thus the a.c. input current Iac shown at (D) in FIG. 11 will closely approximate a sinusoidal wave, with consequent improvement in power factor and input waveform. The rectifier output current I4 is shown at (B) in FIG. 11 as the current flowing through the junction 48 between the diodes D1 and D3 and is the sum of the current IL1 through the primary inductor L1 and the current IL2 through the ancillary charging inductor L2.
It is to be noted that the primary inductor L1 is connected to the tap 10 of the transformer primary N1 via the reverse-blocking diode D5 in this FIG. 10 embodiment. As a result, there will be no flow of the primary inductor current IL1 or of the a.c. input current Iac when the rectifier output voltage V4 is less than the tap voltage due to the voltage Vc1 across the smoothing capacitor C1, as from t0 to t1, from t6 to t8, and from t9 to t10 in FIG. 11. The rectifier output current I4 and a.c. input current Iac are both shown to flow from t1 to t6 and from t8 to t9 at (B) and (D) in FIG. 11.
The FIG. 10 power supply operates in three different modes depending upon the instantaneous value of the a.c. supply voltage Vac shown at (E) in FIG. 11. Let Va be a first voltage value that is equal to the voltage between the tap 10 on the transformer primary N1 and the grounded conductor 45 during the conducting periods of the primary switch Q1, and Vb be a second voltage value that is equal to the voltage Vc1 across the smoothing capacitor C1. Then, in the first 180 degrees of the a.c. supply voltage Vac, the power supply will operate in First Mode during the t0-t1 and t6-t7 periods when the a.c. supply voltage Vac is between 0 and first value Va, in Second Mode during the t1-t3 and t5-t6 periods when the voltage Vac is between first value Va and second value Vb, and in Third Mode during the t3-t5 period when the voltage Vac is higher than the second value Vb.
Incidentally, each negative half-cycle of the a.c. supply voltage Vac, as from t7 to t10 in FIG. 11, is inverted into the same shape as that of each positive half-cycle, as from t0 to t7, as the a.c. supply voltage is rectified by the rectifier circuit 4. The noted three modes of operation repeat themselves during the negative half-cycles. Also, in FIG. 11, the a.c. supply voltage Vac is shown to cross the second value level Vb at the termination of one conducting period Ton of the primary switch Q1. This showing is by way of example only; in practice, the crossing moment may come at other than the end of each conducting period.
In First Mode of operation, as from t0 to t1 and from t6 to t7 in FIG. 11, the primary switch current Iq1 will flow as at (C) in FIG. 11 along the path comprising the smoothing capacitor C1, transformer primary N1, and primary switch Q1 each time this switch conducts. No energy release from the transformer 5 c will occur on its output side during these periods because then the diode D0 is nonconductive. Energy will therefore be stored on the transformer 5 c. The potential at the tap 10 on the transformer primary N1 is now higher than that of the first rectifier output 43, so that there will be no flow of current IL1 through the primary inductor L1. There will be no flow of current IL2 through the ancillary inductor L2, either, because the voltage VC1 across the smoothing capacitor C1 is now higher than the rectifier output voltage V4.
The energy that has been stored as above on the transformer 5 c will be released when the primary switch Q1 subsequently goes off, with the consequent current flow along the path comprising the transformer secondary N2, diode D0, and capacitor C0. Both capacitor C0 and load 20 will therefore be fed even when the a.c. supply voltage Vac is low in instantaneous value.
In Second Mode, as from t1 to t3 and from t5 to t6 in FIG. 11, the potential at the transformer primary tap 10 will be less than that of the first rectifier output 43. There will therefore be a flow of current IL1 through the primary inductor L1. When the primary switch Q1 goes on, the primary inductor current IL1 will flow along the path comprising the first rectifier output 43, primary inductor L1, reverse-blocking diode D5, transformer primary second part N1b, primary switch Q1, and ground-potential conductor 45. Current will also flow along the path comprising the smoothing capacitor C1, transformer primary N1, and primary switch Q1. Thus the current Iq1 now flowing through the primary switch Q1, shown at (C) in FIG. 11, will be the sum of the currents flowing along the two paths just stated. The diode D5 will be off in Second Mode because the smoothing capacitor voltage Vc1 will be higher than the rectifier output voltage V4.
When the primary switch Q1 is off, as from t3 to t4 in FIG. 11, in Second Mode, on the other hand, the primary inductor current IL1 will flow to charge the smoothing capacitor C1, with energy release from the primary inductor L1. There will also be a current flow through the diode D0 of the rectifying and smoothing circuit 6 as a result of energy release from the transformer 5 c and primary inductor L1. The primary inductor current IL1 will decrease in magnitude with the progress of the energy release from the transformer 5 c and primary inductor L1.
In Third Mode, as from t3 to t5 in FIG. 11, the a.c. input voltage Vac and the rectifier output voltage V4 will be higher than the smoothing capacitor voltage Vc1, so that both first and second ancillary charging diodes D6 and D7 will be conductive. Both primary inductor current IL1 and ancillary inductor current IL2 will then flow, as will be hereinafter explained in more detail with reference to FIG. 12.
When the primary switch Q1 is on, as from t0 to t1 in FIG. 12, in response to one of the primary switch control pulses Vg1 shown at (A) in this figure, the primary inductor current IL1 will flow as at (F) in FIG. 12 along the same path as in Second Mode. There will also be a current flow in the circuit comprising the smoothing capacitor C1, transformer primary N1, and primary switch Q1. The primary switch current Iq1, (E) in FIG. 12, is the sum of the primary inductor current IL1, (F) in FIG. 12, and the current discharged by the smoothing capacitor C1. Also, during the conducting period Ton of the primary switch Q1, there will be obtained across the transformer ancillary winding N5 a voltage Vn5, (H) in FIG. 12, depending upon the ratio of the turns of the transformer primary N1 and ancillary winding N5. This voltage Vn5 is oriented to forwardly bias the first ancillary charging diode D6, so that current will flow in the closed circuit comprising the transformer ancillary winding N5, ancillary charging capacitor C2, and first ancillary charging diode D6. The ancillary charging capacitor C2 will be charged with the polarity indicated in FIG. 10, with the consequent development of a voltage VC2 across the same.
As will be noted from (G) in FIG. 12, there has been a sustained flow of current IL2 through the ancillary inductor L2 during the nonconducting period of the primary switch Q1 preceding the t0-t1 conducting period Ton of FIG. 12. This ancillary inductor current IL2 will gradually decrease in magnitude during the t0-t1 period because then the anode potential of the first ancillary charging diode D6 will be higher than that during each nonconducting period Toff of the primary switch Q1. The ancillary inductor current IL2 will flow during this conducting period Ton along the path comprising the first a.c. input terminal 1, filter 3, first rectifier diode D1, second ancillary charging diode D7, ancillary charging inductor L2, first ancillary charging diode D6, smoothing capacitor C1, fourth rectifier diode D4, filter 3, and second a.c. input terminal 2. The ancillary inductor current IL2 will increase in magnitude with the a.c. input voltage Vac.
Then, during the ensuing nonconducting period Toff of the primary switch Q1, as from t1 to t2 in FIG. 12, the smoothing capacitor C1 will be charged by the primary inductor current IL1 as in Second Mode, and the current Ido will flow through the diode D0 of the rectifying and smoothing circuit 6 as at (C) in FIG. 12. Further, as the transformer secondary N2 will be clamped by the voltage V0 across the capacitor C0, a voltage VN5 will develop across the transformer ancillary winding N5 as at (H) in FIG. 12. The transformer ancillary winding voltage VN5 during the nonconducting period Toff will be opposite in polarity to that during the conducting period Ton, reverse-biasing the first ancillary charging diode D6. Since this transformer ancillary winding voltage VN5 during the nonconducting period Toff will be opposite in polarity to the smoothing capacitor voltage VC1, the potential at the right-hand extremity, as viewed in FIG. 10, of the ancillary charging inductor L2 will be less than that during the conducting period.
The ancillary charging inductor current IL2 will rise gradually in magnitude during the nonconducting period Toff. The ancillary charging inductor current IL2 will flow along the path comprising the first a.c. input terminal 1, filter 3, first rectifier diode D1, second ancillary charging diode D7, ancillary charging inductor L2, ancillary charging capacitor C2, transformer ancillary winding N5, smoothing capacitor C1, fourth rectifier diode D4, filter 3, and second a.c. input terminal 2, charging the smoothing capacitor C1 in so doing. The smoothing capacitor C1 is charged by both primary inductor current IL1 and ancillary inductor current IL2. It will therefore be appreciated that the smoothing capacitor C1 is charged to the voltage VC1 that is higher than if, as has been the case heretofore, it is charged only by the primary inductor current IL1. The current I4 through the rectifier circuit 4, shown at (B) in FIG. 12, is the sum of the primary and ancillary inductor currents IL1 and IL2, (F) and (G) in FIG. 12.
Incidentally, FIG. 12 is meant purely to illustrate how the current and voltage signals in question change with time. Their amplitudes are shown simplified or idealized.
The FIG. 10 embodiment possesses the same advantages as that of FIG. 1 resulting from the primary inductor L1 and soft-switching circuit 7. The ancillary charging circuit 30 offers the following additional advantages:
1. The smoothing capacitor C1 is charged not only by the primary charging network comprising the rectifier circuit 4, primary inductor L1, reverse-blocking diode D5, and transformer primary first part N1a, but by the ancillary charging circuit 30 as well. If this smoothing capacitor is to be charged to the same voltage as in FIG. 1, the current IL1 flowing through the primary inductor L1 can be of smaller magnitude than in FIG. 1. Not only can the primary inductor L1 be reduced in size, but also power loss is lessened here for higher overall efficiency of the power supply. The primary inductor current IL1 may be reduced, of course, only to such a level that the desired improvements in input current waveform and power factor does not become unattainable. The ancillary charging circuit 30 has its own power loss. Yet the current flowing through this circuit for charging the smoothing capacitor C1 is only of such magnitude that the resulting power loss is negligible. Altogether, the FIG. 10 switching power supply possesses a decisive advantage over the prior art in terms of efficiency in operation and compactness in size.
2. If the current flowing through the primary inductor L1 for charging the smoothing capacitor C1 is of the same magnitude as in FIG. 1, on the other hand, then the smoothing capacitor will be charged to the voltage VC1 that is higher than that of the FIG. 1 circuitry by the amount charged by the ancillary charging circuit 30. Such higher smoothing capacitor voltage VC1 will be effective to restrict the peaks of the current flowing into the smoothing capacitor C1 at or adjacent the peaks of the rectifier output voltage V4, resulting in the reduction of the higher harmonics of the a.c. input current Iac.
3. The second ancillary charging diode D7 functions to block reverse current flow from ancillary charging circuit 30 toward primary inductor L1. The primary inductor current IL1 is reduced in magnitude in this respect, too, for further curtailment of power loss.
The FIG. 13 power supply is similar to that of FIG. 10 except for the connection of the primary inductor L1 to the transformer primary N1. The primary inductor L1 is connected via the reverse-blocking diode D5 to the junction between transformer primary N1 and primary switch Q1, instead of to the tap 10 as in FIG. 10. The tap 10 is connected only to the soft-switching circuit 7. The relationship between primary inductor L1 and transformer primary N1 in FIG. 13 is akin to that in FIG. 6, so that the FIG. 13 embodiment combines the advantages of the FIGS. 6 and 10 embodiments.
This switching power supply employs a modified ancillary charging circuit 30 a in place of its FIG. 10 counterpart 30 but is identical with the FIG. 10 embodiment in all the other details of construction. The alternate ancillary charging circuit 30 a has the first ancillary charging diode D6 connected between ancillary charging inductor L2 and transformer ancillary winding N5, and the ancillary charging capacitor C2 connected between ancillary charging inductor L2 and smoothing capacitor C1. All the other details of construction are as previously set forth in connection with the FIG. 10 ancillary charging circuit 30.
The voltage that builds up across the transformer ancillary winding N5 during the conducting periods of the primary switch Q1 is oriented to reversely bias the first ancillary charging diode D6. There will therefore be no current flow through this diode D6 that would charge the ancillary charging capacitor C2. A voltage capable of forwardly biasing the first ancillary charging diode D6 will develop across the transformer ancillary winding N5 during the nonconducting periods of the primary switch Q1, so that a closed circuit of the transformer ancillary winding N5, ancillary charging capacitor C2, and first ancillary charging diode D6 will be completed for the flow of a current charging the ancillary charging capacitor C2.
If the voltage drop across the second ancillary charging diode D7 is disregarded, the voltage between the input-side terminal of the ancillary charging inductor L2 and the ground-potential conductor 45 is equal to the rectifier output voltage V4. The voltage between the output-side terminal of the ancillary charging inductor L2 and the ground-potential conductor 45 is equal to the difference between the voltage Vc1 across the smoothing capacitor C1 and the voltage Vc2 across the ancillary charging capacitor C2. Thus the voltage VL2 across the ancillary charging inductor L2 is defined as:
V L2 =V 4−(V c1 −V c2)=V 4 −V c1 +V c2.
It is thus seen that the ancillary charging inductor current IL2 flows only when the sum of V4 and Vc2 is greater than Vc1. As in the FIG. 10 embodiment, the smoothing capacitor C1 is charged both by the current IL1 through the primary inductor L1 and by the current IL2 through the ancillary inductor L2, to the voltage Vc1 that is higher than that of the FIG. 1 embodiment.
The FIG. 15 embodiment is identical with that of FIG. 13 except for replacement of the ancillary charging circuit 30 by its modification 30 a in FIG. 14. Thus, in FIG. 15, the primary inductor L1 is connected via the reverse-blocking diode D5 to the junction between transformer primary N1 and primary switch Q1 as in FIG. 13. This device gains the same advantages as do the FIGS. 13 and 14 embodiments.
The ancillary charging circuit 30 and reverse-blocking diode D5 of the FIG. 10 embodiment lend themselves to use in the switching power supply of FIG. 7 as well. FIG. 16 shows the resulting device. The transformer 5 d shown here is equivalent to the FIG. 7 transformer 5 a plus the ancillary winding N5. The FIG. 16 embodiment combines the advantages of both FIGS. 7 and 10 embodiments.
The FIG. 17 embodiment is similar to that of FIG. 16 except that the primary inductor L1 is connected via the reverse-blocking diode D5 to the junction between transformer primary N1 and primary switch Q1 as in the FIG. 13 embodiment. The transformer 5 e is equivalent to its FIG. 16 counterpart 5 d except for the absence of the tap 10. The resulting power supply exclusive of the ancillary charging circuit 30 is similar to the FIG. 9 device, and the ancillary charging circuit 30 identical with that of FIG. 10, so that the FIG. 17 embodiment combines the advantages of the FIGS. 9 and 10 embodiments.
The ancillary charging circuit 30 of the FIG. 16 embodiment is replaceable by its FIG. 14 counterpart 30 a, as illustrated as a further preferred embodiment of the invention in FIG. 18. This FIG. 18 embodiment combines the advantages of the FIGS. 14 and 16 embodiments.
In FIG. 19 the primary inductor L1 is connected via the reverse-blocking diode D5 to the junction between transformer primary N1 and primary switch Q1, as in FIG. 17. The transformer primary N1 is therefore untapped. All the other details of construction are as set forth above with reference to FIG. 18. Having the same major circuitry, including the soft-switching circuit 7, as that of FIG. 9, and the same ancillary charging circuit 30 a as that of FIG. 14, this FIG. 19 embodiment combines the advantages of the FIGS. 9 and 14 embodiments.
The ancillary charging circuit 30 a of FIG. 14 is modifiable as shown at 30 b in FIG. 20, which shows a further preferred embodiment of the invention that is similar in all the other respects to that of FIG. 14. The modified ancillary charging circuit 30 b incorporates only the transformer ancillary winding N5 and ancillary charging diode D7. Connected between the second rectifier output 44 and smoothing capacitor C1 via the ancillary charging diode D7, the transformer ancillary winding N5 is assumed to possess leakage inductance.
During the nonconducting periods of the primary switch Q1, there will develop across the transformer ancillary winding N5 a voltage Vn5 that is oriented to forwardly bias the ancillary charging diode D7. Current will flow through the ancillary charging diode D7, charging the smoothing capacitor C1, only when the sum of the rectifier output voltage V4 and the transformer ancillary winding voltage Vn5 grows higher than the voltage Vc1 across the smoothing capacitor C1.
This FIG. 20 embodiment offers the same advantages as does that of FIG. 14 except for the lack of smoothing effects by the inductor L2 and capacitor C2 of the FIG. 14 ancillary charging circuit 30 a. Counterbalancing this shortcoming are the simplicity in construction and compactness in size of the ancillary charging circuit 30 b.
The FIG. 20 ancillary charging circuit 30 b lends itself to use with the transformer 5 c of FIGS. 13 and 15. The cathode of the reverse-blocking diode D5 may then be connected to the junction between transformer primary N1 and primary switch Q1 instead of to the tap 10 on the transformer primary. As an additional modification of the FIG. 20 embodiment, the soft-switching circuit 7 may be replaced by its FIG. 7 counterpart 7 a; in other words, the ancillary charging circuit 30 b of FIG. 20 may be substituted for the ancillary charging circuit 30 or 30 a of FIGS. 10 and 13-19.
The switching power supply according to the invention is here shown comprising a modified rectifier circuit 4 a and a modified ancillary charging circuit 30 c in places of the rectifier circuit 4 and the ancillary charging circuit 30, respectively, of the FIG. 10 embodiment and is identical therewith in all the other respects. The modified rectifier circuit 4 a has two diodes D11 and D12 in addition to the four noted diodes D1-D4. The fifth diode D11 has its anode connected to the first a.c. input 41, and its cathode to the additional rectifier output 44. The sixth diode D12 has its anode connected to the second a.c. input 42, and its cathode to the additional rectifier output 44. The additional rectifier output 44 is therefore supplied not with the outputs from the first and third diodes D1 and D3 but with the outputs from the fifth and sixth diodes D11 and D12. These diodes D11 and D12 are substantially equal in electrical characteristics to the diodes D1 and D3, so that the voltage between additional rectifier output 44 and grounded rectifier output 45 is substantially equal to the voltage V4 between first rectifier output 43 and grounded rectifier output 45.
The modified ancillary charging circuit 30 c is similar to the FIG. 10 ancillary charging circuit 30 except for the absence of the second ancillary charging diode D7. The modified circuit 7 d will nevertheless function just like the original circuit 7 as the two additional diodes D11 and D12 of the rectifier circuit 4 a serve to block reverse current flow. It is understood that these additional diodes D11 and D12 are high-frequency diodes capable of responding to changes in the current through the ancillary inductor L2 with the repeated conduction and nonconduction of the primary switch Q1. The omission of the diode D7 is inadvisable in cases where low-frequency diodes are used at D11 and D12.
The rectifier circuit 4 a and ancillary charging circuit 30 c of the FIG. 21 embodiment perform essentially the same functions as their FIG. 10 counterparts 4 and 30. The FIG. 21 embodiment thus wins the same advantages as that of FIG. 10. The rectifier circuit 4 a could be used in the FIGS. 13-20 embodiments as well.
1. All the ancillary charging circuits 30, 30 a, 30 b could do without the second ancillary diode D7. In the absence of this second ancillary diode D7 the a.c. input current Iac would flow during the t0-t1, t6-t8 and t9-t10 periods of FIG. 11 as well.
2. A high-frequency capacitor, with a capacitance less than that of the smoothing capacitor C1, could be connected between the rectifier output conductors 43 and 45 in all the embodiments disclosed. In the FIG. 1 circuitry, for instance, the current flowing along the path comprising the primary inductor L1, reverse-blocking diode D5, transformer primary first part N1a, smoothing capacitor C1, and rectifier circuit 4 during the nonconducting periods of the primary switch Q1 could be caused to bypass the rectifier circuit if such a high-frequency capacitor were incorporated. The result would be the elimination of noise due to the diodes D1-D4.
3. An autotransformer could be used in places of the transformers 5, 5 a-5 c.
4. All the switching power supplies could do without the transformer secondary N2, and the rectifying and smoothing circuit 6 could be connected in parallel with the primary switch Q1 to provide a step-up power supply.
6. An insulated-gate bipolar transistor or any other suitable semiconductor switches could be used in place of the FET switch Q1.
7. The invention could be applied to “forward” switching power supplies in which the diode D0 on the output side of the transformer conducts during the conducting periods of the primary switch Q1.
8. The soft-switching circuit 7 could be connected to the junction 10 a between transformer primary N1 and primary switch Q1, as indicated by the broken lines in FIGS. 1, 6, 10, 13-15, 20 and 21, instead of to the tap 10 on the transformer primary.
(c) a rectifier circuit connected to the pair of input terminals for rectifying the a.c. input voltage, the rectifier circuit having a first and a second output for providing a rectifier output voltage;
(d) a transformer connected to the pair of a.c. input terminals via the rectifier circuit on one hand and, on the other hand, to the pair of d.c. output terminals, the transformer having a primary and an ancillary winding which are electromagnetically coupled together;
(e) a rectifying and smoothing circuit connected between the transformer and the pair of d.c. output terminals for providing the d.c. output voltage;
(f) a smoothing capacitor connected between a first extremity of the primary winding of the transformer and the second output of the rectifier circuit;
(g) an inductor connected between the first output of the rectifier circuit and the smoothing capacitor via at least part of the primary winding of the transformer;
(h) a primary switch connected between a second extremity of the primary winding of the transformer and the second output of the rectifier circuit;
(i) soft-switching capacitance means associated with the primary switch;
(j) an ancillary switch connected in series with the ancillary winding of the transformer;
(k) current supply means connected to the ancillary winding of the transformer for supplying thereto a current of sufficient magnitude to cause the primary winding of the transformer to develop a voltage that enables the soft-switching capacitance means to discharge; and
(l) a switch control circuit connected to the primary switch for on-off control of the primary switch at a repetition frequency higher than the frequency of the a.c. input voltage, the switch control circuit being also connected to the ancillary switch in order to initiate conduction through the ancillary switch earlier than the beginning of each conducting period of the primary switch and to terminate conduction through the ancillary switch not later than the end of each conducting period of the primary switch.
2. The switching power supply of claim 1 wherein the primary winding of the transformer has a tap, and wherein the current supply means comprises:
(a) an ancillary capacitor connected between the tap of the transformer primary and the serial connection of the ancillary switch and the ancillary winding of the transformer; and
(b) an ancillary diode connected reversely in parallel with the serial connection of the ancillary switch and the ancillary winding of the transformer;
(c) the serial connection of the ancillary switch and the ancillary winding of the transformer being connected to the ancillary capacitor on the one hand and, on the other hand, to the second output of the rectifier circuit.
3. The switching power supply of claim 1 wherein the current supply means comprises:
(a) an ancillary capacitor connected between a junction between the transformer primary and the primary switch and the serial connection of the ancillary switch and the ancillary winding of the transformer; and
4. The switching power supply of claim 2 or 3 further comprising a reverse-blocking diode connected in series with the ancillary winding of the transformer and with the ancillary switch.
5. The switching power supply of claim 1 wherein the current supply means comprises:
(a) a second ancillary winding electromagnetically coupled to the primary winding of the transformer and connected in series with the serial connection of the ancillary switch and the first recited ancillary winding of the transformer;
(b) an ancillary diode for rectifying a voltage across the second ancillary winding; and
(c) an ancillary capacitor connected in parallel with the second ancillary winding via the ancillary diode, and with the serial connection of the ancillary switch and the first ancillary winding of the transformer via the second ancillary winding.
6. The switching power supply of claim 1 wherein the rectifier circuit has a third output for providing substantially the same rectifier output voltage as that between the first and the second output of the rectifier circuit, and wherein an ancillary charging circuit is provided which comprises an additional ancillary winding connected between the third output of the rectifier circuit and the smoothing capacitor, the additional ancillary winding being electromagnetically coupled to the primary winding of the transformer for providing a voltage for charging the smoothing capacitor.
7. The switching power supply of claim 6 wherein the ancillary charging circuit further comprises:
(a) an ancillary charging inductor connected to the third output of the rectifier circuit;
(b) an ancillary charging capacitor connected between the ancillary charging inductor and the additional ancillary winding of the transformer; and
(c) an ancillary charging diode connected in parallel with the serial connection of the ancillary charging capacitor and the additional ancillary winding of the transformer.
8. The switching power supply of claim 7 further comprising a second ancillary charging diode connected in series with the ancillary charging inductor.
9. The switching power supply of claim 6 wherein the ancillary charging circuit further comprises:
(b) an ancillary charging diode connected between the ancillary charging inductor and the additional ancillary winding of the transformer; and
(c) an ancillary charging capacitor connected in parallel with the serial connection of the ancillary charging diode and the additional ancillary winding of the transformer.
10. The switching power supply of claim 9 further comprising a second ancillary charging diode connected in series with the ancillary charging inductor.
11. The switching power supply of claim 6 wherein the ancillary charging circuit further comprises a second ancillary charging diode connected between the third output of the rectifier circuit and the additional ancillary winding of the transformer.
12. The switching power supply of claim 6 further comprising a reverse-blocking diode connected in series with the inductor.
13. The switching power supply of claim 6 wherein the rectifier circuit comprises:
(a) a first diode having a first electrode connected to one of the pair of a.c. input terminals, and a second electrode connected to the first and the third output of the rectifier circuit;
(b) a second diode having a first electrode connected to the second output of the rectifier circuit, and a second electrode connected to said one a.c. input terminal;
(c) a third diode having a first electrode connected to the other of the pair of a.c. input terminals, and a second electrode connected to the first and the third output of the rectifier circuit; and
(d) a fourth diode having a first electrode connected to the second output of the rectifier circuit, and a second electrode connected to said other a.c. input terminal.
14. The switching power supply of claim 6 wherein the rectifier circuit comprises:
(a) a first diode having a first electrode connected to one of the pair of a.c. input terminals, and a second electrode connected to the first output of the rectifier circuit;
(c) a third diode having a first electrode connected to the other of the pair of a.c. input terminals, and a second electrode connected to the first output of the rectifier circuit;
(d) a fourth diode having a first electrode connected to the second output of the rectifier circuit, and a second electrode connected to said other a.c. input terminal;
(e) a fifth diode having a first electrode connected to said one a.c. input terminal, and a second electrode connected to the third output of the rectifier circuit; and
(f) a sixth diode having a first electrode connected to said other a.c. input terminal, and a second electrode connected to the third output of the rectifier circuit.
US10222391 2001-08-29 2002-08-16 Soft-switching power supply Expired - Fee Related US6671188B2 (en)
JP2001259304A JP3528921B2 (en) 2001-08-29 2001-08-29 Switching power supply unit
JP2001-259304 2001-08-29
US20030043600A1 true US20030043600A1 (en) 2003-03-06
US6671188B2 true US6671188B2 (en) 2003-12-30
ID=19086695
US10222391 Expired - Fee Related US6671188B2 (en) 2001-08-29 2002-08-16 Soft-switching power supply
US (1) US6671188B2 (en)
EP (1) EP1289109A3 (en)
JP (1) JP3528921B2 (en)
US20070236966A1 (en) * 2006-04-06 2007-10-11 Junpei Uruno Uniderectional dc-dc converter
US20110058393A1 (en) * 2009-09-04 2011-03-10 Tohru Suzuki Switching power supply device
US20150062988A1 (en) * 2013-08-28 2015-03-05 University Of Central Florida Research Foundation, Inc. Hybrid zero-voltage switching (zvs) control for power inverters
US20090128278A1 (en) * 2007-11-09 2009-05-21 Vox Power Limited Switched mode power supply transformer
US9413254B2 (en) * 2014-09-17 2016-08-09 Continental Automotive Systems, Inc. DC-DC conversion circuit and method of protecting devices therein
CN105720840A (en) * 2015-06-24 2016-06-29 中兴通讯股份有限公司 Power conversion device and configuring method thereof
CN105391291B (en) * 2015-11-26 2018-03-16 华为技术有限公司 A power supply circuit
JPH08154379A (en) 1994-11-28 1996-06-11 Sanken Electric Co Ltd Dc power supply device
US7557546B2 (en) * 2006-04-06 2009-07-07 Hitachi, Ltd. Unidirectional DC-DC converter
US9484840B2 (en) * 2013-08-28 2016-11-01 University Of Central Florida Research Foundation, Inc. Hybrid zero-voltage switching (ZVS) control for power inverters
JP2003070250A (en) 2003-03-07 application
EP1289109A3 (en) 2005-03-16 application
EP1289109A2 (en) 2003-03-05 application
JP3528921B2 (en) 2004-05-24 grant
US20030043600A1 (en) 2003-03-06 application
Owner name: SANKEN ELECTRIC CO., LTD., A CORPORATION OF JAPAN,
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:MORITA, KOICHI;REEL/FRAME:013209/0146