Source: https://patents.google.com/patent/KR101488632B1/en
Timestamp: 2020-02-19 08:58:11
Document Index: 66482090

Matched Legal Cases: ['§119', '§ 119', 'Application No. 61', 'Application No. 61', 'Application No. 61', 'Application No. 61', 'art 15']

KR101488632B1 - Transmitters for wireless power transmission - Google Patents
Transmitters for wireless power transmission Download PDF
KR101488632B1
KR101488632B1 KR1020137025176A KR20137025176A KR101488632B1 KR 101488632 B1 KR101488632 B1 KR 101488632B1 KR 1020137025176 A KR1020137025176 A KR 1020137025176A KR 20137025176 A KR20137025176 A KR 20137025176A KR 101488632 B1 KR101488632 B1 KR 101488632B1
KR1020137025176A
KR20130112963A (en
2008-09-17 Priority to US9785908P priority Critical
2008-09-20 Priority to US9874208P priority
2008-09-20 Priority to US61/098,742 priority
2009-09-16 Priority to US12/561,069 priority
2009-09-16 Priority to US12/561,069 priority patent/US8532724B2/en
2009-09-17 Application filed by 퀄컴 인코포레이티드 filed Critical 퀄컴 인코포레이티드
2009-09-17 Priority to PCT/US2009/057355 priority patent/WO2010033727A2/en
2013-10-14 Publication of KR20130112963A publication Critical patent/KR20130112963A/en
2015-02-04 Publication of KR101488632B1 publication Critical patent/KR101488632B1/en
The wireless power transmitter 300 includes a transmit antenna 304 configured as a resonant tank that includes a loop inductor 480 and an antenna capacitance CA. The transmitter further includes a matching circuit 308 operatively coupled between the transmit antenna and the amplifier and a Class-E amplifier 306 configured to drive the transmit antenna. The matching circuit includes an inductance (LF) and a capacitance (CF). The matching circuit of the amplifier and the inductance LF of the inductance L2 may be realized by a single inductor 478. [ Also, the matching circuit's capacitance, CF, and antenna capacitance, CA, may be realized by a single capacitor, The receiver 608 includes a rectifier circuit 600 including a receive antenna 618 including a parallel resonator and at least two rectifier diodes.
TRANSMITTERS FOR WIRELESS POWER TRANSMISSION &lt; RTI ID = 0.0 &gt;
35 Claim of priority under U.S.C. §119
The present application claims priority under 35 USC § 119 (e) to the following applications.
U.S. Provisional Patent Application No. 61 / 098,742 entitled "MAGNETIC POWER USING A CLASS E AMPLIFIER", filed on September 19, 2008, the entire disclosure of which is incorporated herein by reference.
U.S. Provisional Patent Application No. 61 / 097,859 entitled " HIGH EFFICIENCY TECHNIQUES AT HIGH FREQUENCY " filed on September 17, 2008, the entire disclosure of which is incorporated herein by reference.
U.S. Provisional Patent Application No. 61 / 147,081 entitled " WIRELESS POWER ELECTRONIC CIRCUIT " filed on January 24, 2009, the entire disclosure of which is incorporated herein by reference.
U.S. Provisional Patent Application No. 61 / 218,838 entitled " DEVELOPMENT OF HF POWER CONVERSION ELECTRONICS "filed on June 19, 2009, the entire disclosure of which is incorporated herein by reference.
FIELD OF THE INVENTION The present invention relates generally to wireless charging, and more specifically, to devices, systems, and methods associated with portable wireless charging systems.
Typically, each power device, such as a wireless electronic device, requires its own wired charger and power source, whose power source is typically an alternating current (AC) power outlet. Such a wired configuration becomes more difficult to handle when multiple devices require charging. Approaches are being developed that use over-the-air or wireless power transmission between the transmitter and the receiver coupled to the electronic device to be charged. The receive antenna collects the radiated power and rectifies it to power available for powering the device or charging the device &apos; s battery. The wireless energy transmission may be based on a coupling between the transmitting antenna, the receiving antenna, and the rectifying circuit embedded in the host electronic device to be powered or charged. Transmitters including transmit antennas are faced with conflicting design constraints such as relatively small volume, high efficiency, low Bill of Materials (BOM), and high reliability. Therefore, there is a need to improve the transmitter design for wireless power transmission to meet various design goals.
Figure 1 illustrates a simplified block diagram of a wireless power transmission system.
Figure 2 illustrates a simplified schematic diagram of a wireless power transmission system.
Figure 3 illustrates a schematic diagram of a loop antenna according to exemplary embodiments.
4 illustrates a functional block diagram of a wireless power transmission system in accordance with an exemplary embodiment.
5 illustrates a block diagram of a wireless power transmitter in accordance with exemplary embodiments.
6A and 6B illustrate a Class-E amplifier including waveforms according to an exemplary embodiment.
7 illustrates a circuit diagram of a loaded asymmetric Class-E amplifier in accordance with an exemplary embodiment.
8 illustrates a circuit diagram of a loaded symmetric Class-E amplifier in accordance with an exemplary embodiment.
Figure 9 illustrates a circuit diagram of a loaded dual half bridge amplifier according to an exemplary embodiment.
10 illustrates a circuit diagram of a filter and matching circuit including a waveform according to an exemplary embodiment.
11A and 11B illustrate circuit diagrams of intermediate driver circuits according to exemplary embodiments.
12 illustrates a circuit diagram of portions of a wireless power transmitter in accordance with an exemplary embodiment.
13 is a flowchart of a method of transmitting wireless power in accordance with an exemplary embodiment.
14 illustrates a circuit diagram of a wireless power receiver in accordance with an exemplary embodiment.
In this specification, the word "exemplary" is used to mean "serving as an example, instance, or illustration. &Quot; Any embodiment described herein as "exemplary " is not necessarily to be construed as preferred or advantageous over other embodiments.
The following detailed description with reference to the accompanying drawings is intended as a description of exemplary embodiments of the invention and is not intended to represent only those embodiments in which the invention may be practiced. The word "exemplary " used throughout this description means" serving as an example, instance, or illustration ", and need not be construed as preferred or advantageous over other exemplary embodiments. The detailed description includes specific details for the purpose of providing a comprehensive understanding of the exemplary embodiments of the invention. It will be apparent to those skilled in the art that the exemplary embodiments of the present invention may be practiced without these specific details. In some instances, well-known structures and devices are shown in block diagram form in order to avoid obscuring the novelty of the exemplary embodiments presented herein.
As used herein, the term "wireless power" refers to any form of energy associated with electric fields, magnetic fields, electromagnetic fields, or any other form of energy transmitted from a transmitter to a receiver without the use of physical electromagnetic conductors . &Lt; / RTI &gt; In the present specification, it is set forth in order to fill the device including a power conversion in the system, e.g. mobile telephones, cordless phones, iPod ®, MP3 players, headsets, and the like by radio. In general, one basic principle of wireless energy transfer involves a magnetically coupled resonance (i. E., Resonance induction) using frequencies less than, for example, 30 MHz. However, various frequencies may be employed, including frequencies where license-free operation is permitted at relatively high radiation levels, such as less than 135 kHz (LF) or 13.56 MHz (HF). At these frequencies, which are commonly used by radio frequency identification (RFID) systems, systems must comply with interference and safety standards such as EN 300330 in Europe or FCC Part 15 norm in the United States. By way of example and not of limitation, the abbreviations LF and HF are used herein, "LF" refers to f 0 = 135 kHz, and "HF" refers to f 0 = 13.56 MHz.
1 illustrates a wireless power transmission system 100 in accordance with various exemplary embodiments. An input power 102 is provided to the transmitter 104 to generate a magnetic field 106 for providing energy transfer. Receiver 108 couples to magnetic field 106 and generates output power 110 for storage or consumption by a device (not shown) coupled to output power 110. Both the transmitter 104 and the receiver 108 are separated by a distance 112. In one exemplary embodiment, the transmitter 104 and the receiver 108 are configured in a mutual resonant relationship and when the resonant frequency of the receiver 108 and the resonant frequency of the transmitter 104 are matched, The transmission losses between the transmitter 104 and the receiver 108 are minimized when the receiver 108 is located in the "near field"
The transmitter 104 further includes a transmit antenna 114 that provides a means for energy transmission and the receiver 108 further includes a receive antenna 118 that provides a means for energy reception or coupling. The transmit and receive antennas are sized according to the applications and devices to be associated with them. As mentioned, efficient energy transfer occurs by coupling most of the energy in the near field of the transmitting antenna to the receiving antenna, rather than propagating most of the energy into a remote field in the form of an electromagnetic wave. In this near field, a coupling may be established between the transmitting antenna 114 and the receiving antenna 118. In this specification, the area around the antennas 114 and 118, where such near-field coupling may occur, is referred to as the coupling-mode area.
Figure 2 shows a simplified schematic diagram of a wireless power transmission system. The transmitter 104 driven by the input power 102 includes an oscillator 122, a power amplifier or power stage 124, and a filter and matching circuit 126. The oscillator is configured to generate the desired frequency, which may be adjusted in response to the adjustment signal 123. The oscillator signal may be amplified by the power amplifier 124 with an amplification amount responsive to the control signal 125. The filter and matching circuit 126 may be included to filter harmonics or other undesired frequencies and to match the impedance of the transmitter 104 to the transmit antenna 114.
The electronic device 120 includes a receiver 108 that includes a matching circuit 132 and a rectifier and switching circuit 134 to generate a DC power output so that the battery 108, (Not shown) that is coupled to the receiver, or to power the device electronics (not shown) coupled to the receiver. The matching circuit 132 may be included to match the impedance of the receiver 108 to the receive antenna 118.
3, the antennas used in the exemplary embodiments may be configured as a "loop" antenna 150, which may also be referred to as a "self", "resonant" or " Quot; antenna " herein. The loop antennas may be configured to include a physical core such as a ferrite core or an air core. In addition, the core core loop antenna permits placement of other components within the core region. 2) in the plane of the transmitting antenna 114 (FIG. 2) where the coupled-mode region of the transmitting antenna 114 (FIG. 2) may be more effective, May be more readily possible.
Efficient transmission of energy between the transmitter 104 and the receiver 108 occurs during matched or nearly matched resonance between the transmitter 104 and the receiver 108. [ However, even if the resonance between the transmitter 104 and the receiver 108 is not matched, energy may be delivered with lower efficiency. Rather than propagating energy from the transmit antenna to free space, energy transfer occurs by coupling energy from the near field of the transmit antenna to a nearby receive antenna where such a near field is established.
The resonant frequencies of the loop antennas are based on inductance and capacitance. While the inductance at the loop antenna is generally the inductance generated by the loop, the capacitance is generally added to the inductance of the loop antenna to create a resonant structure at the desired resonant frequency. As a non-limiting example, a capacitor 152 and a capacitor 154 may be added to the antenna to create a resonant circuit that produces a sinusoidal or quasi-sinusoidal signal 156. Thus, for larger diameter loop antennas, the size of the capacitance required to induce resonance decreases as the diameter or inductance of the loop increases. In addition, as the diameter of the loop antenna increases, the effective energy transfer area of the near field increases for "vicinity" coupled devices. Of course, other resonant circuits are possible. As another non-limiting example, a capacitor may be disposed in parallel between two terminals of the loop antenna. One of ordinary skill in the art will also recognize that, for transmit antennas, the resonant signal 156 may be an input to the loop antenna 150.
Exemplary embodiments of the present invention include coupling power between two antennas in close proximity to each other. As noted, the near field is the area around the antenna where electromagnetic fields are present, but may or may not be propagated from the antenna. Typically, they are confined to a volume close to the physical volume of the antenna. In the exemplary embodiments of the present invention, antennas, such as single and multi-turn loop antennas, are used because the majority of environments that possibly surround the antennas are dielectrics and thus have a lesser effect on the magnetic field compared to the electric field Is used for both transmit (Tx) and receive (Rx) antenna systems. Also, an antenna or a combination of magnetic and electrical antennas, which is mainly composed of "electrical" antennas (e.g., dipoles and monopoles) is also contemplated.
The Tx antenna is sufficient to achieve good coupling efficiency (e.g., > 10%) for a small Rx antenna at distances considerably greater than allowed by the previously mentioned remote field and inductive approaches It can be operated with a large antenna size and sufficiently low frequency. If the Rx antenna on the host device is placed in the coupling-mode region of the driven Tx loop antenna (i.e., within the near field or strongly coupled system) when the Tx antenna is correctly sized, high coupling efficiencies %) Can be achieved.
As described herein, "proximity" coupling and "vicinity" coupling may require different matching approaches to adapt the power source / sink to the antenna / coupling network. In addition, various exemplary embodiments provide system parameters, design targets, implementation variations, and specifications for both the LF and HF applications and for the transmitter and receiver. For example, some of these parameters and features that are required to better match a particular power conversion approach may change. The system design parameters may include various priorities and tradeoffs. Specifically, transmitter and receiver subsystem considerations may include low complexity, high transmission efficiency of the circuitry that results in a low-cost implementation.
4 illustrates a functional block diagram of a wireless power transmission system configured for direct field coupling between a transmitter and a receiver in accordance with an exemplary embodiment. The wireless power transmission system 200 includes a transmitter 204 and a receiver 208. Transmitter 204 is provided with an input power P TXin to produce a mostly non-radiative field and direct field coupling 206 is defined by the coupling factor k to provide energy transfer. The receiver 208 couples directly to the non-radiating field 206 and generates output power P RXout for storage or consumption by a battery or load 236 coupled to the output port 210. Both the transmitter 204 and the receiver 208 are separated by a predetermined distance. In one exemplary embodiment, the transmitter 204 and the receiver 208 are configured in a mutually resonant relationship and while the receiver 208 is located at the "near field" of the radiated field generated by the transmitter 204, the resonance frequency, that is, the transmission loss between f 0 and the transmitter 204 and the receiver 208, when the resonance frequency of the transmitter 204 of the matching unit 208 are at a minimum.
The transmitter 204 further includes a transmit antenna 214 that provides a means for energy transmission and the receiver 208 further includes a receive antenna 218 that provides a means for energy reception. Transmitter 204 further includes a transmit power conversion unit 220 that at least partially functions as an AC-to-AC converter. The receiver 208 further includes a receive power conversion unit 222 that at least partially functions as an AC-to-DC converter.
A resonant structure capable of efficiently coupling energy from the transmit antenna 214 to the receive antenna 218 via a magnetic field when both transmit antenna 214 and receive antenna 218 are tuned to a common resonant frequency Various transmitter configurations using capacitively loaded wire loops or multi-turn coils to form are described herein. Thus, a highly efficient wireless charging of a strongly coupled set of electronic devices (e.g., mobile phones) is described wherein the transmitting antenna 214 and the receiving antenna 218 are in close proximity, typically in excess of 30% Coupling factors. Accordingly, various transmitter concepts comprised of a wire loop / coil antenna and a transmit power conversion unit are described herein.
5 illustrates a block diagram of a wireless power transmitter in accordance with exemplary embodiments. The transmitter 300 includes a transmission power conversion unit 302 and a transmission antenna 304. Transmit power conversion unit 302 includes an amplifier 306 that is used to drive transmit antenna 304. An example of that amplifier 306 is a class-E amplifier. The filter and matching circuit 308 provides filtering and / or load matching of the drive signals generated by the amplifier 306. The term "amplifier " as used in connection with the class-E amplifier 306 is intended to refer to any of the " amplifiers "," amplifiers ", " Chopper ", or "power stage &quot;.
Transmit power conversion unit 302 further includes an oscillator 310 that generates a substantially non-modulated signal to intermediate driver 312, which in turn drives intermediate driver 312 to drive amplifier 306. The oscillator 310 may be implemented as a stable frequency source providing a square wave signal with a 50% duty cycle. The intermediate driver 312 is configured to provide appropriate drive for controlling the transistors (e.g., MOSFETs) in the amplifier 306. The different operating voltages required by the oscillator 310, the intermediate driver 312 and the amplifier 306 are generated by the DC / DC converter 314 in response to the input voltage 316. In one exemplary embodiment, the amplifier 306 receives the oscillator signal 318 at a frequency of 13.56 MHz and amplifies the oscillator signal to a power level of, for example, approximately 7 watts.
The exemplary embodiment of FIG. 5 provides an implementation based on a reduced number of components and, due to the fixed duty cycle operation, does not require additional circuitry to control the duty cycle. In addition, the exemplary embodiment of FIG. 5 may be implemented with a single transistor to produce low harmonic content on the output signal due to the resonant load network needed for Class-E operation.
The implementation also allows the antenna input impedance 322 and the load impedance 322 for class-E operation of the amplifier 306 to be designed to design a class-E implementation of the filter and matching circuit 308, 320 must be characterized. Additional figures and description herein disclose measurements and modeling to determine these impedances.
6A illustrates an amplifier configured as a Class-E amplifier in accordance with an exemplary embodiment. An example of a transmitter is composed of various amplifiers suitable for a wireless power transmitter operating at, for example, 13.56 MHz. The class-E amplifier 320 includes an active device switch 330, a load network 332, and a load 334 illustrated as a pure resistive load. The class-E amplifier 320 of FIG. 6A illustrates a single-ended class-E amplifier.
The load network 332, including inductor L1 340, capacitor C1 338, capacitor C2 336 and inductor L2 334, is coupled to the active device switch 330 through the active device switch 330 in accordance with the zero-voltage and zero- Are used to shape the current and voltage waveforms in a switching manner. This significantly reduces the switching losses since the main contributor to the inefficiency is the power loss that occurs in the active device switch 330. [ In addition, the parasitic capacitance (not shown) of the active device switch 330 (typically a FFT) is used as part of capacitor C1 338 to eliminate the negative effects of parasitic capacitance.
6B illustrates the resulting voltage and current waveforms of the active device switch 330 in a Class-E configuration. At the switch-on instant (at the center of the plot), the current and voltage on the active device switch 330 is nearly zero, leading to reduced switching losses. The same is true at the switch-off moment (at the end of the plot) when the voltage rises only when the current is already zero.
The components for the class-E amplifier 306 may be determined according to the following equation.
(This equation was originally provided by Nathan O. Sokal, inventor of Class E amplifiers. Some reference should be made here (for example, Sokal NO, Sokal AD, "Class E a New Class of High Efficiency Tuned Single Ended Switching Power Amplifiers" IEEE Journal of Solid-State Circuits, Vol. SC-10, No. 3, June 1975).
By implementation, the quality factor of the load network (Q L =? L 2 / R Load ) must be greater than 1.7879, otherwise capacitor C2 336 becomes negative and the Class-E configuration becomes inoperable. In addition, capacitor C2 336 must be greater than the collector-emitter capacitance (or drain-source capacitance) of active device switch 330. Therefore, all components of the load network depend on R Load . Since R Load varies with the coupling factor k for the receiver in the case of wireless power, the load network needs to be dynamically adjusted or designed for good tradeoffs taking into account all operating conditions.
The class-E amplifier 320 of FIG. 6A may be adapted for wireless power transmission. FIG. 7 illustrates a circuit diagram of an asymmetric Class-E amplifier 350 in accordance with an exemplary embodiment. In the transmission antenna input port, the magnetically coupled transmit antennas, and composed of the loaded receive antenna coupling network is LR circuit (equivalent resistor in Figure 7 R _eqv (362) and the equivalent inductance L _ eqv (364)) Can be expressed by a first approximation. Equivalent inductance L eqv _ (364) is part of the load network (also compared with the components inductor L2 (334) of 6a), the equivalent resistance R eqv _ (362) is the load resistance. As a result, the inductor L eqv _ (364) is supplemented by an additional series inductor, thereby increasing the quality factor (quality factor) of the load network. The quality factor must exceed 1.79, otherwise the class-E amplifier 350 can not be properly designed as illustrated for equations 1 through 6.
The supply voltage 352 provides power from which RF signals are generated based on switching of the control signal 356 driving the active device switch 358. The load network circuit includes inductor L1 354, capacitor C1 360, and capacitor C2 368.
Class-E amplifier 350 may generate harmonic content in the antenna current. To remove even-order harmonics, a symmetric class-E stage may be used. Odd-order harmonics need to be filtered by an additional filtering circuit. 8 illustrates a circuit diagram of a class-E amplifier 400 according to an exemplary embodiment. The symmetric class-E amplifier 400 includes an asymmetric class-E stage 414 comprising a first class-E stage 416 and a second class-E stage 420 configured as a mirror of the first class-E stage 416, Amplifier 350 (FIG. 7). The signal generators 406 and 426 operate 180 degrees out of phase with each other and drive the switches 408 and 428 in 180 ° phase shifted waveforms, respectively, to generate a push-pull operation.
Two stages share the same load containing the equivalent resistance R eqv _ (412) and the equivalent inductance L eqv _ (414). If maintained the equivalent resistance R eqv _ (412) and the equivalent inductance L _eqv (414) is not changed as compared to the asymmetric class -E amplifier 350 of Figure 7, capacitors C1 to C4 (410, 418, 430 , 438 will have to be doubled to maintain Class E operation. This is because the effective inductance seen each switch (408, 428) can be explained by the fact that one half of the equivalent inductance L _ eqv (i.e., inductance L _ eqv is divided into two equal halves, and is grounded at daechingjeom ).
In the first class-E stage 416, the supply voltage 402 provides power and from that power, based on the switching of the control signal 406 driving the active device switch 408, do. The first load network circuit includes inductor L1 404, capacitor C1 410, and capacitor C2 418. In the second class-E stage 420, the supply voltage 422 provides power and from that power, based on the switching of the control signal 426 driving the active device switch 428, do. The second load network circuit includes inductor L2 424, capacitor C3 430, and capacitor C4 438.
The symmetric class-E amplifier 400 further removes even-order harmonic content from the current provided to the transmit antenna. Such even-order harmonic reduction reduces the filtering circuitry otherwise needed to compensate for the supplemental second harmonic filtering. It can also provide a higher RF output power compared to an asymmetric Class-E stage, when both are operated from the same supply voltage.
It is desirable that the class-E amplifiers remain stable under different load conditions because the various electronic devices (or with respect to the transmitter) of the receiver of the electronic device or the various receiver positions generate different load conditions. Changing the load condition for the Class-E amplifier without adapting the load network of the Class-E amplifier will lead to reduced efficiency, and ultimately higher stress, on the active components. However, depending on the type of load change, the impact may be smaller or larger. Various test cases were simulated according to the component values listed in Table 1.
Table 1: Simulated test cases for Class-E amplifiers and their component values (Note: RL = target load resistance, Vcc = supply voltage for Class-E amplifier)
Now, by varying the load to be capacitive or inductive, the desired operating region for the Class-E power stage can be found. Circuit simulations indicate that a Class-E power stage can be designed to operate efficiently for various loads, such as those generated by different receiver coupling conditions that need to be supported.
The component values and the required supply voltage were calculated using the equations of Equations 1 to 6. The calculated values were optimized in the simulation to obtain the best possible efficiency with the target load (purely resistive).
9 illustrates a circuit diagram of a dual half bridge amplifier according to an exemplary embodiment. The dual half bridge amplifier 450 may be considered as a modified dual circuit of a half bridge inverter that drives a parallel tank circuit (not shown) and drives a series tank circuit (not shown). The switching voltage and current waveforms are transformational duals for those of Class-D circuitry. Compared to the class-E stage, the dual half-bridge amplifier 450 does not require any inductance or additional shunt capacitors to supplement the load network. In contrast to the conventional half bridge topology, the dual half bridge provides low dV / dt voltage waveforms, and switching is ideally performed at zero voltage moments. The switch transistor junction capacitance (e.g., the drain-source capacitance of the FET) may be considered an integral part of the capacitance required to achieve resonance in the antenna parallel tank circuit. Thus, when switches are opened and closed at zero voltage instants, there is no sudden charge and discharge of junction capacitance. However, the dual half bridge amplifier, and may as a result of any changes in the wireless power link (coupling network), such as less than indicated to be more sensitive to the equivalent inductance L _ eqv, thus changing the resonant frequency of the parallel tank circuit. . To achieve or maintain zero voltage switching, the switch voltage needs to be phase aligned with the switch current.
In the first approximation, magnetically coupled transmit and consisting of the loaded receive antenna coupling network is represented in its input port by the LR series circuit (L _ eqv (470), R _ eqv (468)) . A parallel capacitor C 1 466 is added to compensate for the inductive portion of the antenna. Proper design and adjustment of the capacitor C 1 466 is important because it produces a very high efficiency operation of the dual half bridge where any non-compensated reactive portion in the load will cause a phase-shift between the switch voltage and the switch current Thereby making it impossible to switch the transistors into a lossless mode. When the equivalent inductor L _ eqv (470) and the equivalent resistance R _ eqv (468) is so changed according to the coupling to the receiver, and in all the couplings and load conditions, the efficiency is to be maintained, the capacitor C 1 (466 ) Must be dynamically adjusted.
The supply voltage 460 provides power from which RF signals are generated based on switching of control signals 452 and 454 that respectively drive the active device switches 4456 and 458. Chokes L2 462 and L3 464 are used to provide a substantially constant current to the active device switches or load and to filter RF currents from the supply voltage 460 ). The dual half bridge amplifier 450 may also be configurable in a receiver configured to operate as a synchronous rectifier operating in a quadrant VI of the positive as part of a wireless power receiver.
FIG. 10 illustrates the circuit diagram of the filter and matching circuit 308 of FIG. 5 and their respective frequency responses. A filter and matching circuit 308, also known as a "resonant transformer" or "L-section" provides an effective approach to achieving narrowband matching and certain additional filtering effects. The impedance slope is a function of the filter 306 and the matching circuit 308 as an amplifier 306 configured as a class-E amplifier since the filter and matching circuit 308 exhibits high impedance for harmonics (e.g., 27.12 MHz and 40.68 MHz) 5). &Lt; / RTI &gt; The bandwidth or Q-factor of the filter and matching circuit 308 is related to the ratio of resistor R1 446 to resistor R2 448. The higher impedance-ratio leads to a narrower bandwidth, hence a higher filtering effect. When a class-E amplifier designed for a low target load impedance (e.g., 8 ohms) is matched to an antenna 304 that is a parallel tank (FIG. 5) typically representing high input impedance, a matching network with a high impedance ratio As a result. In the case of a fill-pad antenna this impedance may be 700 OMEGA when loaded in the receiver. This approach is particularly useful for class E amplifiers that typically need to operate from low DC supply voltages and perform almost perfectly optimally with respect to RF power output and efficiency quantum modes when designed for low target load impedances It seems interesting.
11A and 11B illustrate circuit diagrams of intermediate driver circuits according to exemplary embodiments. As illustrated in FIG. 5, the intermediate driver 312 drives the amplifier 306. The power consumption by the intermediate driver 312 reduces the overall efficiency while the output signal of the intermediate driver 312 affects the switching behavior of the amplifier 306 to affect the efficiency of the amplifier 306 configured as a class- The selection of the intermediate driver 312 contributes to the efficiency of the transmitter 300. [
Figures 11A and 11B illustrate two different intermediate driver types. Figure 11A illustrates a resonant intermediate driver 312 'that utilizes the energy stored in the gate capacitance of transistor 482 (e.g., a MOSFET) by adding inductor 480 to build a series tank circuit. Although such an approach appears to perform adequately for higher power levels and lower frequencies, it may be desirable to provide additional circuitry for the resonant gate driver (such as inductors, diodes, etc.) for power levels less than 10 watts at 13.56 MHz, , And more complex control signals) may introduce additional complexity.
11B illustrates a non-inductive intermediate driver 312 &quot; which exhibits similar efficiency as the resonant intermediate driver 312 'of FIG. 11A. By way of example, the non-conductive intermediate driver 312 'may include a totem-pole output stage having an N-channel transistor (e.g., a MOSFET) and a P-channel transistor And may be constructed as push-pull gate drivers including By implementation, the intermediate driver must provide low r DSon values, fast switching speeds, and low inductance design to prevent ringing in order to achieve high efficiency with a push-pull middle driver. To reduce resistance losses in the driver, several push-pull stages may be used in parallel.
A description of a wireless power transmitter configured to include an amplifier configured as a Class-E amplifier has been provided. Various implementation considerations generally involve the realization that low inductance values can be realized with a higher quality factor than high inductance values for a given volume. 5, since each additional DC-DC conversion introduces power loss and requires additional volume within the electronic device, the oscillator 310 in the wireless power transmitter 300, the intermediate driver 312 ), And other auxiliary components (e.g., controllers) preferably operate from the same auxiliary voltage. Since with by the the R DSon of the MOSFET type used is quite high, increasing the drain current (which can be changed in accordance with a future semiconductor) higher loss may lead, the information additional design considerations and the drain voltage (e. G. , 75 V for 100 V type) and active device switches (e.g., MOSFETs) at low drain currents. Thus, the target load impedance of the Class-E amplifier (set by the L-section matching circuit) is preferably in the range where a good trade-off between high power output and high efficiency for a given supply voltage is achieved. Optimized values in accordance with exemplary embodiments are in the range of 5 ohms to 15 ohms.
12 illustrates a circuit diagram of portions of a wireless power transmitter in accordance with an exemplary embodiment. The class-E amplifier 306 'of FIG. 12 illustrates a partial schematic of the class-E amplifier shown in FIG. 5 and described in more detail in FIGS. 6a, 7, and 8. Specifically, the class-E amplifier 306 'illustrates active device switch 330, capacitor C2 336, and inductor L2 334. Figure 12 also illustrates the filter and matching circuit 308, shown in Figure 5 and described in more detail with reference to Figure 10. The filter and matching circuit 308 of FIG. 5 includes an inductor LF 472 and a capacitor CF 474.
Figure 12 illustrates a series-configured arrangement of the inductor L2 334 of the class-E amplifier 306 'and the inductor LF 472 of the filter and matching circuit 308. Thus, amplifier inductor L2 334 and filtering inductor LF 472 may be combined into a single element inductor 476. [ The wireless power transmitter 300 is configured as a tank circuit that includes antenna inductor LA 480 and antenna capacitor CA 478 that will typically include losses that may be modeled by a series loss resistance (not shown in FIG. 12) And further includes an antenna 304. In addition, circuit elements 484 and 486 are included to model losses due to the presence of self-capacitance (e.g., in the case of a multi-turn loop antenna) and the presence of, for example, electrical stray fields and lossy dielectric materials. The magnetic coupling between the antenna inductor LA 480 and the receiving antenna is not shown in FIG. FIG. 12 illustrates a parallel-configured arrangement of filtering capacitor CF 474 and antenna capacitor CA 478. FIG. Thus, the filtering capacitor CF 474 and the antenna capacitor CA 478 may be combined into a single element capacitor 482.
12 illustrates an example of how a transmit antenna may be efficiently driven by a Class-E amplifier that is reduced to a single inductor 476, a single capacitor 482, and a loop antenna inductor 480 of components. do. Thus, a wireless power transmitter has been described that generates low BOMs due to the use of amplifiers, matching filters, and transmit antenna combinations that allow a combination of reactive components. In addition, the choice of amplifier, matching filter, and transmit antenna also allows for a combination of reactive components, thereby also generating a reduced number of components. Also, in an exemplary embodiment using a symmetric Class-E amplifier, the second harmonics of the antenna current are eliminated.
13 illustrates a flowchart of a method of transmitting wireless power in accordance with an exemplary embodiment. The method 500 for transmitting wireless power is supported by the various structures and circuits described herein. The method 500 includes driving (502) a transmit antenna from an amplifier through a matching circuit. The method 500 further includes resonating (504) the transmit antenna according to the transmit antenna capacitance realized in the matching capacitor and the matching circuit.
As an example of a wireless power receiver, Figure 14 illustrates a circuit diagram of a wireless power receiver in accordance with an exemplary embodiment. According to an exemplary embodiment, the wireless power receiver 608 includes a resonant receive antenna 618 that includes an inductive loop L 2 632 and a capacitor C 2 634, and a passive double diode full wave rectifier circuit 600. . The rectifier circuit 600 includes a diode D 21 628 and a diode D 22 630. The rectifier circuit 600 further includes a high frequency (HF) choke L HFC 624 and a high frequency (HF) block capacitor C HFB 626. The DC path is closed through the antenna loop. HF choke 624 acts as a current sink, with a 50% diode conduction cycle D, and a voltage transformation factor M of 0.5. Also, the input impedance seen at terminals A 2 and A 2 'at the fundamental frequency is approximately four times the load resistance R L (636).
The proper selection of diodes for the rectifier circuit 600 may reduce circuit losses and increase overall efficiency. For rectification efficiency, the diodes are connected to various parameters including peak repetitive inverse voltage (V RRM ), average rectified forward current (I 0 ), maximum instantaneous forward voltage (V F ), and junction capacitance (C j ) May be selected on the basis of. V RRM and I 0 are the maximum ratings of the diodes, while V F and C j are characteristic values that affect the efficiency of the rectifier.
Various diodes have been tested and the voltage, current, and instantaneous power of each diode during the simulation have been calculated to characterize the switching behavior of each type. Different switching behavior of the tested diodes was observed, with the diode with the largest C j (MBRA340T3) exhibiting the worst switching behavior, but with the smallest on-state loss due to the reduced forward voltage. For the various diode types tested, on-state losses were dominant. Specifically, the switching loss varies with the junction capacitance, and the on-state loss varies with the forward voltage. Thus, the total loss depends on the ratio of C j and U F , and on the operating point of the diode depending on the load resistance R L (636).
The configuration of the two parallel PMEG4010EH diodes proved to be the best choice because the switching losses of this diode type are very small and the on-state losses are reduced due to the parallel configuration. Rectifier diodes may be implemented as double diodes (second diodes shown as phantoms) to reduce conduction losses. The current is divided equally into both diodes, which changes the operating point of each diode compared to a single diode solution. It was also observed that a single MBRS2040LT3 performed equally well because the forward voltage was significantly lower and the switching losses were still moderate compared to PMEG4010EH. Thus, for one exemplary embodiment, a diode with a junction capacitance of about 50 pF and a forward voltage of about 308 mV at 1 A is an acceptable option.
Those skilled in the art will appreciate that control information and signals may be represented using any of a variety of different technologies and techniques. For example, data, instructions, commands, information, signals, bits, symbols, and chips that may be referenced throughout the above description may refer to voltages, currents, electromagnetic waves, magnetic fields or magnetic particles, Chains or optical particles, or any combination thereof.
Those skilled in the art will recognize that the various illustrative logical blocks, modules, circuits, and algorithm steps described in connection with the embodiments disclosed herein may be implemented as electronic hardware, computer software controlled, or combinations of both As will be appreciated by those skilled in the art. To clearly illustrate the interchangeability of hardware and software, the various illustrative components, blocks, modules, circuits, and steps described above have been described above generally in terms of their functionality. Whether such functionality is implemented as hardware or software depends upon the particular application and design constraints imposed on the overall system. Skilled artisans may implement the described functionality in varying ways for each particular application, but such implementation decisions should not be interpreted as causing a departure from the scope of the exemplary embodiments of the present invention.
The various illustrative logical blocks, modules, and circuits described in connection with the embodiments disclosed herein may be implemented or performed with a general purpose processor, a digital signal processor (DSP), an application specific integrated circuit (ASIC), a field programmable gate array Other programmable logic devices, discrete gate or transistor logic, discrete hardware components, or any combination thereof designed to perform the functions described herein. A general purpose processor may be a microprocessor, but, in the alternative, the processor may be any conventional processor, controller, microcontroller, or state machine. The processor may also be implemented as a combination of, for example, a DSP and a microprocessor, a plurality of microprocessors, one or more microprocessors in cooperation with a DSP core, or any other such configuration.
The control steps of a method or algorithm described in connection with the embodiments disclosed herein may be embodied directly in hardware, in a software module executed by a processor, or in a combination of the two. The software module may be a random access memory (RAM), a flash memory, a read only memory (ROM), an electrically programmable ROM (EPROM), an electrically erasable programmable ROM (EEPROM), registers, a hard disk, Or any other form of storage medium known in the art. An exemplary storage medium is coupled to the processor such that the processor can read information from, and write information to, the storage medium. Alternatively, the storage medium may be integral to the processor. The processor and the storage medium may reside in an ASIC. The ASIC may reside in a user terminal. Alternatively, the processor and the storage medium may reside as discrete components in a user terminal.
In one or more exemplary embodiments, the described control functions may be implemented in hardware, software, firmware, or any combination thereof. When implemented in software, the functions may be stored or transmitted as one or more instructions or code on a computer readable medium. Computer-readable media includes both communication media and computer storage media including any medium that facilitates transfer of a computer program from one location to another. The storage medium may be any available media that can be accessed by a computer. By way of example, and not limitation, such computer-readable media can be RAM, ROM, EEPROM, CD-ROM, or other optical disk storage, magnetic disk storage or other magnetic storage devices, Or any other medium that can be used to carry or store desired program code in the form of data structures. Also, any connection is suitably referred to as a computer readable medium. For example, if the software is transmitted from a web site, server, or other remote source using wireless technologies such as coaxial cable, fiber optic cable, twisted pair, digital subscriber line (DSL), or infrared, radio and microwave, Cable, fiber optic cable, twisted pair, DSL, or wireless technologies such as infrared, radio and microwave are included in the definition of the medium. As used herein, a disk and a disc comprise a compact disc (CD), a laser disc, an optical disc, a digital versatile disc (DVD), a floppy disc and a Blu-ray disc, discs reproduce data magnetically, while discs reproduce data optically with lasers. Also, the above combinations should be included within the scope of computer readable media.
A wireless power transmitter, the wireless power transmitter comprising:
A transmission circuit configured as a resonant tank, the transmission circuit comprising a first capacitor coupled to a first inductor, the transmission circuit wirelessly transmitting power in response to a time-varying signal to power or charge the receiver device The transmitting circuit being configured to transmit a signal; And
A driver circuit operably coupled to the transmitting circuit, wherein the driver circuit is configured to generate the time-varying signal,
As the first driver stage circuit,
A first switch configured to control the time-varying signal in response to a first control signal;
A second capacitor coupled in parallel to the first switch;
A third capacitor coupled to the first switch and the second capacitor; And
The first driver stage circuit comprising a first inductor coupled to the first switch and the second capacitor; And
A second driver stage circuit,
A second switch configured to control the time-varying signal in response to a second control signal;
A fourth capacitor coupled in parallel to the second switch;
A fifth capacitor coupled to the second switch and the fourth capacitor; And
And a third inductor coupled to the second switch and the fourth capacitor. The second driver stage circuit
Wherein the transmitting circuit is coupled to the driver circuit in parallel.
Wherein the first driver stage circuit is configured as a class-E amplifier circuit and the second driver stage circuit is configured as a class-E amplifier circuit.
Wherein the quality factor of the driver circuit is at least 1.79.
Further comprising a first supply voltage coupled in series with the second inductor and a second supply voltage coupled in series with the third inductor.
A method for transmitting wireless power, the method comprising:
Transmitting power wirelessly by a transmission circuit in response to a time-varying signal, the transmission circuit comprising a first capacitor configured as a resonant tank and coupled to a first inductor, step; And
Generating a time-varying signal by a driver circuit, the driver circuit comprising a first switch configured to control the time-varying signal in response to a first control signal, a second capacitor coupled in parallel to the first switch, A first driver stage circuit including a third capacitor coupled to the first switch and the second capacitor and a second inductor coupled to the first switch and the second capacitor, Circuit includes a second switch configured to control the time-varying signal in response to a second control signal, a fourth capacitor coupled in series with the second switch, a second capacitor coupled to the second switch and the fourth capacitor, 5 capacitor, and a third inductor coupled to the second switch and the fourth capacitor, the second driver stage circuit comprising: And transmitting the radio power to the base station.
Supplying power to the driver circuit by a first supply voltage, and
Further comprising supplying power to the driver circuit by a second supply voltage,
Wherein the first supply voltage is coupled in series with the second inductor and the second supply voltage is coupled in series with the third inductor.
A driver circuit for use in wireless power transmission, the driver circuit comprising:
Means for transmitting power wirelessly in response to a time-varying signal, said transmitting means comprising means for storing energy in a first magnetic field, and means for storing energy in a first electric field coupled; And
Means for generating said time-varying signal, said means for generating comprising: first means for controlling said time-varying signal in response to a first control signal; first means for generating energy in a second electric field coupled in parallel to said controlling means; Means for storing energy in a third electric field coupled to said first means for controlling and means for storing energy in said second electric field and means for storing energy in said second electric field and means for storing energy in said second electric field, And means for storing energy in a second magnetic field coupled to the means for storing, wherein the means for generating the time-varying signal comprises means for controlling the time-varying signal in response to a second control signal Means for storing energy in a fourth electric field coupled in parallel to said second means for controlling, means for storing energy, means for controlling said second means and said fourth electric field Means for storing energy in a fifth electric field coupled to means for storing energy in the first electric field and means for storing energy in a third magnetic field coupled to means for storing energy in the second means for controlling and in the fourth electric field, And a second generating stage, wherein the second generating stage comprises the means for generating.
Said transmitting means being coupled in parallel to said generating means.
Wherein the first means for controlling is configured as a class-E amplifier circuit, and the second means for controlling is configured as a class-E amplifier circuit.
Wherein the quality factor of the means for generating is at least 1.79.
First means for supplying power coupled in series with means for storing energy in the second magnetic field, and second means for supplying power coupled in series with means for storing energy in the third magnetic field A driver circuit.
Wherein the driver circuit comprises the first driver stage circuit configured as a class-E amplifier circuit and the second driver stage circuit configured as a mirror of the first driver stage circuit. transmitter.
Wherein the control signals are configured to shift 180 degrees from each other to drive the first and second switches with a 180 degree phase shifted waveform.
Wherein the first capacitor of the transmission circuit is equal to, or coupled with at least one of the second, third, fourth or fifth capacitors.
Wherein the first and second supply voltages comprise the same supply voltage.
Wherein the driver circuit comprises the first driver stage circuit configured as a class-E amplifier circuit and the second driver stage circuit configured as a mirror of the first driver stage circuit. / RTI &gt;
Wherein the control signals are configured to be 180 degrees out of phase with each other to drive the first and second switches with a 180 degree phase shifted waveform.
Wherein the first capacitor of the transmitting circuit is equal to or at least associated with at least one of the second, third, fourth or fifth capacitors.
The means for generating the time-varying signal includes first means for controlling the time-varying signal configured as a class-E amplifier circuit and second means for controlling the time-varying signal configured as a mirror of first means for controlling the time-varying signal A &lt; / RTI &gt; symmetrical class-E amplifier.
Wherein the control signals are configured to be 180 degrees phase shifted from each other to drive the first means for controlling and the second means for controlling each having a 180 degree phase shifted waveform.
Wherein the means for storing energy in the first electric field of the transmitting circuit comprises means for storing energy in at least one of the means for storing energy in a second, third, fourth or fifth electric field, Driver circuit.
KR1020137025176A 2008-09-17 2009-09-17 Transmitters for wireless power transmission KR101488632B1 (en)
US9874208P true 2008-09-20 2008-09-20
US61/098,742 2008-09-20
US12/561,069 2009-09-16
US12/561,069 US8532724B2 (en) 2008-09-17 2009-09-16 Transmitters for wireless power transmission
PCT/US2009/057355 WO2010033727A2 (en) 2008-09-17 2009-09-17 Transmitters for wireless power transmission
KR20130112963A KR20130112963A (en) 2013-10-14
KR101488632B1 true KR101488632B1 (en) 2015-02-04
ID=41152045
KR1020137001842A KR20130025434A (en) 2008-09-17 2009-09-17 Transmitters for wireless power transmission
KR1020117008732A KR101435249B1 (en) 2008-09-17 2009-09-17 Transmitters for wireless power transmission
KR1020137025176A KR101488632B1 (en) 2008-09-17 2009-09-17 Transmitters for wireless power transmission
KR1020137001843A KR101343706B1 (en) 2008-09-17 2009-09-17 transmitters for wireless power transmission
US (2) US8532724B2 (en)
EP (1) EP2342796B1 (en)
JP (2) JP2012503469A (en)
KR (4) KR20130025434A (en)
CN (2) CN104617678B (en)
WO (1) WO2010033727A2 (en)
CN101919708B (en) * 2010-07-05 2011-08-17 深圳市开立科技有限公司 Dual wireless ultrasonic probe and biological ultrasonic echo signal acquisition system
US8552911B2 (en) * 2011-07-11 2013-10-08 Ronald Edgar Ham Automatic electronically tuned electrically small transmitting antenna system
WO2013054386A1 (en) 2011-10-14 2013-04-18 Empire Technology Development Llc Mobile terminal, power transfer system and computer-readable storage medium
WO2013059300A2 (en) * 2011-10-21 2013-04-25 Qualcomm Incorporated Load impedance detection for static or dynamic adjustment of passive loads
WO2013129859A1 (en) * 2012-02-28 2013-09-06 한국전기연구원 Wireless power transmission end including wireless power transmission network
KR101455697B1 (en) * 2012-02-28 2014-11-03 한국전기연구원 A wireless power transmitter including wireless power transmission network
MX350033B (en) * 2012-05-11 2017-08-23 Momentum Dynamics Corp A method of and apparatus for generating an adjustable reactance.
KR101938650B1 (en) * 2012-05-31 2019-01-15 삼성전자주식회사 Near-Field Wireless Transceiver Circuit and Mobile Terminal
KR101931256B1 (en) 2012-07-25 2018-12-20 삼성전자주식회사 Wireless power reception apparatus and method
US10084345B2 (en) * 2013-03-14 2018-09-25 Mediatek Singapore Pte. Ltd. Resonant wireless power driver with adjustable power output
CN103746462B (en) 2013-07-11 2016-01-20 重庆米亚车辆技术有限公司 A kind of bilateral LCC compensating network for wireless power transmission and tuning methods thereof
KR20160058794A (en) * 2013-09-10 2016-05-25 이피션트 파워 컨버젼 코퍼레이션 High efficiency voltage mode class d topology
US9882435B2 (en) * 2013-10-31 2018-01-30 Mitsubishi Electric Engineering Company, Limited Resonant type high frequency power supply device and switching circuit for resonant type high frequency power supply device
US9735605B2 (en) * 2014-06-17 2017-08-15 Qualcomm Incorporated Methods and systems for object detection and sensing for wireless charging systems
WO2016073867A1 (en) * 2014-11-07 2016-05-12 Murata Manufacturing Co., Ltd. Variable-distance wireless-power-transfer system with fixed tuning and power limiting
DE202015001106U1 (en) 2015-02-12 2015-05-04 IAB - Institut für Angewandte Bauforschung Weimar gemeinnützige GmbH Arrangement of wirelessly couplable components
WO2016161280A1 (en) 2015-04-01 2016-10-06 The Regents Of The University Of Michigan Double-sided lclc-compensated topology for capacitive power transfer
US10298058B2 (en) * 2015-05-04 2019-05-21 The Regents Of The University Of Colorado, A Body Corporate Wireless power transfer
CN104901630B (en) * 2015-05-27 2018-04-17 复旦大学 Realize the tunable radio frequency phase difference power amplification circuit of linear ablation
US10431991B2 (en) 2015-09-23 2019-10-01 Intel Corporation Tuning in a wireless power transmitter
JP6487825B2 (en) * 2015-11-11 2019-03-20 株式会社ダイヘン Non-contact power transmission system and power transmission device
JP2017093180A (en) * 2015-11-11 2017-05-25 株式会社ダイヘン Noncontact power transmission system, and power transmission device
CN107785939A (en) * 2016-08-24 2018-03-09 东莞宝德电子有限公司 Wireless charging circuit and its charging panel
US20180337559A1 (en) * 2017-05-18 2018-11-22 Integrated Device Technology, Inc. Power limiting in a wireless power transmitter
JPH10256957A (en) * 1997-03-13 1998-09-25 Nagano Japan Radio Co Device and system for transmitting power
KR20050005480A (en) * 2002-05-16 2005-01-13 코닌클리즈케 필립스 일렉트로닉스 엔.브이. Single stage power converter for contact-less energy transfer
JP2853207B2 (en) 1989-10-12 1999-02-03 三菱電機株式会社 Power supply system
DE69836468T2 (en) * 1997-08-08 2007-09-13 Meins, Jürgen, Prof. Dr. Ing. Method and device for contactless power supply
JP3840765B2 (en) 1997-11-21 2006-11-01 神鋼電機株式会社 Primary power supply side power supply device for contactless power transfer system
EP1017146B1 (en) 1998-05-18 2005-10-19 Seiko Epson Corporation Overcharge protection, charger, electronic device and timepiece
JP3649374B2 (en) 1998-11-30 2005-05-18 ソニー株式会社 Antenna device and card-like storage medium
WO2001001553A1 (en) 1999-06-25 2001-01-04 The Board Of Trustees Of The University Of Illinois Dynamically-switched power converter
JP4491883B2 (en) 2000-01-07 2010-06-30 シンフォニアテクノロジー株式会社 Non-contact power feeding device
JP2002010534A (en) 2000-06-22 2002-01-11 Atr Adaptive Communications Res Lab Receiving device
WO2003028197A2 (en) 2001-09-26 2003-04-03 Koninklijke Philips Electronics N.V. Split topology power supply architecture
JP3663397B2 (en) * 2002-08-30 2005-06-22 株式会社東芝 High frequency power amplifier
JP2004206937A (en) 2002-12-24 2004-07-22 Matsushita Electric Ind Co Ltd High frequency oscillator
JP4657574B2 (en) 2002-12-25 2011-03-23 パナソニック株式会社 Non-contact IC card reader / writer
JP4216647B2 (en) * 2003-05-29 2009-01-28 古野電気株式会社 Ultrasonic transmitter, ultrasonic transmitter / receiver, and detector
JP3931163B2 (en) * 2003-08-14 2007-06-13 松下電器産業株式会社 Antenna matching device
EP1646122A1 (en) 2004-10-06 2006-04-12 Nokia Corporation Multilayer printed circuit board comprising a battery charging circuitry and an induction coil
DE102005005812A1 (en) * 2005-02-09 2006-08-17 Atmel Germany Gmbh Circuit arrangement and method for supplying power to a transponder
WO2007026428A1 (en) 2005-08-31 2007-03-08 Kyocera Corporation Piezoelectric resonator
JP4563950B2 (en) 2006-03-14 2010-10-20 富士通株式会社 Contactless charging system
JP2008125198A (en) 2006-11-09 2008-05-29 Ishida Co Ltd Non-contact feeder system
JP4899917B2 (en) * 2007-02-20 2012-03-21 セイコーエプソン株式会社 Power transmission control device, power transmission device, electronic device, and tan δ detection circuit
2009-09-16 US US12/561,069 patent/US8532724B2/en active Active
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CN102144239B (en) 2015-11-25 bidirectional wireless power transmission
EP3039770B1 (en) 2020-01-22 Impedance tuning
US9698761B2 (en) 2017-07-04 Dynamic resonant matching circuit for wireless power receivers
2013-09-24 A107 Divisional application of patent