Source: http://www.google.de/patents/US5757845?hl=de
Timestamp: 2013-05-19 06:36:32
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Patent US5757845 - Adaptive spread spectrum receiver - Google PatenteSuche Bilder Maps Play YouTube News Gmail Drive Mehr » Erweiterte Patentsuche | Webprotokoll | Anmelden Erweiterte Patentsuche PatenteSampled signals SPS of a spread-spectrum received signal are fed to matched filters 12.sub.1, to 12.sub.4, wherein they are despread. A spreading code of a desired signal is used as a despreading code in the matched filter 12.sub.1, and a plurality of spreading codes which are orthogonal to the spreading...http://www.google.de/patents/US5757845?utm_source=gb-gplus-sharePatent US5757845 - Adaptive spread spectrum receiver Ver�ffentlichungsnummerUS5757845 APublikationstypErteilung Anmeldenummer08/446,717 Ver�ffentlichungsdatum26. Mai 1998Eingetragen9. Febr. 1995 Priorit�tsdatum10. Febr. 1994Auch ver�ffentlicht unterCN1059530CCN1122176AEP0701344A1EP0701344A4EP0701344B1WO1995022214A1 ErfinderKazuhiko FukawaHiroshi SuzukiUrspr�nglich Bevollm�chtigterNtt Mobile Communications NetworkNtt Mobile Communications Network, Inc. US-Klassifikation375/152375/E01.18375/E01.36370/342375/E01.32Internationale KlassifikationH04B1/707H04B1/713 UnternehmensklassifikationH04B1/7115H04B1/7093 Europ�ische KlassifikationH04B1/7093H04B1/7115ReferenzenPatentzitate (9) Referenziert von (56)Externe LinksUSPTO USPTO-Zuordnung EspacenetAdaptive spread spectrum receiverUS 5757845 A Zusammenfassung Sampled signals SPS of a spread-spectrum received signal are fed to matched filters 12.sub.1, to 12.sub.4, wherein they are despread. A spreading code of a desired signal is used as a despreading code in the matched filter 12.sub.1, and a plurality of spreading codes which are orthogonal to the spreading code of the desired signal and orthogonal to one another are used for despreading in the matched filters 12.sub.2 to 12.sub.4. Outputs x.sub.1 to x.sub.4 from the filters 12.sub.1 to 12.sub.4 are multipled by weighting coefficients w.sub.1 to w.sub.4 ; the multiplied outputs are linearly combined into a combined signal DCS. A coefficient control part 37 is supplied with the signals x.sub.1 to X.sub.4 to be multiplied by the weighting coefficients w.sub.1 to w.sub.4 and the combined signal DCS and determines the weighting coefficients w.sub.1 to w.sub.4 by an algorithm that minimizes the average power of the combined signal DCS under a constraint on the weighting coefficients w.sub.1 to w.sub.4.
We claim: 1. An adaptive spread spectrum receiver comprising: sample means which samples a received signal at regular time intervals and outputs sampled signals; and signal extraction means which despreads and linearly combines said sampled signals and outputs a combined signal; wherein said signal extraction means comprises: despreading/combining means which despreads and linearly combines said sampled signals by use of weighting coefficients to obtain said combined signal and outputs said combined signal and signals to be multiplied by said weighting coefficients; and coefficient control means which is supplied with said signal to be multiplied and said combined signal and calculates said weighting coefficients which minimize the average power of said combined signal under a constraint on said weighting coefficients.
2. The adaptive spread spectrum receiver of claim 1, wherein said despreading/combining means comprises: despreading means which despreads said sampled signals by a plurality of despreading codes into a plurality of despread signals and outputs said plurality of despread signals as said signals to be multiplied; and linearly combining means which multiplies said plurality of despread signals by said weighting coefficients and combines the multiplied signals into said combined signal.
5. The adaptive spread spectrum receiver of claim 1, wherein said sample means is means which samples one or more received signals at regular time intervals and outputs one or more sampled signals; wherein said signal extraction means is diversity signal extraction means which despreads and linearly combines said sampled signals by said despreading/combining means and said coefficient control means and outputs a plurality of branch combined signals; and which comprises diversity demodulation means which diversity-combines/demodulates said plurality of branch combined signals and outputs a decision signal.
6. The adaptive spread spectrum receiver of claim 5, wherein said sample means is means which outputs, as said sampled signals, received signals generated from received waves of a plurality of antennas; and wherein said diversity signal extraction means is means which comprises said despreading/combining means and said coefficient control means for each of said sampled signals and outputs combined signals outputted by said plurality of despreading/combining means as said plurality of branch combined signals.
7. The adaptive spread spectrum receiver of claim 5, wherein said sample means is means which outputs, as a single sampled signal, a received signal generated from a received wave of a single antenna; and wherein said diversity signal extraction means is means which performs processings of said despreading/combining means and said coefficient control means with different timings and outputs the results of such signal processings as said plurality of branch combined signals.
8. The adaptive spread spectrum receiver of claim 7, wherein said despreading/combining means comprises: despreading means which despreads said sampled signal by a plurality of despreading codes to obtain a plurality of despread signals and outputs them as said signals to be multiplied; and a plurality of linearly combining means which sample said plurality of despread signals with different timings, multiply the sampled despread signals of different timings by weighting coefficients for the respective timings and combines the multiplied signals into said branch combined signals; and wherein said coefficient control means is a plurality of coefficient control means which uses, as said signals to be multiplied, said sampled despread signals sampled with said different timings and generate said weighting coefficients for said different timings.
9. The adaptive spread spectrum receiver of claim 8, wherein said plurality of despreading codes of said despreading means are a spreading code of a desired signal and one or more spreading codes orthogonal thereto; and wherein the timing for sampling said despread signals despread by said one or more spreading codes orthogonal to said spreading code of said desired signal is not synchronized with the timing of said despread signal despread by said spreading code of said desired signal in terms of sampling phase.
11. The adaptive spread spectrum receiver of claim 7, wherein said despreading/combining means is means which convolutes a sequence of said sampled signals and a sequence of said weighting coefficients by a transversal filter and outputs said plurality of branch combined signals and said sequence of sampled signals as said signals to be multiplied; and wherein said coefficient control means is means which outputs said weighting coefficients for said different timings on the basis of said signals to be multiplied and said plurality of branch combined signals.
Sampled signals, by sampling received signals from respective antennas at regular time intervals, are fed into input terminals 11.sub.1 to 11.sub.4. Connected to the input terminals 11.sub.1 to 11.sub.4 are matched filters (MF) 12.sub.1 to 12.sub.4 of the same construction, respectively. The matched filters 12.sub.1 to 12.sub.4 each detect the correlation between the spreading code of the desired signal and the sampled signals fed thereto. The outputs of the matched filters 12.sub.1 to 12.sub.4 each contain interfering signal components due to the correlation with spreading codes of the other users, in addition to the desired signal. The outputs of the matched filters 12.sub.1 to 12.sub.4 are multiplied by weighting coefficients w.sub.1 to w.sub.4 in multipliers 13.sub.1 to 13.sub.4, respectively, and the outputs of the multipliers are combined together by an adder 15 into a combined signal. The combined signal is fed to a decision circuit 16 for hard decision; the resulting decision signal is outputted at an output terminal 17. A subtractor 18 calculates the difference between the combined signal and the decision signal and outputs it as an estimation error E; a coefficient control part 19 uses the estimation error E and the input signals MS of the multipliers 13.sub.1 to 13.sub.4 to control the weighting coefficients w.sub.1 to W.sub.4 by employing an adaptive algorithm that minimizes the square of the estimation error E. That is, the weighting coefficients are controlled so that the average power of interference signal components and a noise signal contained in the combined output signal of the adder 15 becomes minimum.
This method requires a plurality of antennas for diversity reception, that is, requires a large amount of hardware complexity; hence, the method is hard to employ in mobile radio receivers. Furthermore, the signals received in the respective branches are combined so that they become in-phase with one another when no interfering signals exist, and when interference signals exist, they are combined so that the interference signal components are removed. To perform this, the phases and amplitudes of the weighting coefficients w.sub.1 to W.sub.4 are adaptively controlled; this control is very difficult to realize in the fast fading environments as in mobile radio communications.
PRIOR ART EXAMPLE 2 In FIG. 2, there is shown the configuration of the prior art that has a diversity effect on a multipath signal received by a single antenna and performs interference cancellation (Abdulrahman, M., D. D. Falconer and A. U. H. Sheikh, "Equalization for Interference Cancellation In Spread Spectrum Multiple Access Systems," Proc. 42nd Vehicular Technology Conference, pp. 71-74, May 1992). In FIG. 2, the multipliers 13.sub.1 to 13.sub.4 and the adder 15, which form the combining circuit in FIG. 1, are replaced with a transversal filter 21 that is equivalent to their combination. The received signal outputted by the single antenna is sampled at regular time intervals and fed as a sampled signal SPS to the input terminal 11. The sampled signal SPS is fed to the matched filter 12, which calculates its correlation with the spreading code of a desired signal to obtain the despread signal. The despread signal reflects the impulse response of the multipath channel and contains multipath delayed signal components of different delay times. The despread signal containing the multipath delayed signal components is inputted into the transversal filter 21 and products of respective tap outputs of the transversal filter 21 multiplied by tap coefficients supplied as a tap coefficient vector W to the taps of the transversal filter 21 are added up together thereby obtaining convolution between the despread signal and the tap coefficients. As the result of this, a combined signal free from interference is provided. The decision circuit 16 inputs thereinto the combined signal and makes a signal decision by hard decision and feeds the decision signal to the output terminal 17. The subtractor 18 outputs, as the estimation error E, the difference between the combined signal and the decision signal. The coefficient control part 19 inputs thereinto the estimation error E and an output signal sequence MS of the matched filter 12 which is fed to the transversal filter 21, and controls the tap coefficient W of the transversal filter 21 so that the square of the estimation error E becomes minimum.
In FIG. 3, the input sampled signal through the input terminal 11 is fed to the matched filters 12.sub.1 to 12.sub.4, wherein its correlation with spreading codes of respective users is calculated. The matched filter .sup.12 1 uses the spreading code of the desired signal and the other matched filters 12.sub.2 to 12.sub.4 use spreading codes of other users. The output of the matched filter 12.sub.1 contains interference signals as well as the desired signal. Since the interference signal can be expressed as a linear combination of output signals of the matched filters 12.sub.2 to 12.sub.4, it is possible to completely prevent the interference signal from being contained in the combined signal which is the output of the adder 15, by adjusting or controlling the weighting coefficients w.sub.1 to w.sub.4 by which the outputs of the matched filters 12.sub.1 to 12.sub.4 are multiplied in the multipliers 13.sub.1 to 13.sub.4, respectively. This is mathematically equivalent to extracting a component orthogonal to the interference signal as a despread signal of the desired signal. In the decorrelator with such an operation, an inverse matrix calculator 25 calculates an inverse matrix of the correlation matrix of the spreading codes on the basis of information about the spreading codes and reception timing of the users, and outputs particular elements of the inverse matrix as the weighting coefficients w.sub.1 to W.sub.4.
At first, the input sampled signal SPS via the input terminal 11 is fed to the four matched filters 12.sub.1 to 12.sub.4 forming the despreading part 38. The matched filters 12.sub.1 to 12.sub.4 each calculate the correlation between the sampled signal and the spreading code; the resulting despread signals x.sub.1 (i) to x.sub.4 (i) at a discrete time instant i are outputted as the signals to be multiplied MPS. In this case, the matched filter 12.sub.1 uses a spreading code of a desired signal and the other matched filters 12.sub.2 to 12.sub.4 each use predetermined spreading codes different from that of the desired signal. Unlike in the prior art example of FIG. 3, the spreading codes for use in the matched filters 12.sub.2 to 12.sub.4 need not be the same as those of other users. Furthermore, the spreading code that is used in any one of the matched filters 12.sub.1 to 12.sub.4 need not always be the spreading code of the desired signal; the cross-correlation between the spreading code for use in any one of the matched filters and the spreading code of the desired signal needs only to be sufficiently higher than the cross-correlation between the spreading codes for the other matched filters and the spreading code of the desired signal.
The linearly combining part 39 is composed of the multipliers 13.sub.1 to 13.sub.4 and the adder 15. The despread signals x.sub.1 (i) to x.sub.4 (i) of the despreading part 38 are multiplied by the weighting coefficients w.sub.1 to W.sub.4, respectively, and the multiplied outputs are added together by the adder 15 to obtain the combined signal DCS, which is fed to the output terminal 26. The coefficient control part 37 is supplied with the despread signals x.sub.1 (i) to x.sub.4 (i) and the combined signal DCS and calculates the weighting coefficients w.sub.1 to W.sub.4 by the algorithm that minimizes the average power of the combined signal DCS under the weighting coefficient constraint. The matched filters 12.sub.1 to 12.sub.4 can be replaced with correlators--this applies to the matched filters described hereinafter.
In this embodiment, letting the optimum value of the four-dimensional weighting coefficient vector W= w.sub.1 *, w.sub.2 *, w.sub.3 *, w.sub.4 *!.sup.T be represented by Wo= wo.sub.1 *, wo.sub.2 *, wo.sub.3 *, wo.sub.4 *!.sup.T, it is evident from the aforementioned literature by Frost, for example, that the optimum value Wo under the weighting coefficient constraint is given by the following equation. In the above, the symbol * denotes a complex conjugate and .sup.T transposition.
Wo=&#945;R.sup.-1 S                                       (1)
where α a scalar value, R is a four-by-four correlation matrix of the despread signals and S is a four-dimensional steering vector. Using a despread signal vector X(i)= x.sub.1 (i), x.sub.2 (i), x.sub.3 (i), x.sub.4 (i)!.sup.T, the correlation matrix R is given as follows:
R=&amp;lt;X(i)x.sup.H (i)&amp;gt;                                        (2)
where i is a discrete time instant using the symbol duration T as a unit, wo.sub.j is the optimum value of the weighting coefficient w.sub.j, x.sub.j (i) is the despread signal of a j-th matched filter 12.sub.j (j=1, 2, 3, 4) at the time instant i, .sup.H denotes complex conjugate transposition and &lt;&gt; denotes an ensemble average. The matrix R can be approximated as follows:
where N.sub.t is a large natural number. The larger the number N.sub.t, the higher the approximation accuracy; though dependent on the system, the natural number N.sub.t may preferably be set to a value such that a change in the communication conditions, such as the initiation of communication by another user, will not occur during the period N.sub.t.
The steering vector S is, in this case, a vector whose element is the cross-correlation ρ.sub.jk between the spreading code j (j=1) of the desired signal and the spreading code k that is used in the matched filter 12, and it is given as follows:
S= &#961;11, &#961;12, &#961;13, &#961;14!.sup.T               (4)
The rate at which the desired signal is contained in the output of the multiplier 13.sub.1 is wo.sub.1ρ11. Similarly, the rates at which the desired wave is contained in the outputs of the multipliers 13.sub.2 to 13.sub.4 are wo.sub.2ρ12., wo.sub.3ρ13 and wo.sub.4ρ14, respectively. Since the combined signal DCS is the sum of the outputs of the multipliers 13.sub.1 to 13.sub.4, the rate at which the desired signal is contained in the combined signal DCS is (wo.sub.1ρ11 +wo.sub.2ρ12 +wo.sub.3ρ13 +wo.sub.4ρ14), which corresponds to W.sup.H S. When the weighting coefficients are controlled so that the signal level of the desired signal contained in the combined signal remains constant, the constraint on the weighting coefficients is expressed as follows:
As algorithms for requiring the optimum value Wo, there are a method of calculating it directly by using Eqs. (1), (3) and (4) and a method of calculating it in a recursive form. The recursive form can be derived by use of a theory which utilizes the lemma of an inverse matrix about R.N.sub.t, taking into account the constraint on the weighting coefficient W. The algorithm thus derived is such as follows:
W(i)=&#946;.sub.i W(i-1)-&#946;.sub.i K(i)Y*(i)            (8)
where Y(i) is the combined signal, P(i) is the inverse matrix of R.N.sub.t at the time instant i, K(i) is the Kalman gain vector and βi=i/(i-1), where i≦2. In the steady state, W(i) converges to α.sup.-1 Wo which is a constant multiple of Wo expressed by Eq. (1). Whether it converges to α.sup.-1 Wo or Wo, the ratio between the signal power of the desired signal and that of the interference signals contained in the combined signal remains unchanged and the transmission performance also remains unchanged accordingly. Therefore, a description will be given of the calculation of α.sup.-1 Wo which involves less computational complexity. In this recursive form the steering vector is contained in the initial condition {W(1)=P(1)S}.
The despreading part 38 is formed by the matched filter 12.sub.1 and orthogonal code filters (OCF) 41.sub.1 to 41.sub.3 ; the matched filter 12.sub.1 is assigned the spreading code of the desired signal and the orthogonal code filters 41.sub.1 to 41.sub.3 are assigned spreading codes that are orthogonal to the spreading code of the desired signal and orthogonal to one another. The orthogonal code filters constitute an orthogonalization part 42. In the orthogonalization part 42 there is no need of using the same spreading codes as those of other users.
Even when the spreading code of the desired signal is used as the despreading code for the matched filter 12.sub.1, signal components of the desired and the interference wave are contained in the despread signal component. In such an instance, however, the signal component of the desired signal is not contained in the output signals of the orthogonalization part 42 but only the signal components of the interference signals are contained. By fixing the weighting coefficient for the output signal of the matched filter 12.sub.1 as a constant and controlling the weighting coefficients in such a manner as to minimize the power of the combined signal, it is possible to keep constant the power of the signal component of the desired signal contained in the combined signal and minimize the power of the signal component of the interference signals in the combined signal. The optimum value Wo of the four-dimensional weighting coefficient vector in this case is calculated in the same manner as in the aforementioned embodiment. In this instance, however, since there is no correlation between the despreading codes in the orthogonalization part 42 and the spreading code of the desired signal, the steering vector S becomes as follows:
When the steering vector S is set to such a value as given by Eq. (11), the coefficient constraint corresponds to setting Wo.sub.1 =1. The same algorithms as referred to previously in respect of Embodiment 1 can be applied to calculate the optimum value Wo.
The received signal is fed via the input terminal 31 to the sampling circuit 32, which samples it and outputs the sampled signal SPS. The signal extraction part 33 receives the sampled signal SPS, performs the despreading and linearly combining operations and outputs the combined signal. The demodulation part 34 demodulates the combined signal and feeds the decision signal to the output terminal 17. The signal extraction part 33 comprises the transversal filter 43 and the coefficient control part 37. The transversal filter 43 comprises: a plurality of cascade-connected delay elements 43D.sub.1, 43D.sub.2 and 43D.sub.3 each having a delay time equal to one chip duration Tc; multipliers 43M.sub.0 to 43M.sub.3 for multiplying the input of the first delay stage 43D.sub.1 and the outputs of the respective delay elements by the tap coefficients W (w.sub.1, w.sub.2, w.sub.3, w.sub.4); and adders 43A.sub.1, 43A.sub.2 and 43A.sub.3 for adding the multiplied outputs. The transversal filter 43 operates equivalently to the combination of the despreading part 38 and the linearly combining part 39 in FIG. 6; it convolutes the sampled signal SPS and the tap coefficients W and outputs the combined signal DCS. The coefficient control part 37 inputs thereinto the sampled signal MPS set in the transversal filter 43 and the combined signal DCS and calculates the tap coefficients W by the algorithm that minimizes the average power of the combined signal DCS under the constraint on the tap coefficients W.
With the scheme using the transversal filter 43, since this filter possesses the function of the despreading part 38 including the orthogonalization part 42 in FIG. 6, the steering vector S becomes the product of a vector C.sub.1 using the spreading code of the desired signal as its elements and the identity matrix I; this is expressed by the following equation.
The rate of the desired signal component in the output of the transversal filter 43 can be expressed by W.sup.H C.sub.1 ; it will be seen that this component could be held constant by using C.sub.1 in place of S in W.sup.H S=1 of Eq. (5) which shows the constraint on the weighting coefficients. The algorithms that can be used to calculate the optimum value Wo are the same as those mentioned previously with reference to Embodiment 1.
The combined signal DCS is inputted into a branch metric generator 45 via the input terminal 26. A combined signal at the current time kT and combined signals at times (k-1)T to (k-4)T respectively set in four delay circuits 46.sub.1 to 46.sub.4, each having a delay time equal to one symbol duration T, are inversely modulated by a symbol sequence candidate {a.sub.m (k)} which is fed by a maximum likelihood sequence estimator 47, whereby inversely modulated signals are generated. Incidentally, the modulation scheme in this example is a modulation scheme in which the amplitude and the inverse modulation processing can be done by multiplying the combined signal DCS by a complex conjugate {a.sub.m *(k)} of the symbol sequence candidate in multipliers 48.sub.1 to 48.sub.4. Next, multipliers 49.sub.1 to 49.sub.4 and an adder 51 estimate an inversely modulated signal at the time kT by using the inversely modulated signals at times (k-1)T to (k-4)T and output an inversely modulated signal estimated value. Assuming that the channel variation is slow, the coefficients of the multipliers 49.sub.1 to 49.sub.4 need only to be set to such a value as for averaging, 1/4 in this example. When the symbol sequence candidate {a.sub.m (k)} coincides with the true value of the transmitted symbol sequence, the inversely modulated signals approximately coincide with the carrier signal; consequently, the above-mentioned average value that is fed by the adder 51 becomes the carrier component of the received signal.
A subtractor 52 detects and outputs a difference ε between the inversely modulated signal at time kT, the output of a multiplier 48.sub.0 which has multiplied the input sampled signal SPS, by the complex conjugate a.sub.m *(k) of a complex symbol candidate at time kT and the inversely modulated signal estimated value (the output from the adder 51). The output ε is squared by a square calculator 53, by which the squared output is fed as a likelihood information signal to the maximum likelihood sequence estimator 47. The maximum likelihood sequence estimator 47 uses the likelihood information signal to calculate a log likelihood function, then selects by the Viterbi algorithm a symbol sequence candidate that maximizes the log likelihood function, and outputs it as the decision signal to the output terminal 17. While this embodiment has been described to employ the four delay circuits 46.sub.1 to 46.sub.4, the number of delay circuits is not limited specifically to four but may also be extended to L(L≧1).
In FIG. 10 there are indicated by the curves 10a and 10b the results of the average bit error rate performance by computer simulations conducted to demonstrate the effectiveness of the present invention. The abscissa represents the maximum Doppler shift frequency f.sub.D of the received signal which is caused by the movement of the receiver, and E.sub.b /N.sub.o represents the received signal power versus noise power ratio per bit. In the computer simulations, the orthogonalization was done by the configuration of Embodiment 2 (FIG. 7) and the predictive coherent detector depicted in FIG. 9 was used as the demodulation part 34. For the purpose of comparison, the results of the average bit error rate performance obtained by employing the DS-CDMA adaptive interference canceller shown in FIG. 4 are indicated by the curves 10c and 10d. The process gain Gp used was 16, the number of users was 16 and the received timing of each user was assumed to be synchronized with one another. The modulation scheme used is a 10 kb/s BPSK modulation and the spreading codes used are those which have a cross-correlation under 0.25. The channel model is a Rayleigh fading model. It will be seen from FIG. 10 that the present invention is superior to the conventional DS-CDMA adaptive interference canceller.
EMBODIMENT 5 In mobile radio communications, the diversity reception scheme is utilized with a view to suppressing severe degradation of the transmission performance by the fading-induced variation of the propagation path. FIG. 11 illustrates an embodiment of the present invention applied to the diversity reception. The sampling circuit 32 samples one or more received signals at regular time intervals and outputs one or more sampled signals. In the case of antenna diversity, a plurality of received signals are handled, whereas in the case of path diversity, a single received signal is handled. The illustrated example is a two-branch antenna diversity scheme. A diversity signal extraction part 55 comprises a despreading/combining part 56 which inputs thereinto sampled signals SPS.sub.1 and SPS.sub.2 and despreads and linearly combines them, and a coefficient control part 57; the diversity signal extraction part outputs a plurality of branch combined signals DCS. The despreading/combining part 56 and the coefficient control part 57 are similar to the despreading/combining part 36 and the coefficient control part 37 in FIG. 6, but the diversity signal extraction part differs from the signal extraction part in FIG. 6 in that the former outputs the plurality of branch combined signals DCS.sub.1 and DCS.sub.2. A diversity demodulation part (DIV-DEM) 58 combines and demodulates the plurality of branch combined signals DCS.sub.1 and DCS.sub.2 and outputs a decision signal. The timing control part effects timing control for each part.
FIGS. 12A, 12B and 12C illustrate examples of the construction of the diversity demodulation part 58 for use in the two-branch diversity reception scheme. These examples are conventionally known. In FIG. 12A there is shown an extended differential detection structure. The combined signals DCS.sub.1 and DCS.sub.2, which are inputted via input terminals 26.sub.1 and 26.sub.2 for respective diversity branches, and signals, which are generated by delaying the combined signals for one symbol duration T in delay elements 58A.sub.1 and 58A.sub.2 and subjecting the delayed signals to a complex conjugate calculation in complex conjugate calculation parts 58B.sub.1 and 58B.sub.2, are multiplied by multipliers 58C.sub.1 and 58C.sub.2, respectively; thus, the received signal is differentially detected. The multiplied outputs are added together by an adder 61, and the added output is fed to the decision circuit 16, which makes the signal decision by hard decision and provides the decision signal to the output terminal 17.
FIG. 12B shows an extension of the coherent detection to the diversity reception scheme. The combined signals DCS.sub.1 and DCS.sub.2, which are inputted via the input terminals 26.sub.1 and 26.sub.2 for respective diversity branches, are fed to multipliers 58D.sub.1 and 58D.sub.2, which multiplies them by estimated carrier synchronizing signals SY.sub.1 and SY.sub.2 from a control part 62, respectively, and outputs carrier-phase-synchronized signals. The multiplied signals are added together by the adder 61, whose added output is fed to the decision circuit 16. The decision circuit 16 makes the signal decision by hard decision and provided the decision signal to the output terminal 17. The subtractor 18 outputs, as an estimation error signal, the difference between the input of the decision circuit 16 and the output therefrom. A control circuit 62 is supplied with the estimation error signal outputted by the subtractor 18, the combined signals DCS.sub.1 and DCS.sub.2 via the input terminals 26.sub.1 and 26.sub.2, and estimates the above-mentioned estimated carrier synchronizing signals SY.sub.1 and SY.sub.2 so that the square of the absolute value of the estimation error becomes minimum.
FIG. 12C illustrates an extension of the predictive coherent detection of FIG. 9 to the diversity reception scheme. Branch metric generators 45.sub.1 and 45.sub.2, which are identical in construction to that in FIG. 9, are each set for each diversity branch; these branch metric generators are supplied with the combined signals DCS.sub.1 and DCS.sub.2, respectively, and the symbol sequence candidate outputted by the maximum likelihood sequence estimator 47 in common to them and outputs the likelihood information signals. The maximum likelihood sequence estimator 47 calculates the log likelihood function on the basis of the likelihood information signals, then selects by the Viterbi algorithm a symbol sequence candidate that maximizes the log likelihood function, and feeds it as the decision signal to the output terminal 17.
EMBODIMENT 6 In FIG. 13 there is illustrated a concrete embodiment for the antenna diversity scheme. In the antenna diversity scheme, the sampling circuit 32 receives, as a plurality of received signals, the signals generated from the received waves of a plurality (two in this example) of antennas and outputs a plurality of sampled signals SPS.sub.1 and SPS.sub.2 sampled in sampling parts (SMP) 32.sub.1 and 32.sub.2. In the diversity signal extraction part 55, branch signal extraction parts 33.sub.1 and 33.sub.2, each composed of the despreading/combining part 36 and the coefficient control part 37 described previously in respect of FIGS. 5, 6 and 7, are set corresponding to the sampled signals, respectively; the diversity signal extraction part outputs a plurality of branch combined signals DCS.sub.1 and DCS.sub.2. Incidentally, the branch signal extraction parts 33.sub.1 and 33.sub.2 may each be formed by the signal extraction part 33 with the transversal filter 43 described previously with respect to FIG. 8.
It is an example of the two-branch antenna diversity scheme that is depicted in FIG. 13. Received signals of first and second diversity branches are inputted into the sampling circuit via input terminals 31.sub.1 and 31.sub.2. The received signals of the respective diversity branches are sampled by the sampling circuits 32.sub.1 and 32.sub.2 and the sampled signals are fed to the branch signal extraction parts 33.sub.1 and 33.sub.2, by which the combined signals DCS.sub.1 and DCS.sub.2 are generated. The diversity demodulation part 58 combines the combined signals DCS.sub.1 and DCS.sub.2 of the respective diversity branches and demodulates the combined signals, then outputs the decision signal to the output terminal 17. While this embodiment has been described in the case of the two-branch diversity reception scheme, an extension to the diversity reception scheme with three or more branches can be implemented with ease.
EMBODIMENT 7 When the When the transmitted wave propagates over two different paths of a multipath channel, the impulse response can be expressed by two impulses 64.sub.1 and 64.sub.2 as shown in FIG. 14. The wave that propagates over a path 1 reached the receiving end with a delay time t.sub.1 ; the wave that propagates over a path 2 reaches the receiving end with a delay time t.sub.2.
FIG. 15 illustrates an example of the path diversity configuration. In the path diversity scheme, the sampling circuit (SMP) 32 samples a single received signal generated from the received wave of a single antenna and outputs a single sampled signal SPS. The despreading/combining processing and the coefficient control processing in the diversity signal extraction part 55 are carried out at a plurality of different timings t.sub.1 and t.sub.2 of the sampled signal SPS, and a plurality of branch combined signals DCS.sub.1 and DCS.sub.2 are outputted. The timing signal for the processing is generated by the timing control part 35. The diversity demodulation part (DIV-DEM) 58 combines and demodulates the branch combined signals DCS.sub.1 and DCS.sub.2.
EMBODIMENT 8 In FIG. 16 there is shown an embodiment for generating the branch combined signal of the path diversity scheme by a despreading part. The diversity signal extraction part 55 is comprised of one despreading part 38, a plurality of sample-hold (SH) parts 65.sub.1 and 65.sub.2 and a plurality of combining control parts 83.sub.1 and 83.sub.2 each composed of the linearly combining part 39 and the coefficient control part 37 both shown in FIG. 6. As is the case with FIG. 6, the despreading part 38 despreads a single sampled signal SPS by a plurality of despreading codes and outputs a plurality of despread signals. These despread signals are fed to the sample-hold parts 65.sub.1 and 65.sub.2, wherein they are sampled at the timing t.sub.1 and t.sub.2, respectively. The thus sampled despread signals are fed to the combining control parts 83.sub.1 and 83.sub.2, which multiply them by respective weighting coefficients and combine them, then output the branch combined signals DCS.sub.1 and DCS.sub.2. In this case, the weighting coefficients are adaptively controlled so that the combined output under the coefficient-constraint becomes minimum. Incidentally, the propagation path is a two-path model as is the case with FIG. 14.
The sampled signal SPS via the input terminal 11 is despread by a plurality of despreading codes in the despreading part 38, from which a plurality of despread signals are provided. These despread signals are sampled and held by the sample-hold part 65.sub.1 with the timing t.sub.1 and by the sample-hold part 65.sub.2 with the timing t.sub.2, respectively. The combining control part 83.sub.1 combines the plurality of despread signals outputted by the sample-hold part 65.sub.1 and the combining control part 83.sub.2 combines the plurality of despread signals outputted by the sample-hold part 65.sub.2. The despreading part 38, the sample-hold parts 65.sub.1 and 65.sub.2 and combining control parts 83.sub.1 and 83.sub.2 correspond to the diversity signal extraction part as a whole. The diversity demodulation part 58 combines and demodulates the combined signals outputted by the combining control parts 83.sub.1 and 83.sub.2, and feeds a decision signal to the output terminal 17.
EMBODIMENT 9 FIG. 17 illustrates an embodiment in which branch combined signals of the path diversity scheme are generated by the despreading part 38 using an orthogonalization part. The propagation path is a two-path model. In the despreading part 38, as described previously with reference to FIG. 7, the spreading code of the desired signal is used in the matched filter 12.sub.1 and the plurality of spreading codes, which are orthogonal to the spreading code of the desired signal and orthogonal to one another, are used in the orthogonalization part 42. The timing t.sub.3 for sampling the output of the orthogonalization part 42 is synchronous with the sample timing t.sub.1 as in the embodiment of FIG. 16, but under special conditions they may be asynchronous (t.sub.3 ≠t.sub.1) as shown.
The sampled signal SPS via the input terminal 11 is fed to the despreading part 38. The despreading part 38 is identical with that depicted in FIG. 7, for instance; it comprises the matched filter 12.sub.1 which uses the spreading code of the desired signal and the orthogonalization part 42 which uses the plurality of spreading codes orthogonal to the above. The output signal of the matched filter 12.sub.1, that is, the despread signal produced by despreading the sampled signal with the spreading code of the desired signal, is inputted into the sample-hold parts 65.sub.1 and 65.sub.2. The sample-hold part 65.sub.1 samples the above-mentioned despread signal at the timing t.sub.1 and holds it for the symbol duration T, and the sample-hold part 65.sub.2 similarly samples the despreading signal at the timing t.sub.2 and holds it for the symbol duration T. Three output signals of the orthogonalization part 42, that is, three despread signals generated by despreading the sampled signal with three despreading codes orthogonal to the spreading code of the desired signal, are fed to three sample-hold parts 65.sub.3 to 65.sub.5, respectively. The sample-hold parts 65.sub.3 to 65.sub.5 sample the three despread signals at the timing t.sub.3 which is not always synchronous with the timing t.sub.1 and t.sub.2 and hold them for the symbol duration T. The combining control part 83.sub.1 linearly combines the output signals of the sample-hold part 65.sub.1 and 65.sub.3 to 65.sub.5 and outputs the combined signal DCS.sub.1. Similarly, the combining control part 83.sub.2 linearly combines the output signals of the sample-hold parts 65.sub.2 and 65.sub.3 to 65.sub.5 and outputs the combined signal DCS.sub.2. The diversity demodulation part 58 combines and demodulates the combined signals of the combining control parts 83.sub.1 and 83.sub.2 and feeds a decision signal to the output terminal 17.
This configuration can gain the path diversity effect because the desired signal which passes over the two different paths corresponding to the sample timing t.sub.1 and t.sub.2 are regarded as independent signals. With respect to interference signal components, however, it is necessary to meet conditions that they undergo closely correlated changes at these two sample timings and that an interference signal component closely correlated to those at the timings t.sub.1 and t.sub.2 also be sampled at the timing t.sub.3 different from those of t.sub.1 and t.sub.2. While this embodiment has been described with reference to the two-path model for the propagation path, the configuration of this embodiment can easily be extended to the path diversity scheme with three or more paths. By sampling and holding the plurality of despread signals of the orthogonalization part 42 with only one timing t.sub.3 as described above, the number of sample-hold parts used can be reduced accordingly.
The despreading part 38 is composed of the same matched filter 12.sub.1 as in FIG. 17 and orthogonalization parts of the same number as that of paths to be considered. In this example, the propagation path is a two-path model and two orthogonalization parts 42.sub.1 and 42.sub.2 are set. Let t.sub.1 represent the timing of direct path of a desired signal that are received under the propagation over the two paths and t.sub.2 the timing of delayed paths. Furthermore, let Cd denote the spreading code of the desired signal, Cd(+δ) denote a code that is obtained by shifting the chip of the spreading code Cd of the desired signal in the positive direction by the timing difference, t.sub.2 -t.sub.1 =67 , between the two paths, and Cd(-δ) denote a code that is obtained by shifting the chip of the spreading code by δ in the negative direction. The matched filter 12.sub.1 uses the spreading code Cd of the desired signal, the orthogonalization part 42.sub.1 uses a plurality of despreading codes orthogonal to both the spreading code Cd of the desired signal and the shift code Cd(+δ), and the orthogonalization part 42.sub.2 uses a plurality of spreading codes orthogonal to both of the spreading code Cd and the shift code Cd(-δ). The orthogonalization part 42.sub.1 operates on the basis of the timing t.sub.1 of the direct path. On the other hand, the orthogonalization part 42.sub.2 operates on the basis of the timing t.sub.2 of the delayed path and its output signals are sampled in the sample-hold parts 65.sub.5 and 65.sub.6 with the timing t.sub.2 of the delayed path. Incidentally, the number of output signals outputted by each of the orthogonalization parts 42.sub.1 and 42.sub.2 is two, smaller than the number of paths by one, unlike in the case of the orthogonalization part in FIG. 7.
The sampled signal is inputted into the despreading part via the input terminal 11. The sampled signal SPS is despread by the spreading code of the desired signal in the matched filter 12.sub.1 and the despread signal is fed to the sample-hold parts 65.sub.1 and 65.sub.2. The sample-hold part 65.sub.1 samples the despread signal with the timing t.sub.1 and holds it for the symbol duration T; the sample-hold part 65.sub.2 samples the despread signal with the timing t.sub.2 and holds it for the symbol duration T. The orthogonalization parts 42.sub.1 and 42.sub.2 each output a plurality of despread signals generated by despreading the sampled signal by using the codes orthogonal to both the spreading code of the desired signal and the codes obtained by shifting it in accordance with the timing difference between the propagation paths. These despread signals are fed to the sample-hold parts 65.sub.3, 65.sub.4 and 65.sub.5, 65.sub.6. The sample-hold parts 65.sub.3 and 65.sub.4 sample the plurality of despread signals of the orthogonalization part 42.sub.1 with the timing t.sub.1 and hold them for the symbol duration T. On the other hand, the sample-hold parts 65.sub.5 and 65.sub.6 sample the plurality of despread signals of the orthogonalization part 42.sub.2 with the timing t.sub.2 and hold them for the symbol duration T.
The combining control part 83.sub.1 combines the output signals of the sample-holds parts 65.sub.1, 65.sub.3 and 65.sub.4 and outputs the combined signal DCS.sub.1. As the result of this, the spreading signal components of other users contained in the despread signals with the timing t.sub.1 are removed therefrom. Similarly, the combining control part 83.sub.2 combines the output signals of the sample-hold parts 65.sub.2, 65.sub.5 and 65.sub.6 and outputs the combined signal DCS.sub.2. As the result of this, the spreading signal components of other users contained in the despread signals obtained with the timing t.sub.2 are removed therefrom. The diversity demodulation part 58 diversity-demodulates the combined signals DCS.sub.1 and DCS.sub.2 of the combining control parts 83.sub.1 and 83.sub.2, obtained with the timing t.sub.1 and t.sub.2, and outputs a decision signal to the output terminal 17, as is the case with FIGS. 15, 16 and 17.
EMBODIMENT 11 FIG. 19 illustrates an embodiment designed to generate the branch combined signal of the path diversity scheme by a despreading/combining part using a transversal filter. The propagation path is a two-path model. The result of convolution of the sampled signal SPS and the tap coefficients W of the transversal filter 43 is sampled with different timings t.sub.1 and t.sub.2 corresponding to the propagation paths and the sampled signals are outputted as the plurality of branch combined signals DCS.sub.1 and DCS.sub.2. Furthermore, a sequence of sampled signals SPS is outputted as the signals to be multiplied MPS. The coefficient control parts 37.sub.1 and 37.sub.2 use the signals MPS and the branch combined signals DCS.sub.1 and DCS.sub.2 to output the tap coefficients W.sub.1 and W.sub.2 corresponding to the timings t.sub.1 and t.sub.2, respectively.
The sampled signal SPS via the input terminal 11 is inputted into the transversal filter 43 identical in construction with that shown in FIG. 8, which performs the despreading and linearly combining operations and outputs the combined signal DCS. The combined signal DCS is fed to the sample-hold parts 65.sub.1 and 65.sub.2. The sample-hold part 65.sub.1 samples the combined signal with the timing t.sub.1 and holds it for the symbol duration T, and the sample-hold part 65.sub.2 samples the combined signal with the timing t.sub.2 and holds it for the symbol duration T. The sampled signal sequence MPS (see FIG. 8) set in the transversal filter 43 is fed to the sample-hold parts 65.sub.3 and 65.sub.4, wherein it is sampled with the timings t.sub.1 and t.sub.2 and held for the symbol duration T.
The coefficient control part 37.sub.1 is supplied with the combined signal DCS.sub.1 outputted by the sample-hold part 65.sub.1 and the sampled signal sequence SPS outputted by the sample-hold part 65.sub.3 which is set in the transversal filter 43, whereas the coefficient control part 37.sub.2 is supplied with the composite signal DCS.sub.2 outputted by the sample-hold part 65.sub.2 and the sampled signal sequence MPS outputted by the sample-hold part 65.sub.4 which is set in the transversal filter 43. These coefficient control parts calculate and output the tap coefficients W.sub.1 and W.sub.2 which minimize the average power of the combined signal under the constraints of the tap coefficients W.sub.1 and W.sub.2. A switching circuit 67 selectively sets the tap coefficients W.sub.1 and W.sub.2 outputted by the coefficient control part 37.sub.1 and 37.sub.2 in the transversal filter 43 so that the transversal filter 43 is allowed to output a desired combined signal. The timing control part 35 controls the operation timing of each of the sample-hold parts 65.sub.1 to 65.sub.4 and the switching circuit 67. The diversity demodulation part 58 combines and demodulates the combined signals DCS.sub.1 and DCS.sub.2 outputted by the sample-hold parts 65.sub.1 and 65.sub.2 and feeds a decision signal to the output terminal 17.
FIG. 23A shows two signal points Sp.sub.1 and Sp.sub.2 in the case of generating BPSK modulation in the modulation part 69 in FIG. 22; the in-phase component I(t) of the modulated signal varies with the digital signal DS every period T as shown in FIG. 23B. Incidentally, the quadrature component Q(t) of the modulated signal remains zero. FIG. 23C shows four signal points Sp.sub.1 to Sp.sub.4 in the case of generating QPSK modulation in the modulation part 69; the in-phase component I(t) and the quadrature component Q(t) of the modulated signal vary with the digital signal DS every 2T as depicted in FIG. 23D.
Moreover, the multilevel modulation narrows the frequency band, and hence enables the process gain Gp to be increased. Accordingly, the multiplexity, i.e. the channel capacity can be increased. With the conventional orthogonalization scheme, since no sufficient orthogonality can be obtained, it is impossible to provide a margin of the ratio E.sub.j /N.sub.o by multiplexing, and consequently, the multiplexity has to be further decreased. Since employing multilevel modulation increases the accuracy of orthogonalization, however, it is possible to ensure an increase in the multiplexity by narrowing the frequency band and ensure the operation in the asynchronous system.
EMBODIMENT 3 OF TRANSMITTER Let it be assumed that the transmission signal is generated by applying an RZ signal to the transmission processing part 70 in FIG. 22. Assuming that the channel impulse response is that of a two-path model as depicted in FIG. 14, a direct (preceding) path 29a and a delayed path 29b are combined as shown in FIG. 29A, in which AT is t.sub.2 -t.sub.1. When the modulated signal is generated with a symbol waveform by the RZ signal as shown in FIG. 29B, signal leak of other symbols, that is, the intersymbol interference can be suppressed during one symbol period T, and hence, degradation of the multipath propagation can be suppressed. In the actual application of this example, ΔT in FIG. 29B needs only to be set to a value nearly equal to a delay spread of the propagation path.
EMBODIMENT 5 OF TRANSMITTER In FIGS. 32 and 34 there are illustrated a transmitter and a receiver for use in the case where the adaptive spread spectrum scheme and the frequency hopping one are combined. As is the case with the FIG. 21 embodiment, the transmission processing part 70 of the transmitter modulates the digital signal DS in the modulator 69, spectrum-spreads the modulated signal by the spreading code Cs in the spreading modulation part 71 and feeds the signal to the terminal 72. In FIG. 32, the carrier frequency is caused to hop by a frequency hopping synthesizer 77 in synchronization with carrier signals of all the other users at regular time intervals, then the output outputted by the terminal 72 is multiplied by the frequency-hopped carrier in a multiplier 78 and the multiplied output is transmitted. By the frequency hopping, for example, as depicted in FIG. 33, the carrier frequencies for transmission to all the users #1 to #K are caused to hop between frequencies f.sub.1 and f.sub.2 at regular time intervals T.sub.B, where T.sub.B is an integral multiple of the symbol duration T of the modulated signal.
EMBODIMENT 7 OF TRANSMITTER For example, when the spreading takes place using a spreading code Cs of a pulse train waveform-shaped at a roll-off rate of 1.0, a waveform similar to that shown in FIG. 27 is obtained; letting the chip period be represented by Tc, the pulse waveform becomes zero at sample timing, .+-.(2m+1)Tc/2, m=1, 2, . . . , which is an odd multiple of Tc/2, except sample timing .+-.Tc/2 (in the case of FIG. 27, the pulse waveform also becomes zero at sample timing which is an even multiple of .+-.Tc/2). Hence, the cross-correlation between the same rolled-off spreading codes, after shifting them by an odd multiple of Tc/2 relative to each other, contains many multiplication-additions with zero points and the correlation value becomes small. For example, FIG. 37 shows the configuration of the transmission processing part 70 intended to decrease the correlation value through utilization of the above phenomenon. For half of the modulated signals spread by spreading codes of many users, which are transmitted from a base station, a delay circuit 82 is connected to the output of the spreading part 71 to shift the timing of the modulated and spread transmission signals to such timing as 3Tc/2 or -3Tc/2, by which the multiplication-addition with zeros increases in the cross-correlation between the signals, reducing the correlation value. The lower the cross-correlation, the higher the interference cancelling performance; hence, such a timing shift appreciably improves the transmission performance. As the number of users increases, the number of spreading codes required also increases. On the other hand, the number of spreading codes having mutually high orthogonality in the asynchronous state as well is limited, but by shifting the timing of respective transmission signals among different user groups to reduce the correlation between the spreading codes, it is possible to use the respective spreading code in common to users of different groups.
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